IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. 66, NO.

12, DECEMBER 2017 10899

A Single-Phase Integrated Onboard Battery Charger

Using Propulsion System for Plug-in
Electric Vehicles
Chuan Shi, Student Member, IEEE, Yichao Tang, Student Member, IEEE, and Alireza Khaligh, Senior Member, IEEE

Abstract—Plug-in electric vehicles (PEVs) are equipped with

onboard level-1 or level-2 chargers for home overnight or of-
fice daytime charging. In addition, off-board chargers can pro-
vide fast charging for traveling long distances. However, off-board
high-power chargers are bulky, expensive, and require compre-
hensive evolution of charging infrastructures. An integrated on-
board charger capable of fast charging of PEVs will combine the
benefits of both the conventional onboard and off-board charg-
ers, without additional weight, volume, and cost. In this paper, an
innovative single-phase integrated charger, using the PEV propul-
sion machine and its traction converter, is introduced. The charger
topology is capable of power factor correction and battery volt-
age/current regulation without any bulky add-on components. Ac
machine windings are utilized as mutually coupled inductors, to
construct a two-channel interleaved boost converter. The circuit
analyses of the proposed technology, based on a permanent mag-
net synchronous machine (PMSM), are discussed in details. Exper-
imental results of a 3-kW proof-of-concept prototype are carried
out using a 220-Vrm s , 3-phase, 8-pole PMSM. A nearly unity power
factor and 3.96% total harmonic distortion of input ac current are
acquired with a maximum efficiency of 93.1%.
Index Terms—Electric vehicle, integrated charger, interleaved Fig. 1. Conventional power electronics interfaces for a plug-in electric vehicle.
boost converter, off-board charger, onboard charger, permanent
magnet synchronous machine, power factor correction.
I. INTRODUCTION fast charging [5], [6]. However, they are large, expensive and
ONVENTIONAL level-1 and level-2 onboard chargers require a comprehensive evolution of charging infrastructures.
C are utilized to charge plug-in electric vehicle (PEV) bat-
teries using single-phase power outlets [1], [2]. However, due to
A high-power integrated charger is a promising solution, which
not only provides onboard fast charging without substantially-
their charging power limitations, it might take between 4 to 20 added cost and weight, but also alleviates the range anxiety
hours to fully charge a PEV battery [3], [4]. The level-1 onboard concern among PEV owners.
chargers have power levels up to 1.92 kW (with typical power The onboard and off-board PEIs for a PEV are illustrated
levels as 1.4 kW and 1.9 kW). In the U.S., standard 120 V/15 A in Fig. 1. Typically, a charger consists of two stages: an ac-
single-phase grounded outlets can be utilized for level-1 charg- dc stage for rectification of grid ac voltage and power factor
ing. The level-2 onboard chargers have power levels up to 19.2 correction (PFC) and a dc-dc stage for battery current and volt-
kW (with typical power levels as 3.3 kW and 6.6 kW) [5]. On age regulation [7]–[9]. In addition, PEVs utilize high-power
the other hand, off-board chargers are capable of high-power three-phase-ac machines such as induction machines (IM) or
permanent magnet synchronous machines (PMSM) for elec-
Manuscript received November 12, 2016; revised February 10, 2017 and May tric propulsion (Tesla Model S 310-kW IM, Toyota Rav4 EV
22, 2017; accepted July 14, 2017. Date of publication July 19, 2017; date of 115-kW IM, Nissan Leaf 80-kW PMSM, Chevy Volt 111-kW
current version December 14, 2017. This work is supported by the National
Science Foundation Grant No. 1507546, which is gratefully acknowledged. The PMSM, among others) [10]–[12]. In a typical three-phase-ac
review of this paper was coordinated by Dr. B. Akin. (Corresponding author: machine propulsion system, a bidirectional three-phase traction
Alireza Khaligh.) converter enables battery power flow to the ac machine during
The authors are with theMaryland Power Electronics Laboratory, Department
of Electrical and Computer Engineering, Institute for Systems Research, Uni- propulsion and battery charging during regenerative braking
versity of Maryland, College Park, MD 20742 USA (e-mail: cshi@umd.edu; [10]. In order to acquire high efficiency, the dc-link voltage of
ychtang@umd.edu; khaligh@ece.umd.edu). the traction converter (typically 360 V or 720 V [13]) is preferred
Color versions of one or more of the figures in this paper are available online
at http://ieeexplore.ieee.org. to be higher than the rated peak voltage of the ac machine. Fur-
Digital Object Identifier 10.1109/TVT.2017.2729345 thermore, a propulsion machine’s dc-dc bidirectional converter
0018-9545 © 2017 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission.
See http://www.ieee.org/publications standards/publications/rights/index.html for more information.
10900 IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. 66, NO. 12, DECEMBER 2017

is optionally used to regulate dc-link voltage with a wide range

of battery voltage variations. In such a system, the only external
accessible points of the propulsion machine are the three phase
terminals of the ac machine.
To reduce the size, weight and cost of onboard chargers,
integrated chargers are proposed and studied in the literature
[14]–[20]. Researchers have studied integrated non-isolated Fig. 2. Proposed integrated onboard charger using ac propulsion system.
single-stage chargers that combine ac-dc rectifiers and the
propulsion machine’s dc-dc bidirectional converters [14]–[16].
In comparison to two-stage converters, single-stage topologies
reduce the number and size of bulky passive components, such
as inductors. However, to achieve the ac-dc power factor cor-
rection operation, more transistors and diodes might be needed,
which increases the complexity of circuit and may lead to lower
reliability due to existence of additional switching components.
In addition, the propulsion machine’s dc-dc bidirectional con-
verter is usually rated for much larger power levels than on-
board chargers. Therefore, utilizing an integrated high-power
converter as a low-power converter during relatively low-power
charging might not provide high efficiency.
Other groups of efforts have been focused on integrated charg-
ers using propulsion machine windings and its traction converter
[17]–[23]. The topology in [17] requires bulky add-on compo-
nents. Specially designed machines are needed in topologies
proposed by [18]–[21]. The integrated chargers in [21]–[23]
require access to inaccessible points of machine windings. The
winding/ traction converter rearrangement are necessary in [21],
[24]. An integrated topology is proposed in [25], but it has Fig. 3. Operation modes of integrated battery charger with ac machine:
high possibility of propulsion machine rotation during charging. (a) propulsion mode; (b) charging mode.
The contribution of this paper is proposing a novel and simple
method for single-phase integrated charging, using propulsion
system of plug-in electric vehicles. The proposed integrated machines, and (vii) prevention of propulsion machine rotation
charging method only needs one passive diode bridge to connect during charging.
to two terminals of the existing traction system in an electric ve- This paper is organized as follows. Section II presents the inte-
hicle. There is no need for rearrangement of the system topology grated charger scheme. Section III outlines the electrical model
during the transition from the propulsion mode to the charging of ac propulsion machine, followed by steady-state analyses of
mode. The simple structure of the integrated charging reduces charging operation in Section IV. The experimental results are
the cost to implement the high power-level integrated charg- presented in Section V. Finally, Section VI concludes the work.
ing and significantly reduces the size of integrated charging
system.
By using ac propulsion machine and its traction converter, II. INTEGRATED ONBOARD CHARGER
onboard charging can be easily implemented. The proposed in- As shown in Fig. 1, usually the onboard charger operates
tegrated scheme enables onboard high-power charging without independent of the propulsion system [3]. The proposed inte-
a need for an off-board bulky fast charger, since the propulsion grated single-phase onboard charger is realized by connecting a
machine and its traction converter are rated for higher power very small add-on diode bridge between one of machine phase-
levels. Furthermore, there is no need for bulky add-on compo- terminals, and the negative terminal of the dc-link of the traction
nents, such as large inductors or large capacitors, in the proposed converter, as illustrated in Fig. 2. The proposed scheme is appli-
method. Only a small add-on diode bridge at the rated power cable for any three-phase-ac machine propulsion system with
is needed. In charging mode, the ac machine is used as a three- the only feasible access to machine phase terminals (a, b, c) and
winding coupled inductor, to develop a two-channel interleaved there is no need to have access to the winding neutral point or
boost converter. mid-point of the windings.
The advantages of the proposed topology are summarized During propulsion, battery provides the propulsion power
as: (i) simple structure, (ii) no need for bulky add-on compo- through three-phase traction converter, as shown in Fig. 3(a),
nents (such as large inductors or capacitors), (iii) no need for and the additional diode bridge has no influence on the trac-
motor/traction converter rearrangement, (iv) no need to have tion converter operation. In battery charging mode, as shown in
access to inaccessible points of machine windings, (v) effec- Fig. 3(b), the grid ac line voltage is rectified by the diode-bridge
tive input current ripple cancellation, (vi) extendable to other ac and the propulsion machine windings along with the traction
SHI et al.: SINGLE-PHASE INTEGRATED ONBOARD BATTERY CHARGER USING PROPULSION SYSTEM FOR PLUG-IN ELECTRIC VEHICLES 10901

a conceptual cross-sectional view of a 3-phase, 4-pole PMSM

along with two reference frames (a-b-c frame and d-q frame).
In this model, θr is the rotor angle between a-axis and q-axis
in radians; ω = dθr /dt is the angular velocity of rotation in
rad/sec; Pl is the number of pole pairs; and θe = Pl θr is the
electrical angle between a-axis and q-axis. The corresponding
3-phase electrical model is illustrated in Fig. 4(b). ia , ib and
ic are instantaneous stator phase-currents; va , vb and vc are
the instantaneous stator phase voltages; Rs is the stator resis-
tance; Laa , Lbb and Lcc denote the a-axis, b-axis and c-axis
stator self-inductances; and Lab , Lac and Lbc denote the mu-
tual inductances representing the coupling effects among the
Fig. 4. (a) Cross section of a PMSM; (b) simplified electrical model of a
three-phase stator windings. It can be assumed that in stationary
PMSM. condition θe = π/2, when d-axis is aligned with a-axis.
In the two-phase d-q equivalent circuit model of a PMSM,
converter develop a two-channel interleaved boost converter, the d-axis and q-axis stator self-inductances (Ld and Lq ) can be
which is utilized for PFC and output voltage/current regulation. used to represent the three-phase self and mutual inductances
The ac machine serves as a three-winding coupled inductor [28], as
⎧
for energy storage and ripple cancellation. One of the three- ⎪
⎪ Laa = Ls + Lx cos (2θe ) (1)
⎪
⎪
phase traction converter bridges is connected to the positive ⎪
⎪  
⎪
⎪ 2π
terminal of the diode rectifier. In this case, the S1/S2 bridge is ⎪
⎪ Lbb = Ls + Lx cos 2θe + (2)
⎪
⎪ 3
disabled, while the other two bridges S3/S4 and S5/S6 are used ⎪
⎪  
⎪
⎪
to develop an interleaved boost converter. The switches S1 and ⎪
⎪ 2π
S2 are always open in battery charging mode. The anti-parallel ⎪
⎨ Lcc = Ls + Lx cos 2θe − 3 (3)
diode D2 is reverse biased by the positive output voltage of  
⎪
⎪ 2π
diode rectifier. As for the body diode D1, it only turns on to ⎪
⎪ Lab = −Ms + Lx cos 2θe − (4)
⎪
⎪ 3
charge the output capacitor during the start-up when the output ⎪
⎪
⎪
⎪
capacitor voltage is lower than the input voltage. In steady state, ⎪
⎪ Lbc = −Ms + Lx cos (2θe ) (5)
⎪
⎪
D1 is reverse biased since the output voltage is higher than the ⎪
⎪  
⎪
⎪ 2π
input voltage, due to the boost operation of the converter. To ⎪
⎩ Lac = −Ms + Lx cos 2θe + (6)
meet the low total harmonic distortion (THD) requirement of 3
grid, the converter is operated in continuous conduction mode where, Ls = 1/2(Lq + Ld ) is the average inductance; Lx =
(CCM). 1/2(Lq − Ld ) is the inductance fluctuation; Ms = −(Lab +
Both the conventional onboard charger and the proposed Lbc + Lac )/3 is the average mutual inductance; and θe = P θr
single-phase integrated charger can be installed on EVs. The is the electrical angle between a-axis and q-axis.
selection of conventional onboard charging or high power level In the three-phase model, the electrical dynamic equations in
integrated charging depends on the power level of power supply terms of instantaneous stator phase voltages va , vb and vc can
available. be written as
⎧
⎪ ∂λa
⎪
⎪ va = Rs ia + (7)
III. ELECTRICAL MODEL OF AC PROPULSION MACHINE ⎪
⎪ ∂t
⎪
⎪
PMSMs are suitable candidates as PEV propulsion machines ⎨
∂λb
vb = Rs ib + (8)
owing to their superiorities in terms of high power density, ⎪
⎪ ∂t
⎪
⎪
high efficiency and light weight/volume [11]. They usually use ⎪
⎪
round-rotor structure instead of salient-pole for high efficient ⎪
⎩ vc = Rs ic + ∂λc (9)
operation at high rotating speeds (1500 rpm–14000 rpm or ∂t
even higher), which is the case for PEV applications [11]. In where, λa , λb , and λc are stator flux linkages, satisfying
this work, the integration theory is focused on a PMSM based ⎧
⎪ λ = Laa ia + Lab ib + Lac ic + λm a (10)
propulsion system. However, the proposed method and approach ⎨ a
is simply extendable to any other ac propulsion machine. λb = Lab ia + Lbb ib + Lbc ic + λm b (11)
⎪
⎩
The general mathematical model of a three-phase PMSM [26] λc = Lac ia + Lbc ib + Lcc ic + λm c (12)
is utilized based on the assumptions that: (1) stator windings’
where, λm a , λm b , and λm c are rotor permanent magnet flux
magneto-motive forces (MMF) are sinusoidally distributed; and
linkages.
(2) eddy current and hysteresis effects are neglected. The d-q
According to Kirchhoff’s Current Law (KCL), the instanta-
transformation (Park transformation) is utilized to convert the
neous phase-currents follow that
three-phase quantities to two-phase quantities that simplify the
analyses of three-phase electrical model [27]. Fig. 4(a) presents ia + ib + ic = 0 (13)
10902 IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. 66, NO. 12, DECEMBER 2017

Hence, applying (1)–(6) and (13) into (7)–(12), the instanta-

neous stator phase voltages va , vb and vc can be written as,
⎧
⎪ ∂ia
⎪
⎪ va = Rs ia + L1 (14)
⎪
⎪ ∂t
⎪
⎪
⎨
∂ib ∂ic
vb = Rs ib + L2 + L3 (15)
⎪
⎪ ∂t ∂t
⎪
⎪
⎪
⎪
⎪
⎩ ∂ic ∂ib
vc = Rs ic + L2 + L3 (16)
∂t ∂t
where, L1 , L2 and L3 are equivalent inductances in terms of Ld ,
Lq and Ms ,
⎧
⎪
⎪
5 1
⎪ L1 = Ld − Lq + Ms
⎪ (17)
⎪
⎪ 4 4
⎪
⎨
1 1
L = Ld + Lq + Ms (18)
⎪ 2
⎪ 2 2
⎪
⎪
⎪
⎪
⎪
⎩ L3 = 3 Ld − 3 Lq (19)
4 4
where, Ld and Lq are the d-axis and q-axis stator self-
inductances in the two-phase d-q equivalent circuit model of
a PMSM [27]; and Ms = −(Lab + Lbc + Lac )/3 is the av-
erage mutual inductance.
Therefore, the mutual effect appears between phase-b and
phase-c; however, it has no influence on phase-a. The relation-
ship among the phase voltages can be expressed as,
va + vb + vc = 0 (20)
The round-rotor motors are more suitable for high rotating
speeds as propulsion machines in PEVs. In the case of a round
rotor (Ld = Lq ), L1 = L2 = Laa + Ms , and L3 = 0. Thus,
(14)–(16) can be modified as,
⎧
⎪ ∂ia
⎪
⎪ va = Rs ia + Laa + Ms ) ∂t (21)
⎪
⎪
⎪
⎪
⎨
∂ib Fig. 5. Switching states (I)∼(IV) during battery charging.
vb = Rs ib + (Laa + Ms ) (22)
⎪
⎪ ∂t
⎪
⎪
⎪
⎪ the switching duty cycle (D), the steady-state operation is cat-
⎪
⎩ vc = Rs ic + (Laa + Ms ) ∂ic (23) egorized into two different cases: (1) 0 < D < 0.5, in which
∂t
Vo < 2Vac < 2Vo; and (2) 0.5 < D < 1, where Vo > 2Vac .
Equations (21)–(23) in a round-rotor structure indicate that in
The duty cycles mean the duty cycles of the lower side switches,
stationary condition each phase serves as a discrete inductor with
S4 and S6 , as shown in the Fig. 3. Here, CCM operation is se-
the equal inductances (Laa + Ms ). Here, the phase-currents
lected for the inductor currents. In fact, DCM operation can be
are functions of multiple independent variables. Therefore, par-
achieved for the inductor currents with the input current in CCM
tial derivatives are used to derive the equations. The same holds
operation. However, DCM operation for the inductor currents
for the instantaneous phase-currents.
brings larger current stresses for the inductors, the switches and
the diodes in the circuit. The current ratings of these compo-
IV. CHARGING OPERATION USING AC MACHINE nents need to be designed much higher to have enough margins
During charging, the switching operation is similar to that for DCM operation in comparison to the CCM operation at the
of a two-channel interleaved boost with two discrete inductors same power level. For high power-level applications, CCM op-
or one inversely coupled inductor. The interleaving legs (S4 eration is more suitable for the converter considering the current
and S6 ) operate with 180° phase difference in time domain. stresses of the components. In many electric vehicles (such as
However, in comparison to conventional interleaved boost con- Prius Hybrid, Camry hybrid, Fusion hybrid and Nissan Leaf
verters, the converter with 120° spatial out-of-phase distributed EV), there is a bidirectional dc-dc converter between the three-
windings (La , Lb , Lc ) of ac machine has different steady-state phase traction converter and the battery to step up the battery
equivalent inductances. The switching operation is divided into voltage when the battery voltage is lower than its rated volt-
four switching states (I)–(IV), as shown in Fig. 5. According to age due to low SoC (state of charge) [5], [29]–[30]. In battery
SHI et al.: SINGLE-PHASE INTEGRATED ONBOARD BATTERY CHARGER USING PROPULSION SYSTEM FOR PLUG-IN ELECTRIC VEHICLES 10903

mutual inductance. When 0 < D < 0.5, Vo < 2Vac < 2Vo ,
ia and −ib increase linearly; −ic decreases linearly. When
0.5 < D < 1, 2Vac < Vo , ia and −ic decrease linearly; −ib
increases linearly.
In State II, the transistor of the second leg, S6 , is turned on; and
the diode of the first leg, D3 , conducts. In this state: vaII = vaI ,
vbI I = vcI , vcI I = vbI . For a round rotor, the phase-currents can
be expressed as
⎧ II
⎪
⎪ ∂ia 2Vin − Vo
⎪
⎪ = (32)
⎪
⎪ ∂t 3 (Laa + Ms )
⎪
⎪
⎨
∂iII
Vin − 2Vo
− b = (33)
⎪
⎪ ∂t 3 (Laa + Ms )
⎪
⎪
⎪
⎪
⎪
⎪ ∂iI I Vin + Vo
⎩− c = (34)
∂t 3 (Laa + Ms )
Fig. 6. The current waveforms of the integrated charger with PMSM (round
rotor) during charging (a) 0 < D < 0.5; and (b) 0.5 < D < 1. State III only exists when 0 < D < 0.5. It occurs between
State I and State II. In this state, both transistors of two legs,
charging mode, this bidirectional dc-dc converter can be used S4 and S6 , are turned off; and diodes of two legs, D3 and D5 ,
as a buck converter to step down the DC-link voltage when the conduct.
battery voltage is lower than the peak input AC voltage. 
va − vb = Vin − Vo (35)
A. Switching States
va − vc = Vin − Vo (36)
When 0 < D < 0.5, the circuit operation has a periodical
switching sequence of (I)-(III)-(II)-(III)-(I). When 0.5 < D < 1, yielding,
the switching sequence changes to (I)-(IV)-(II)-(IV)-(I). Cur- ⎧
rents of stator inductors and semiconductors corresponding to ⎪ 2 (Vin − Vo )
⎪ III
⎨ va = (37)
different switching states are illustrated in Fig. 6. 3
In State I, the transistor of the first leg, S4 , is turned on; and ⎪
the diode of the second channel, D5 , conducts. ⎩ v III = v III = − Vin − Vo
⎪
(38)
b c
 3
va − vb = Vin (24)
For a round rotor (Lq = Ld ), ia , −ib and −ic decrease lin-
va − vc = Vin − Vo (25) early due to their equivalent discrete inductor. In this state, the
where, Vac and Vo are the input and output voltages of the slope of ia is twice of −ib and −ic :
interleaved boost converter. Using (20), (24) and (25), the stator ⎧ III
phase voltages in State I can be expressed as, ⎪
⎪ ∂ia 2 (Vin − Vo )
⎪
⎨ ∂t = 3 (Laa + Ms ) (39)
⎧
⎪ 2Vin − Vo
⎪
⎪
I
⎪ va =
⎪ 3
(26) ⎪
⎪
⎪ ∂iIII ∂iIII Vin − Vo
⎪
⎪ ⎩− b = − c = (40)
⎨ ∂t ∂t 3 (Laa + Ms )
Vin + Vo
vbI = − (27)
⎪
⎪ 3
⎪
⎪ State IV only exists when 0.5 < D < 1. It happens right
⎪
⎪ Vin − 2Vo
⎪ I
⎩ between Mode I and Mode II. In this state, both transistors of
vc = − (28)
3 two legs, S4 and S6 , are turned on; diodes of two legs, D3 and
According to (21)–(23) and neglecting stator resistance D5 , are reverse biased. The same voltage, Vac , appears across
(Rs = 0), the stator phase-currents can be represented as, the stator windings:
⎧ I 
⎪ ∂ia = 2Vin − Vo
⎪
⎪ (29) va − vb = Vin (41)
⎪
⎪ ∂t 3 (Laa + Ms )
⎪
⎪ va − vc = Vin (42)
⎪
⎨
∂iI Vin + Vo
− b = (30)
⎪
⎪ ∂t 3 (L aa + Ms ) Thereby, one can present the phase voltages as
⎪
⎪
⎪
⎪ ⎧
⎪ ∂iIc
⎪ Vin − 2Vo
⎩− = (31) ⎪
⎪
2Vin
∂t 3 (Laa + Ms ) ⎨ vaI V = (43)
3
Hence, in a round-rotor structure, the equivalent stator wind- ⎪
⎪
ing inductance is three times of sum of self-inductance and ⎩ vbI V = vcI V = − Vin (44)
3
10904 IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. 66, NO. 12, DECEMBER 2017

In the particular case of a round rotor, the phase-currents can Iin and the input current ripple |Δiin |, for 0 < D < 0.5, as
be written as ⎧
⎧ IV ⎪ 1
Iin ,m ax = Iin + |Δiin |
⎪ ∂i 2Vin ⎪
⎪
⎪
⎪ a = (45) ⎪
⎪ 2

⎨ ⎪
⎪ D (1 − 2D)
∂t 3 (Laa + Ms ) ⎪
⎪ 1
⎪
⎪ = + Vin (55)
⎪
⎪ ∂iIb V ∂iI V Vin ⎨ (1 − D)2 RL 6 (1 − D) (Laa + Ms ) fs
⎪
⎩− =− c = (46)
∂t ∂t 3 (Laa + Ms ) ⎪
⎪ 1
⎪
⎪Iin ,m in = Iin − |Δiin |
⎪
⎪ 2

B. Steady State Analyses ⎪

⎪
⎪
⎪ 1 D (1 − 2D)
⎪=
⎩ − Vin (56)
Since the rectified input current |iin | is equal to ia , based on (1 − D)2 RL 6 (1 − D) (Laa + Ms ) fs
(29) and (39), the input current ripple |Δiin | for 0 <D < 0.5 can
be calculated as and for 0.5 < D < 1, as
⎧
2Vin − Vo 2 (Vin − Vo ) ⎪ 1
|Δiin | = DT = − (0.5 − D) T ⎪
⎪Iin ,m ax = I in + 2 |Δiin |
3 (Laa + Ms ) 3 (Laa + Ms ) ⎪
⎪

⎪
⎪
(47) ⎪
⎪ 1 2D − 1
⎪
⎪ = + Vin (57)
For 0.5 < D < 1, the input current ripple can be found from ⎨ (1 − D)2 RL 6 (Laa + Ms ) fs
(29) and (45),
⎪
⎪ 1
2Vin − Vo ⎪
⎪ Iin ,m in = Iin − |Δiin |
⎪
⎪ 2

|Δiin | = − (1 − D) T ⎪
⎪
3 (Laa + Ms ) ⎪
⎪ 1 2D − 1
⎪
⎩ = − Vin (58)
2Vin (1 − D)2 RL 6 (Laa + Ms ) fs
= (D − 0.5) T (48)
3 (Laa + Ms ) For comparison, the maximum and minimum input current
Therefore, the steady-state output-to-input voltage gain Av equations of the traditional single-channel boost converter are
for 0 < D < 1 can be obtained as shown in (59)–(60).
Vo 1 ⎧
Av = = (49) ⎪
⎪
1
Vin 1 − D ⎪ Iin ,m ax = Iin + (ΔIin )
⎪
⎪
⎪ 2

⎪
⎪
which is equivalent to that of a conventional interleaved boost ⎪
⎪ 1 D
⎪
⎪ = + Vin (59)
converter. The input current ripple can be written as ⎨ (1 − D)2 RL Lfs
⎧
⎪ D (1 − 2D) Vin ⎪
⎪ 1
⎪
⎪ , ⎪
⎪ Iin ,m in = Iin − (ΔIin )
⎪
⎪ 1 − D 3 (L + Ms ) fs ⎪
⎪

⎪
⎪ aa ⎪
⎪
2
⎪
⎪ ⎪
⎪ 1 D
⎨ 0 < D < 0.5 (50) ⎪
⎪ = − Vin (60)
|Δiin | = ⎩ (1 − D)2 RL Lfs
⎪
⎪
⎪
⎪ (2D − 1) Vin
⎪
⎪ , It can be seen from (50) and (51) that the input current ripple
⎪
⎪ 3 (Laa + Ms ) fs
⎪
⎩ is suppressed by the equivalent stator inductance equal to the
0.5 < D < 1 (51)sum of self-inductance and average mutual inductance of three
On the other hand, the stator current ripple of phase-b and stator coils. Furthermore, the current cancellation effect of two
phase-c can be calculated as channels can reduce the current ripple. As shown in Fig. 7, the
⎧ effectiveness of ripple cancellation can be represented as the
⎪ D (2 − D) Vin
⎪
⎪ , normalized current ripple, γ(D), expressed as a function of duty
⎪
⎪ 1 − D 3 (Laa + Ms ) fs
⎪
⎪ cycle,
⎪
⎪
⎨ 0 < D < 0.5 (52) ⎧
⎪ 1 − 2D
|Δib,c | = ⎪
⎨ , 0 < D ≤ 0.5 (61)
⎪
⎪ |Δiin | 2 − D
⎪
⎪ (2 − D)
Vin γ (D) = =
⎪
⎪ , |Δib,c | ⎪
⎪
⎪
⎪
3 (Laa + Ms ) fs ⎩ 2D − 1 , 0.5 < D < 1
⎪
(62)
⎩ 2 − D
0.5 < D < 1 (53)
In steady state, considering an ideal converter, where the input For the conventional interleaved boost converter, the normal-
power is equal to the output power with an equivalent resistive ized current ripple, γ(D), in terms of duty cycle is represented
electrical load RL , we could have as,
⎧
Vo 2
⎪ 1 − 2D , 0 < D ≤ 0.5
Vin ⎪
Iin = = (54) ⎪ (63)
RL Vin (1 − D)2 RL |Δiin | ⎨ 1 − D
γ (D) = =
|ΔiL | ⎪
⎪
⎩ 2D − 1 , 0.5 < D < 1
Thus, the maximum value and minimum value of rectified in- ⎪
put current can be calculated based on the average input current (64)
D
SHI et al.: SINGLE-PHASE INTEGRATED ONBOARD BATTERY CHARGER USING PROPULSION SYSTEM FOR PLUG-IN ELECTRIC VEHICLES 10905

and S6 with 180 degree phase shift. The pulse widths of the
PWM signals are determined by the duty cycles.
Due to the symmetry of two-channel interleaved converter, the
steady-state currents in phase-b and phase-c windings are split
equally (Ib = Ic = 1/2|Iin |), even though their instantaneous
values might not be equal.
For a battery load, there are two charging modes: the constant
current (CC) charging and constant voltage (CV) charging. The
CC and CV charging are enabled by the controller of the pro-
posed integrated charger. To achieve the CC mode charging,
the reference battery charging current is sent to the outer loop
of the dual-loop controller, which regulates the output charging
current. To achieve the CV mode charging, the reference bat-
tery voltage is sent to the outer loop of the dual-loop controller,
which regulates the battery voltage.
Fig. 7. Effectiveness of the input current ripple cancellation for a boost con-
verter, a conventional interleaved boost converter, and the proposed integrated D. Electromagnetic Effect
propulsion machine charger.
In charging mode, given the phase-b and phase-c current
flowing directions, the vector-sum of the stator magnetic flux
due to phase-b and phase-c currents would be in the direction of
a-axis, according to the right-hand rule, which is aligned with
phase-a magnetic flux. Therefore, the overall vector-sum of the
three-phase stator magnetic flux is in the direction of a-axis.
As a result, the rotor is stationary with an electrical angle (θe )
equal to π/2 due to the electromagnetic force in the direction of
a-axis. Fig. 4 illustrates the rotor condition of a 3-phase, 2 pole-
pairs (P = 2) PMSM during charging, where the rotor angle
(θr ) is equal to the angle between d-axis and q-axis (π/2P ).
The theoretical analysis is applicable for the motor with any
number of pole-pairs. Here, a motor with 2 pole-pairs is utilized
as shown in Fig. 4.
Fig. 8. Schematic of the dual closed-loop control for the integrated charger. Based on the Park transformation, the stator currents in d-
The normalized input current ripple of the traditional single- axis (id ) and q-axis (iq ) can be expressed in terms of stator
channel boost converter is 1 since the input current ripple is the phase-currents (ia , ib , ic ) and electrical angle (θe ),
same as the inductor current ripple. ⎧   
⎪
⎪ 2 2π
In comparison to a conventional interleaved boost converter, ⎪
⎪ id = ia sinθe + ib sin θe −
⎪
⎪ 3   3
the proposed converter has smaller normalized current ripple ⎪
⎪
⎪
⎪ 2π
(or higher effective ripple cancellation) in the entire duty cycle ⎪
⎨ + ic sin θe + (65)
3
range. The full cancellation of the input current ripple occurs at   
D = 0.5. ⎪
⎪ 2 2π
⎪
⎪ i = i cosθ + i cos θ −
⎪
⎪ q
3
a e b e
3
⎪
⎪  
C. Control Strategy ⎪
⎪ 2π
⎪
⎩ + ic cos θe + (66)
The control strategy is developed with capability of achiev- 3
ing unity PF and less than 5% THD. The controller is com- The produced electromagnetic torque Te can be represented
posed of two closed loops: (1) the input current inner loop to as
shape the sinusoidal input line current; and (2) the output volt-  
3 1
age/current outer loop to regulate the output voltage/current, as Te = λm ia cosθe + (Lq − Ld ) ia 2 sin (2θe ) (67)
2 2
shown in Fig. 8. The inner loop shapes the steady-state phase-b
and phase-c currents (Ib and Ic ) through a phase locked loop When the electrical angle is equal to π/2, the electromagnetic
(PLL) to follow the trajectory of the line voltage. The outer torque, generated by the magnetic flux of the permanent magnets
loop regulates the magnitude of the phase-currents correspond- and the stator phase-currents, is zero based on (67).
ing to output voltage/current. The PWM block is implemented Before the electric vehicles begin to charge, the initial elec-
in DSP TMS320F28335 from Texas Instrument using the in- trical angle of the propulsion machine is set at π/2 radian by the
ternal ePWM modules, which compares the signal from the PI motor-driving control applied in the propulsion mode. Then,
controller with the produced triangle waveforms to generate two the gear of propulsion system is locked, and the charging mode
PWM signals. These two PWM signals are sent to switches S4 begins. Thus, the inductances of phase-B and phase-C motor
10906 IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. 66, NO. 12, DECEMBER 2017

TABLE II
TEST BED PERFORMANCE

Parameters Symbol Value Unit

Input ac voltage V in , rm s 90∼240 V

Input ac current Iin , rm s 13.6 A
Input frequency fa c 60 Hz
Switching frequency fs 15 kHz
Output dc voltage Vo 340∼420 V
Equivalent resistive load RL 58.8 Ω
Maximum output power Po u t 3 kW
Power factor PF 0.98
Input current total harmonic distortion THD 3.96 %
Maximum efficiency Ef f 93.1 %

Fig. 9. Proposed integrated onboard charger using PMSM.

TABLE I
INTEGRATED CHARGER COMPONENT PARAMETERS

Device Part # Value # of device

PMSM AKM62M 3.6 kW/220 1

V,rms/3-phase/2
pole-pairs
Diode bridge GBPC3506W 35 A/420 V,rms 1
rectifier
IGBT in traction FGL60N100 NA 6
converter
DC capacitor LGW2W561 3.3 mF 6
PFC controller TMS320F28335 NA 1
Current sensor LTS25-NP NA 3
Voltage sensor LV20-P NA 2 Fig. 10. Experimental waveforms of input voltage (V in ), phase-b current (Ib )
and phase-c current (Ic ) of the proposed integrated charger. Y-axis (from top to
bottom): Ch2 = V in 250 V/div; Ch4 = Ic 5 A/div; Ch1 = Ib 5 A/div; X-
axis: time 10 ms/div.

windings can be kept constant and equal to each other every

time the EV starts to operate in charging mode. In battery charg- Although the machine and the traction converter are designed
ing mode, the current in phase-b has the same phase-angle as from a drive point of view, the parameters are naturally ap-
the current in phase-c. The current in phase-a is the sum of the plicable for battery charging mode. Each phase winding of
currents in phase-b and phase-c. The total generated torque is the PMSM electric machine has 1.2-mH self-inductance and
kept as zero during charging. 0.5-mH mutual-inductance between each two phase windings.
These inductance values are typical values of the PMSM elec-
tric machines which are widely used in electric vehicles. For an
V. EXPERIMENTAL RESULTS
interleaved boost converter operating at 15 kHz, the inductance
To verify the proposed theory, a 3-kW test bed using a values are appropriate for the converter to operate in CCM op-
3.6-kW propulsion system is developed. The circuit is shown in eration. As for the capacitor Cdc , the capacitance, 3.3-mF, is
Fig. 9. A 3.6-kW, 220-Vrm s , 3-phase, 2 pole-pairs round-rotor selected based on the voltage ripple requirement for the propul-
PMSM is utilized as a three-winding coupled inductor in charg- sion mode. The proposed integrated charger utilizes the DC-link
ing mode. The propulsion system utilizes a 3.6-kW three-phase capacitor of the traction system without any add-on large capaci-
traction converter module with six IGBTs. The Chroma 63212 tor. The DC-Link capacitor is designed based on the requirement
DC electronic load is utilized to test the circuit. The CC mode of the voltage ripple in the inversion operation mode. Usually,
and CV mode of this electronic load can emulate the CC-CV a large capacitance is selected for the DC-link capacitor. This
charging process very well. Thus, the electronic load can be used capacitance is large enough to balance the instantaneous power
to validate the feasibility of the proposed integration scheme. difference when energy is transferred from the DC side to the
The output voltage is regulated from 340 V to 420 V, based on AC side within a voltage ripple limit on the DC-link capaci-
the rated voltage of EV battery. The IGBT-based traction con- tor. In battery charging mode, the energy is transferred in the
verter is used to construct the interleaved boost charger with the opposite direction, and the voltage ripple can meet the same
switching frequency as 15 kHz during charging. The switching requirement as set in the propulsion mode. Thus, capacitor Cdc
frequency is selected based on the specifications of the IGBT in can be directly used in the battery charging mode.
the traction converter. The component parameters of integrated Fig. 10 shows the waveforms of the input line voltage, phase-
charger are listed in Table I. The overall performance of the test b current and phase-c current when the input current is 10 A and
bed is presented in Table II. the output power is 2 kW. The rectified input current is evenly
SHI et al.: SINGLE-PHASE INTEGRATED ONBOARD BATTERY CHARGER USING PROPULSION SYSTEM FOR PLUG-IN ELECTRIC VEHICLES 10907

Fig. 13. Conversion efficiency of the integrated charger test bed at different
input voltage (V in = 120 Vr m s and V in = 240 Vr m s ) and different output
powers (P o u t = 600 W ∼ 3 kW).

Fig. 11. Experimental waveforms of phase-b current (Ib ), phase-c current

(Ic ), gate voltage of S 4 (V s 4, g a te ) and gate voltage of S 6 (V s 6, g a te ) at
D > 0.5. Y-axis (from top to bottom): Ch4 = Ic 2 A/div; Ch2 = Ib 2 A/div;
Ch3 = V s 4, g a te 25 V/div; Ch1 = V s 6, g a te 25 V/div, X-axis: time 40 μs/div.

Fig. 14. Loss breakdown at input voltage (V in = 240 V, r m s ), output voltage

(V o u t = 420 V) and output power (P o u t = 3 kW).

Fig. 12. Experimental waveforms of the input current (Iin ), in-

put voltage (V in ), and output voltage (V d c ) at V in , rm s = 240 V,
Iin , rm s = 13.6 V, V d c = 420 V, P o u t = 3 kW; Y-axis (from top to bot-
tom): Ch3 = V d c 250 V/div; Ch2 = V in 250 V/div; Ch4 = Iin 20 A/div;
X-axis: time 20 ms/div.

shared between the phase-b and the phase-c windings. Fig. 11

shows waveforms of phase-b current, phase-c current and switch
gating signals when the duty cycle is higher than 0.5. The wave-
forms in each switching state confirm the analyses in Section IV.
The current ripples in phase-b and phase-c are high frequency Fig. 15. Experimental waveforms of the transient process (30% load to
70% load), including the input current (Iin ), input voltage (V in ), and
ripple. Due to the large inertia of the rotor the high frequency output voltage (V d c ) at V in , rm s = 240 V, Iin , rm s = 4.3 V − Iin , rm s =
ripples cause no vibrating around a-axis. Furthermore, the aver- 8.5 V, V d c = 400 V; Y-axis (from top to bottom): Ch2 = V in 500 V/div;
age electromagnetic torque exerted on torque in one switching Ch3 = V d c 250 V/div; Ch4 = Iin 20 A/div; X-axis: time 10 ms/div.
period is zero because the current ripples in phase-b and phase-
c are the same with 180 degree phase-shift. The experimental The efficiency curves for 120 Vrm s input and 240 Vrm s input
results have also proved that there is no rotor vibration around at different loading conditions are illustrated in Fig. 13. The effi-
a-axis in battery charging operation. In Figs. 10 and 11, there is ciencies are calculated by measuring the input current, the input
a small amount of unbalanced current between the phase-b and voltage, output current and output voltage at different power lev-
phase-c. In the test, the unbalanced current amount is less than els for Vin = 120 V,rm s and Vin = 240 V,rm s . The efficiency
0.8 A, resulting in very limited uneven thermal dissipation of the curves are obtained based on the calculated efficiencies. By
two interleaved boost channels. This small unbalanced current measuring the RMS values of the input voltage, input current,
does not affect the feasibility of the proposed integrated charger, output current, and output voltage for each component in the
and current balancing techniques can be applied to reduce the circuit, the power losses of each component can be measured.
unbalancing currents [31], [32]. Fig. 14 shows the loss breakdown of the proposed integrated
Fig. 12 shows the waveforms of the input line current, input charger with input voltage Vin = 240 V,rm s , output voltage
line voltage and output voltage at full load. The input voltage Vout = 420 V and output power Pout = 3 kW.
is 240 V,rm s . The output voltage is regulated at 420 V. The Fig. 15 shows the waveforms during the transient process.
efficiency reaches 93.1% at 3 kW output power. The DC-link voltage is kept as 400 V during the transient
10908 IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. 66, NO. 12, DECEMBER 2017

TABLE III
COMPARISON TO OTHER SINGLE-PHASE-BATTERY CHARGERS

Proposed Integrated Charger with Charger with

Charger single-channel interleaved boost
boost PFC [33] PFC [33]

Components Diode Bridge Two stage system Two stage system

(PFC + LLC) (PFC + LLC)
Size Very Small Large Large
Efficiency 93.1% 94.5% 95.0%
Cost Very low Medium High
EMI / noise Best Poor Fair
Fig. 17. CC-CV charging profile.

Fig. 18. Experimental waveforms of the startup process, including

the input current (Iin), input voltage (Vin), and output voltage (Vdc);
Ch2 = Vin 250 V/div; Ch3 = Vdc 500 V/div; Ch4 = Iin 20 A/div; X-
Fig. 16. Harmonics orders at full load compared against the EN61000-3-2 axis: time 20 ms/div.
standard.

process. The experimental results show that the dual-loop con-

troller perfectly regulates the DC-link voltage during the load
transient.
Due to the power loss on the propulsion machine, the ef-
ficiency is slightly lower than the other single-phase-battery
chargers. Instead of utilizing the propulsion machine, the tradi-
tional single-phase-chargers use optimally designed inductors,
which cause less power loss. Table III shows the comparison be-
tween the proposed integrated charger with other conventional
Fig. 19. EMI spectrum over entire conducted EMI range with the CISPR class
single-phase-battery chargers, i.e., the two-stage battery charger B standard.
with single-channel boost PFC converter and the two-stage bat-
tery charger with interleaved boost PFC converter. In terms of
The CC-CV charging plot for a typical 24 kWh battery pack
cost, the integrated charger costs much less than other battery
is shown in Fig. 17.
chargers since only a diode-bridge is required for the proposed
For the startup of the converter, different startup strategies
integrated charger.
provide different transient behaviors. The reference value for the
Based on the analysis of the input current ripples and Fig. 7,
controller gradually increases at a slow rate during the startup
the proposed integrated charger has the minimum input current
process to limit the transient input current. Fig. 18 shows the
ripple in comparison to the other traditional battery chargers.
transient waveforms at the time of startup.
Furthermore, the inductances of the propulsion machines are
In comparison to the CISPR class B standard, the EMI spec-
relatively higher than the inductances of the traditional battery
trum over entire conducted EMI range is shown in Fig. 19. The
chargers [33], resulting even less input current ripple. Thus, the
EMI of the proposed integrated charger meets the requirement
proposed integrated charger has the minimum EMI noise and
of the EMI standard.
the minimum size of the EMI filter. The EMI comparison with
the traditional battery chargers is presented in Table III.
VI. CONCLUSIONS
Fig. 16 compares the harmonics orders between the proposed
integrated charger and the EN61000-3-2 Class D Limits (A) In this paper, an integrated onboard battery charger based on
based on the experimental results at full load. It shows that the a three-phase-ac propulsion system is proposed for PEVs. The
THD of the proposed integrated charger is compliant with the charging topology utilizes an ac propulsion machine and its trac-
EN61000-3-2 standard. tion converter as a two-channel interleaved boost converter for
SHI et al.: SINGLE-PHASE INTEGRATED ONBOARD BATTERY CHARGER USING PROPULSION SYSTEM FOR PLUG-IN ELECTRIC VEHICLES 10909

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hicles,” IEEE Trans. Veh. Technol., vol. 58, no. 8, pp. 3970–3980, trical engineering with distinguished honor from
Oct. 2009. Wuhan University, Wuhan, China, in 2013. He is cur-
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pp. 1–6. Austin, TX, USA, in Summer 2015. His research
[16] S. Dusmez and A. Khaligh, “A novel low cost integrated onboard charger interests include power electronics, plug-in electric
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[17] S. D. Mohamed, A. E. Ahmed, S. A. Ayman, M. M. Ahmed, and ing, and power management of hybrid energy storage
A. Shehab,“A nine-switch-converter-based integrated motor drive and systems. He received the 2016 Harry K. Wells Energy Research Fellowship
battery charger system for EVs using symmetrical six-phase ma- from the University of Maryland Energy Research Center.
10910 IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. 66, NO. 12, DECEMBER 2017

Yichao Tang (S’12) received the B.S. degree from Alireza Khaligh (S’04–M’06–SM’09) is currently
Shanghai Jiaotong University, Shanghai, China, in an Associate Professor in the Department of Electri-
2009, the M.S. degree from Illinois Institute of Tech- cal and Computer Engineering (ECE) and the Insti-
nology, Chicago, IL, USA, in 2011, and the Ph.D. tute for Systems Research, University of Maryland,
degree in electrical engineering from the University College Park (UMD), MD, USA. His major research
of Maryland, College Park, MD, USA. interests include modeling, analysis, design, and con-
His research interests include modeling, analysis, trol of power electronic converters for transportation
design and control of ac–dc, dc–dc, and dc–ac power electrification, renewable energies, energy harvest-
electronic converters, energy harvesting from envi- ing, and microrobotics. He is an author/co-author of
ronmental sources, renewable energies, power auton- more than 160 journal and conference papers. He is
omy of mobile microrobots, electric chargers for EV the Area Editor for “Vehicular Electronics and Sys-
and PEV, as well as power conditioning systems for more electric aircraft and tems Area” of the IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY. He is
shipboard power systems. currently an Associate Editor of the IEEE TRANSACTIONS ON POWER ELECTRON-
ICS, an Associate Editor of the IEEE JOURNAL OF EMERGING AND SELECTED
TOPICS IN POWER ELECTRONICS, and an Associate Editor of the IEEE TRANS-
ACTIONS ON TRANSPORTATION ELECTRIFICATION.
Dr. Khaligh received the various awards and recognitions including 2017
Outstanding Young Alumnus Award from Illinois Institute of Technology, the
2016 E. Robert Kent Junior Faculty Teaching Award from Clark School of En-
gineering at UMD, the 2016 Junior Faculty Outstanding Research Award from
Clark School of Engineering at UMD, the 2015 Junior Faculty Fellowship from
the Institute for Systems Research at UMD, 2013 George Corcoran Memorial
Award from the ECE Department at UMD, three Best Vehicular Electronics
Awards from the IEEE Vehicular Electronics Society (VTS), and 2010 Ralph
R. Teetor Educational Award from Society of Automotive Engineers. He was
the General Chair of the 2016 IEEE Applied Power Electronic Conference and
Expo, Long Beach, CA, USA, the General Chair of the 2013 IEEE Transporta-
tion Electrification Conference and Expo, Dearborn, MI, USA, and the Program
Chair of the 2011 IEEE Vehicle Power and Propulsion Conference, Chicago, IL,
USA. He is a Distinguished Lecturer of the IEEE Vehicular Technology Society
and also a Distinguished Lecturer of the the IEEE Industry Applications Society.