Stereo 60W, Mono 120W Audio Power Amplifier with Mute

LM4780 Overture™ Audio Power Amplifier Series

July 2003

LM4780 Overture™ Audio Power Amplifier Series

Stereo 60W, Mono 120W Audio Power Amplifier with
Mute
General Description Key Specifications
The LM4780 is a stereo audio amplifier capable of typically n Output Power/Channel with 0.5% THD+N, 1kHz into
delivering 60W per channel of continuous average output 8Ω 60W (typ)
power into an 8Ω load with less than 0.5% THD+N from n THD+N at 2 x 30W into 8Ω (20Hz - 20kHz) 0.03% (typ)
20Hz - 20kHz. n THD+N at 2 x 30W into 6Ω (20Hz - 20kHz) 0.05% (typ)
The LM4780 is fully protected utilizing National’s Self Peak n THD+N at 2 x 30W into 4Ω (20Hz - 20kHz) 0.07% (typ)
Instantaneous Temperature (˚Ke) (SPiKeTM) protection cir- n Mute Attenuation 110dB (typ)
cuitry. SPiKe provides a dynamically optimized Safe Operat- n PSRR 85dB (min)
ing Area (SOA). SPiKe protection completely safeguards the n Slew Rate 19V/µs (typ)
LM4780’s outputs against over-voltage, under-voltage, over-
loads, shorts to the supplies or GND, thermal runaway and
instantaneous temperature peaks. The advanced protection Features
features of the LM4780 places it in a class above discrete n SPiKe Protection
and hybrid amplifiers. n Low external component count
Each amplifier of the LM4780 has an independent smooth n Quiet fade-in/out mute mode
transition fade-in/out mute. n Wide supply range: 20V - 84V
The LM4780 can easily be configured for bridge or parallel n Signal-to-Noise Ratio ≥ 97dB (ref. to PO = 1W)
operation for 120W mono solutions.
Applications
n Audio amplifier for component stereo
n Audio amplifier for compact stereo
n Audio amplifier for self-powered speakers
n Audio amplifier for high-end and HD TVs

Typical Application

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FIGURE 1. Typical Audio Amplifier Application Circuit

SPiKe™ Protection and Overture™ are trademarks of National Semiconductor Corporation.

© 2003 National Semiconductor Corporation DS200586 www.national.com

LM4780
Connection Diagrams
Plastic Package (Note 14)

200586D6
Top View
Order Number LM4780TA
See NS Package Number TA27A

TO-220 Top Marking

200586A2
Top View
U - Wafer Fab Code
Z - Assemble Plant Code
XY - Date Code
TT - Die Run Traceability
L4780TA - LM4780TA

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 LM4780
Absolute Maximum Ratings (Notes 1, Soldering Information
2) TA Package (10 seconds) 260˚C
If Military/Aerospace specified devices are required, Storage Temperature -40˚C to +150˚C
please contact the National Semiconductor Sales Office/ Thermal Resistance
Distributors for availability and specifications. θJA 30˚C/W
+ -
Supply Voltage |V | + |V | θJC 0.8˚C/W
(No Signal) 94V
Supply Voltage |V+| + |V-| Operating Ratings (Notes 1, 2)
(Input Signal) 84V
Temperature Range
Common Mode Input Voltage (V+ or V-) and
TMIN ≤ TA ≤ TMAX −20˚C ≤ TA ≤ +85˚C
|V+| + |V-| ≤ 80V
Supply Voltage |V+| + |V-| 20V ≤ VTOTAL ≤ 84V
Differential Input Voltage (Note 13) 60V
Output Current Internally Limited
Power Dissipation (Note 3) 125W Note: Operation is guaranteed up to 84V; however, distor-
tion may be introduced from SPiKe protection circuitry if
ESD Susceptability (Note 4) 3.0kV
proper thermal considerations are not taken into account.
ESD Susceptability (Note 5) 200V Refer to the Thermal Considerations section for more
Junction Temperature (TJMAX) (Note 9) 150˚C information.

Electrical Characteristics (Notes 1, 2)

The following specifications apply for V+ = +35V, V- = −35V, IMUTE = -1mA and RL = 8Ω unless otherwise specified. Limits ap-
ply for TA = 25˚C.
Symbol Parameter Conditions LM4780 Units
Typical Limit (Limits)
(Note 6) (Notes 7, 8)
+ - -
|V | + |V | Power Supply Voltage (Note GND − V ≥ 9V 18 20 V (min)
10) 84 V (max)
AM Mute Attenuation IMUTE = 0mA 110 80 dB (min)
THD+N = 0.5% (max)
f = 1kHz; f = 20kHz
PO Output Power (RMS) |V+| = |V-| = 25V, RL = 4Ω 55 50 W (min)
|V+| = |V-| = 30V, RL = 6Ω 55 50 W (min)
|V+| = |V-| = 35V, RL = 8Ω 60 50 W (min)
PO = 30W, f = 20Hz - 20kHz
AV = 26dB
Total Harmonic Distortion +
THD+N |V+| = |V-| = 25V, RL = 4Ω 0.07 %
Noise
|V+| = |V-| = 30V, RL = 6Ω 0.05 %
|V+| = |V-| = 35V, RL = 8Ω 0.03 %
PO = 10W, f = 1kHz 70 dB
Xtalk Channel Separation (Note 11)
PO = 10W, f = 10kHz 72 dB
SR Slew Rate VIN = 2.0VP-P, tRISE = 2ns 19 8 V/µs (min)
IDD Total Quiescent Power VCM = 0V, 110 170 mA (max)
Supply Current VO = 0V, IO = 0A
VOS Input Offset Voltage VCM = 0V, IO = 0mA 1 10 mV (max)
IB Input Bias Current VCM = 0V, IO = 0mA 0.2 1 µA (max)
+ -
IO Output Current Limit |V | = |V | = 20V, tON = 10ms 11.5 7 A (min)
VOD Output Dropout Voltage |V+ - VO|, V+ = 28V, IO = +100mA 1.6 2.0 V (max)
(Note 12) |V- - VO|, V- = -28V, IO = -100mA 2.5 3.0 V (max)
V+ = 40V to 20V, V- = -40V,
120 85 dB (min)
Power Supply Rejection Ratio VCM = 0V, IO = 0mA
PSRR
(Note 15) V+ = 40V, V- = -40V to -20V,
105 85 dB (min)
VCM = 0V, IO = 0mA
CMRR Common Mode Rejection Ratio V+ = 60V to 20V, V- = -20V to -60V, 110 85 dB (min)
(Note 15) VCM = 20V to -20V, IO = 0mA

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LM4780
Electrical Characteristics (Notes 1, 2) (Continued)
The following specifications apply for V+ = +35V, V- = −35V, IMUTE = -1mA and RL = 8Ω unless otherwise specified. Limits ap-
ply for TA = 25˚C.
Symbol Parameter Conditions LM4780 Units
Typical Limit (Limits)
(Note 6) (Notes 7, 8)
AVOL Open Loop Voltage Gain RL = 2kΩ, ∆VO = 40V 115 90 dB (min)
GBWP Gain Bandwidth Product fIN = 100kHz, VIN = 50mVRMS 8 2 MHz (min)
eIN Input Noise IHF-A-Weighting Filter, 2.0 10 µV (max)
RIN = 600Ω (Input Referred)
PO = 1WRMS; A-Weighted Filter
97 dB
fIN = 1kHz, RS = 25Ω
SNR Signal-to-Noise Ratio
PO = 50WRMS; A-Weighted Filter
114 dB
fIN = 1kHz, RS = 25Ω
60Hz, 7kHz, 4:1 (SMPTE) 0.004 %
IMD Intermodulation Distortion
60Hz, 7kHz, 1:1 (SMPTE) 0.009 %

Note 1: All voltages are measured with respect to the ground pins, unless otherwise specified.
Note 2: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is
functional, but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical specifications under particular test conditions which
guarantee specific performance limits. This assumes that the device is within the Operating Ratings. Specifications are not guaranteed for parameters where no limit
is given; however, the typical value is a good indication of device performance.
Note 3: The maximum power dissipation must be de-rated at elevated temperatures and is dictated by TJMAX, θJC, and the ambient temperature TA. The maximum
allowable power dissipation is PDMAX = (TJMAX -TA)/θJC or the number given in the Absolute Maximum Ratings, whichever is lower. For the LM4780, TJMAX = 150˚C
and the typical θJC is 0.8˚C/W. Refer to the Thermal Considerations section for more information.
Note 4: Human body model, 100pF discharged through a 1.5kΩ resistor.
Note 5: Machine Model: a 220pF - 240pF discharged through all pins.
Note 6: Typical specifications are measured at 25˚C and represent the parametric norm.
Note 7: Tested limits are guaranteed to National’s AOQL (Average Outgoing Quality Level).
Note 8: Datasheet min/max specification limits are guaranteed by design, test, or statistical analysis.
Note 9: The maximum operating junction temperature is 150˚C. However, the instantaneous Safe Operating Area temperature is 250˚C.
Note 10: V- must have at least - 9V at its pin with reference to GND in order for the under-voltage protection circuitry to be disabled. In addition, the voltage
differential between V+ and V- must be greater than 14V.
Note 11: Cross talk performance was measured using the demo board shown in the datasheet. PCB layout will affect cross talk. It is recommended that input and
output traces be separated by as much distance as possible. Return ground traces from outputs should be independent back to a single ground point and use as
wide of traces as possible.
Note 12: The output dropout voltage is defined as the supply voltage minus the clipping voltage. Refer to the Clipping Voltage vs. Supply Voltage graph in the
Typical Performance Characteristics section.
Note 13: The Differential Input Voltage Absolute Maximum Rating is based on supply voltages V+ = 40V and V- = - 40V.
Note 14: The TA27A is a non-isolated package. The package’s metal back, and any heat sink to which it is mounted are connected to the V- potential when using
only thermal compound. If a mica washer is used in addition to thermal compound, θCS (case to sink) is increased, but the heat sink will be electrically isolated from
V-.
Note 15: DC electrical test.
Note 16: CCIR/ARM: A Practical Noise Measurement Method; by Ray Dolby, David Robinson and Kenneth Gundry, AES Preprint No. 1353 (F-3).

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 LM4780
Bridged Amplifier Application Circuit

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FIGURE 2. Bridged Amplifier Application Circuit

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LM4780
Parallel Amplifier Application Circuit

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FIGURE 3. Parallel Amplifier Application Circuit

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 LM4780
Single Supply Application Circuit

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FIGURE 4. Single Supply Amplifier Application Circuit

Note: *Optional components dependent upon specific design requirements.

Auxiliary Amplifier Application Circuit

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FIGURE 5. Special Audio Amplifier Application Circuit

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LM4780
External Components Description
(Figures 1-5)

Components Functional Description

1 RB Prevents current from entering the amplifier’s non-inverting input. This current may pass through to the load
during system power down, because of the amplifier’s low input impedance when the undervoltage circuitry
is off. This phenomenon occurs when the V+ and V- supply voltages are below 1.5V.
2 Ri Inverting input resistance. Along with Rf, sets AC gain.
3 Rf Feedback resistance. Along with Ri, sets AC gain.
4 Rf2 Feedback resistance. Works with Cf and Rf creating a lowpass filter that lowers AC gain at high
(Note 17) frequencies. The -3dB point of the pole occurs when: (Rf - Ri)/2 = Rf // [1/(2πfcCf) + Rf2] for the
Non-Inverting configuration shown in Figure 5.
5 Cf Compensation capacitor. Works with Rf and Rf2 to reduce AC gain at higher frequencies.
(Note 17)
6 CC Compensation capacitor. Reduces the gain at higher frequencies to avoid quasi-saturation oscillations of the
(Note 17) output transistor. Also suppresses external electromagnetic switching noise created from fluorescent lamps.
7 Ci Feedback capacitor which ensures unity gain at DC. Along with Ri also creates a highpass filter at fc =
(Note 17) 1/(2πRiCi).
8 CS Provides power supply filtering and bypassing. Refer to the Supply Bypassing application section for proper
placement and selection of bypass capacitors.
9 RV Acts as a volume control by setting the input voltage level.
(Note 17)
10 RIN Sets the amplifier’s input terminals DC bias point when CIN is present in the circuit. Also works with CIN to
(Note 17) create a highpass filter at fC = 1/(2πRINCIN). If the value of RIN is too large, oscillations may be observed on
the outputs when the inputs are floating. Recommended values are 10kΩ to 47kΩ. Refer to Figure 5.
11 CIN Input capacitor. Prevents the input signal’s DC offsets from being passed onto the amplifier’s inputs.
(Note 17)
12 RSN Works with CSN to stabilize the output stage by creating a pole that reduces high frequency instabilities.
(Note 17)
13 CSN Works with RSN to stabilize the output stage by creating a pole that reduces high frequency instabilities. The
(Note 17) pole is set at fC = 1/(2πRSNCSN). Refer to Figure 5.
14 L (Note 17) Provides high impedance at high frequencies so that R may decouple a highly capacitive load and reduce
the Q of the series resonant circuit. Also provides a low impedance at low frequencies to short out R and
15 R (Note 17)
pass audio signals to the load. Refer to Figure 5.
16 RA Provides DC voltage biasing for the transistor Q1 in single supply operation.
17 CA Provides bias filtering for single supply operation.
18 RINP Limits the voltage difference between the amplifier’s inputs for single supply operation. Refer to the Clicks
(Note 17) and Pops application section for a more detailed explanation of the function of RINP.
19 RBI Provides input bias current for single supply operation. Refer to the Clicks and Pops application section for
a more detailed explanation of the function of RBI.
20 RE Establishes a fixed DC current for the transistor Q1 in single supply operation. This resistor stabilizes the
half-supply point along with CA.
21 RM Mute resistance set up to allow 0.5mA to be drawn from each MUTE pin to turn the muting function off.
→ RM is calculated using: RM ≤ (|VEE| − 2.6V)/l where l ≥ 0.5mA. Refer to the Mute Attenuation vs Mute
Current curves in the Typical Performance Characteristics section.
22 CM Mute capacitance set up to create a large time constant for turn-on and turn-off muting.
23 S1 Mute switch. When open or switched to GND, the amplifier will be in mute mode.
24 ROUT Reduces current flow between outputs that are caused by Gain or DC offset differences between the
amplifiers.

Note 17: Optional components dependent upon specific design requirements.

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 LM4780
OPTIONAL EXTERNAL tion capacitor, Cf. These two components are low imped-
ances at certain frequencies. They may couple signals from
COMPONENT INTERACTION the input to the output. Please take careful note of basic
The optional external components have specific desired amplifier component functionality when selecting the value of
functions. Their values are chosen to reduce the bandwidth these components and their placement on a printed circuit
and eliminate unwanted high frequency oscillation. They board (PCB).
may, however, cause certain undesirable effects when they The optional external components shown in Figure 4 and
interact. Interaction may occur when the components pro- Figure 5, and described above, are applicable in both single
duce reactions that are nearly equal to one another. One and split supply voltage configurations.
example is the coupling capacitor, CC, and the compensa-

Typical Performance Characteristics

THD+N vs Frequency THD+N vs Frequency
± 25V, POUT = 1W & 30W/Channel ± 30V, POUT = 1W & 30W/Channel
RL= 4Ω, 80kHz BW RL= 6Ω, 80kHz BW

200586E3 200586E4

THD+N vs Frequency THD+N vs Output Power/Channel

± 35V, POUT = 1W & 30W/Channel ± 25V, RL= 4Ω, 80kHz BW
RL= 8Ω, 80kHz BW

200586E5 200586E8

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LM4780
Typical Performance Characteristics (Continued)

THD+N vs Output Power/Channel THD+N vs Output Power/Channel

± 30V, RL= 6Ω, 80kHz BW ± 35V, RL= 8Ω, 80kHz BW

200586E9 200586F0

Output Power/Channel Output Power/Channel

vs Supply Voltage vs Supply Voltage
f = 1kHz, RL = 4Ω, 80kHz BW f = 1kHz, RL = 6Ω, 80kHzBW

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Output Power/Channel Total Power Dissipation

vs Supply Voltage vs Output Power/Channel
f = 1kHz, RL = 8Ω, 80kHz BW 1% THD (max), RL = 4Ω, 80kHz BW

20058623 200586A6

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 LM4780
Typical Performance Characteristics (Continued)

Total Power Dissipation Total Power Dissipation

vs Output Power/Channel vs Output Power/Channel
1% THD (max), RL = 6Ω, 80kHz BW 1% THD (max), RL = 8Ω, 80kHz BW

200586A7 200586A8

Crosstalk vs Frequency Crosstalk vs Frequency

± 25V, POUT = 10W ± 35V, POUT = 10W
RL = 4Ω, 80kHz BW RL = 8Ω, 80kHz BW

200586C5 200586A5

Mute Attenuation Supply Current

vs Mute Pin Current vs Supply Voltage
POUT = 10W/Channel

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LM4780
Typical Performance Characteristics (Continued)

Power Supply
Large Signal Response Rejection Ratio

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200586C8

Common Mode Open Loop

Rejection Ratio Frequency Response

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200586C9

Clipping Voltage Clipping Voltage

vs Supply Voltage vs Supply Voltage

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 LM4780
Typical Performance Characteristics (Continued)

THD+N vs Frequency THD+N vs Frequency

± 25V, POUT = 1W & 50W ± 35V, POUT = 1W & 50W
Bridge Mode (Note 18), RL = 8Ω, 80kHz BW Parallel Mode (Note 19), RL = 4Ω, 80kHz BW

200586E6 200586E7

THD+N vs Output Power THD+N vs Output Power

± 25V, Bridge Mode (Note 18) ± 35V, Parallel Mode (Note 19)
RL = 8Ω, 80kHz BW RL = 4Ω, 80kHz BW

200586F1 200586F2

Output Power vs Output Power vs

Supply Voltage, Bridge Mode (Note 18) Supply Voltage, Parallel Mode (Note 19)
f = 1kHz, RL = 8Ω, 80kHz BW f = 1kHz, RL = 4Ω, 80kHz BW

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LM4780
Typical Performance Characteristics (Continued)

SPiKe
Safe Area Protection Response

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200586D3

Frequency Response of Demo Board

POUT = 10W/Channel = 0dB
RIN = 47kΩ, RL = 8Ω, No Filters

200586E2

Note 18: Bridge mode graphs were taken using the demo board and invert-
ing the signal to the channel B input.
Note 19: Parallel mode graphs were taken using the demo board and con-
necting each output through a 0.1Ω/3W resistor to the load.

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 LM4780
Application Information OVER-VOLTAGE PROTECTION
The LM4780 contains over-voltage protection circuitry that
MUTE MODE limits the output current while also providing voltage clamp-
ing. The clamp does not, however, use internal clamping
The muting function allows the user to mute the amplifier.
diodes. The clamping effect is quite the same because the
This can be accomplished as shown in the Typical Applica-
output transistors are designed to work alternately by sinking
tion Circuit. The resistor RM is chosen with reference to the
large current spikes.
negative supply voltage and is used in conjunction with a
switch. The switch, when opened or switched to GND, cuts
SPiKe PROTECTION
off the current flow from the MUTE pins to −VEE, thus placing
the LM4780 into mute mode. Refer to the Mute Attenuation The LM4780 is protected from instantaneous peak-
vs Mute Current curves in the Typical Performance Char- temperature stressing of the power transistor array. The Safe
acteristics section for values of attenuation per current out Operating graph in the Typical Performance Characteris-
of each MUTE pin. The resistance RM is calculated by the tics section shows the area of device operation where
following equation: SPiKe Protection Circuitry is not enabled. The SPiKe Pro-
tection Response waveform graph shows the waveform dis-
RM ≤ (|−VEE| − 2.6V) / IMUTE
tortion when SPiKe is enabled. Please refer to AN-898 for
Where IMUTE ≥ 0.5mA for each MUTE pin. more detailed information.
The MUTE pins can be tied together so that only one resistor
is required for the mute function. The mute resistor value THERMAL PROTECTION
must be chosen so that a minimum of 1mA is pulled through The LM4780 has a sophisticated thermal protection scheme
the resistor RM. This ensures that each amplifier is fully to prevent long-term thermal stress of the device. When the
operational. Taking into account supply line fluctuations, it is temperature on the die exceeds 150˚C, the LM4780 shuts
a good idea to pull out 1mA per MUTE pin or 2mA total if down. It starts operating again when the die temperature
both pins are tied together. drops to about 145˚C, but if the temperature again begins to
A turn-on MUTE or soft start circuit may also be used during rise, shutdown will occur again above 150˚C. Therefore, the
power up. A simple circuit like the one shown below may be device is allowed to heat up to a relatively high temperature
used. if the fault condition is temporary, but a sustained fault will
cause the device to cycle in a Schmitt Trigger fashion be-
tween the thermal shutdown temperature limits of 150˚C and
145˚C. This greatly reduces the stress imposed on the IC by
thermal cycling, which in turn improves its reliability under
sustained fault conditions.
Since the die temperature is directly dependent upon the
heat sink used, the heat sink should be chosen so that
200586A3
thermal shutdown is not activated during normal operation.
The RC combination of CM and RM1 may cause the voltage Using the best heat sink possible within the cost and space
at point A to change more slowly than the -VEE supply constraints of the system will improve the long-term reliability
voltage. Until the voltage at point A is low enough to have of any power semiconductor device, as discussed in the
0.5mA of current per MUTE pin flow through RM2, the IC will Determining the Correct Heat Sink section.
be in mute mode. The series combination of RM1 and RM2
DETERMlNlNG MAXIMUM POWER DISSIPATION
needs to satisfy the mute equation above for all operating
voltages or mute mode may be activated during normal Power dissipation within the integrated circuit package is a
operation. For a longer turn-on mute time, a larger time very important parameter requiring a thorough understand-
constant, τ = RC = RM1CM (sec), is needed. For the values ing if optimum power output is to be obtained. An incorrect
show above and with the MUTE pins tied together, the maximum power dissipation calculation may result in inad-
LM4780 will enter play mode when the voltage at point A is equate heat sinking causing thermal shutdown and thus
-17.6V. The voltage at point A is found with Equation (1) limiting the output power.
below. Equation (2) shows the theoretical maximum power dissipa-
VA(t) = (Vf - VO)e-t/τ (Volts) (1) tion point of each amplifier in a single-ended configuration
where VCC is the total supply voltage.
where:
PDMAX = (VCC)2 / 2π2RL (2)
t = time (sec)
τ = RC (sec)
Thus by knowing the total supply voltage and rated output
Vo = Voltage on C at t = 0 (Volts)
load, the maximum power dissipation point can be calcu-
Vf = Final voltage, -VEE in this circuit (Volts) lated. The package dissipation is twice the number which
results from Equation (2) since there are two amplifiers in
UNDER-VOLTAGE PROTECTION each LM4780. Refer to the graphs of Power Dissipation
Upon system power-up, the under-voltage protection cir- versus Output Power in the Typical Performance Charac-
cuitry allows the power supplies and their corresponding teristics section which show the actual full range of power
capacitors to come up close to their full values before turning dissipation not just the maximum theoretical point that re-
on the LM4780. Since the supplies have essentially settled sults from Equation (2).
to their final value, no DC output spikes occur. At power
down, the outputs of the LM4780 are forced to ground before
the power supply voltages fully decay preventing transients
on the output.

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LM4780
Application Information (Continued) BRIDGED AMPLIFIER APPLICATION
The LM4780 has two operational amplifiers internally, allow-
DETERMINING THE CORRECT HEAT SINK ing for a few different amplifier configurations. One of these
The choice of a heat sink for a high-power audio amplifier is configurations is referred to as “bridged mode” and involves
made entirely to keep the die temperature at a level such driving the load differentially through the LM4780’s outputs.
that the thermal protection circuitry is not activated under This configuration is shown in Figure 2. Bridged mode op-
normal circumstances. eration is different from the classical single-ended amplifier
The thermal resistance from the die to the outside air, θJA configuration where one side of its load is connected to
(junction to ambient), is a combination of three thermal re- ground.
sistances, θJC (junction to case), θCS (case to sink), and θSA A bridge amplifier design has a distinct advantage over the
(sink to ambient). The thermal resistance, θJC (junction to single-ended configuration, as it provides differential drive to
case), of the LM4780T is 0.8˚C/W. Using Thermalloy Ther- the load, thus doubling output swing for a specified supply
macote thermal compound, the thermal resistance, θCS voltage. Theoretically, four times the output power is pos-
(case to sink), is about 0.2˚C/W. Since convection heat flow sible as compared to a single-ended amplifier under the
(power dissipation) is analogous to current flow, thermal same conditions. This increase in attainable output power
resistance is analogous to electrical resistance, and tem- assumes that the amplifier is not current limited or clipped.
perature drops are analogous to voltage drops, the power A direct consequence of the increased power delivered to
dissipation out of the LM4780 is equal to the following: the load by a bridge amplifier is an increase in internal power
PDMAX = (TJMAX−TAMB) / θJA (3) dissipation. For each operational amplifier in a bridge con-
figuration, the internal power dissipation will increase by a
factor of two over the single ended dissipation. Thus, for an
where TJMAX = 150˚C, TAMB is the system ambient tempera-
audio power amplifier such as the LM4780, which has two
ture and θJA = θJC + θCS + θSA.
operational amplifiers in one package, the package dissipa-
tion will increase by a factor of four. To calculate the
LM4780’s maximum power dissipation point for a bridged
load, multiply Equation (2) by a factor of four.
This value of PDMAX can be used to calculate the correct size
heat sink for a bridged amplifier application. Since the inter-
nal dissipation for a given power supply and load is in-
creased by using bridged-mode, the heatsink’s θSA will have
20058652
to decrease accordingly as shown by Equation (4). Refer to
Once the maximum package power dissipation has been the section, Determining the Correct Heat Sink, for a more
calculated using Equation (2), the maximum thermal resis- detailed discussion of proper heat sinking for a given appli-
tance, θSA, (heat sink to ambient) in ˚C/W for a heat sink can cation.
be calculated. This calculation is made using Equation (4)
which is derived by solving for θSA in Equation (3). PARALLEL AMPLIFIER APPLICATION
θSA = [(TJMAX−TAMB)−PDMAX(θJC +θCS)] / PDMAX (4) Parallel configuration is normally used when higher output
current is needed for driving lower impedance loads (i.e. 4Ω
or lower) to obtain higher output power levels. As shown in
Again it must be noted that the value of θSA is dependent Figure 3 , the parallel amplifier configuration consist of de-
upon the system designer’s amplifier requirements. If the signing the amplifiers in the IC to have identical gain, con-
ambient temperature that the audio amplifier is to be working necting the inputs in parallel and then connecting the outputs
under is higher than 25˚C, then the thermal resistance for the in parallel through a small external output resistor. Any num-
heat sink, given all other things are equal, will need to be ber of amplifiers can be connected in parallel to obtain the
smaller. needed output current or to divide the power dissipation
across multiple IC packages. Ideally, each amplifier shares
SUPPLY BYPASSING
the output current equally. Due to slight differences in gain
The LM4780 has excellent power supply rejection and does the current sharing will not be equal among all channels. If
not require a regulated supply. However, to improve system current is not shared equally among all channels then the
performance as well as eliminate possible oscillations, the power dissipation will also not be equal among all channels.
LM4780 should have its supply leads bypassed with low- It is recommended that 0.1% tolerance resistors be used to
inductance capacitors having short leads that are located set the gain (Ri and Rf) for a minimal amount of difference in
close to the package terminals. Inadequate power supply current sharing.
bypassing will manifest itself by a low frequency oscillation
When operating two or more amplifiers in parallel mode the
known as “motorboating” or by high frequency instabilities.
impedance seen by each amplifier is equal to the total load
These instabilities can be eliminated through multiple by-
impedance multiplied by the number of amplifiers driving the
passing utilizing a large tantalum or electrolytic capacitor
load in parallel as shown by Equation (5) below:
(10µF or larger) which is used to absorb low frequency
variations and a small ceramic capacitor (0.1µF) to prevent RL(parallel) = RL(total) * Number of amplifiers (5)
any high frequency feedback through the power supply lines. Once the impedance seen by each amplifier in the parallel
If adequate bypassing is not provided, the current in the configuration is known then Equation (2) can be used with
supply leads which is a rectified component of the load this calculated impedance to find the amount of power dis-
current may be fed back into internal circuitry. This signal sipation for each amplifier. Total power dissipation (PDMAX)
causes distortion at high frequencies requiring that the sup- within an IC package is found by adding up the power
plies be bypassed at the package terminals with an electro- dissipation for each amplifier in the IC package. Using the
lytic capacitor of 470µF or more. calculated PDMAX the correct heat sink size can be deter-

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 LM4780
Application Information (Continued) than 50V/V. Gain settings below 10V/V may experience
instability and using the LM4780 for gains higher than 50V/V
mined. Refer to the section, Determining the Correct Heat will see an increase in noise and THD.
Sink, for more information and detailed discussion of proper The combination of Ri with Ci (see Figure 1) creates a high
heat sinking. pass filter. The low frequency response is determined by
these two components. The -3dB point can be found from
SINGLE-SUPPLY AMPLIFIER APPLICATION Equation (7) shown below:
The typical application of the LM4780 is a split supply am- fi = 1 / (2πRiCi) (Hz) (7)
plifier. But as shown in Figure 4, the LM4780 can also be
If an input coupling capacitor is used to block DC from the
used in a single power supply configuration. This involves
inputs as shown in Figure 5, there will be another high pass
using some external components to create a half-supply bias
filter created with the combination of CIN and RIN. When
which is used as the reference for the inputs and outputs.
using a input coupling capacitor RIN is needed to set the DC
Thus, the signal will swing around half-supply much like it
bias point on the amplifier’s input terminal. The resulting
swings around ground in a split-supply application. Along
-3dB frequency response due to the combination of CIN and
with proper circuit biasing, a few other considerations must
RIN can be found from Equation (8) shown below:
be accounted for to take advantage of all of the LM4780
functions, like the mute function. fIN = 1 / (2πRINCIN) (Hz) (8)
With large values of RIN oscillations may be observed on the
CLICKS AND POPS outputs when the inputs are left floating. Decreasing the
In the typical application of the LM4780 as a split-supply value of RIN or not letting the inputs float will remove the
audio power amplifier, the IC exhibits excellent “click” and oscillations. If the value of RIN is decreased then the value of
“pop” performance when utilizing the mute mode. In addition, CIN will need to increase in order to maintain the same -3dB
the device employs Under-Voltage Protection, which elimi- frequency response.
nates unwanted power-up and power-down transients. The
basis for these functions are a stable and constant half- HIGH PERFORMANCE CONSIDERATIONS
supply potential. In a split-supply application, ground is the Using low cost electrolytic capacitors in the signal path such
stable half-supply potential. But in a single-supply applica- as CIN and Ci (see Figures 1 - 5) will result in very good
tion, the half-supply needs to charge up at the same rate as performance. However, electrolytic capacitors are less linear
the supply rail, VCC. This makes the task of attaining a than other premium capacitors. Higher THD+N performance
clickless and popless turn-on more challenging. Any uneven may be obtained by using high quality polypropylene capaci-
charging of the amplifier inputs will result in output clicks and tors in the signal path. A more cost effective solution may be
pops due to the differential input topology of the LM4780. the use of smaller value premium capacitors in parallel with
To achieve a transient free power-up and power-down, the the larger electrolytic capacitors. This will maintain signal
voltage seen at the input terminals should be ideally the quality in the upper audio band where any degradation is
same. Such a signal will be common-mode in nature, and most noticeable while also coupling in the signals in the
will be rejected by the LM4780. In Figure 4, the resistor RINP lower audio band for good bass response.
serves to keep the inputs at the same potential by limiting the Distortion is introduced as the audio signal approaches the
voltage difference possible between the two nodes. This lower -3dB point, determined as discussed in the section
should significantly reduce any type of turn-on pop, due to an above. By using larger values of capacitors such that the
uneven charging of the amplifier inputs. This charging is -3dB point is well outside of the audio band will reduce this
based on a specific application loading and thus, the system distortion and improve THD+N performance.
designer may need to adjust these values for optimal perfor- Increasing the value of the large supply bypass capacitors
mance. will improve burst power output. The larger the supply by-
As shown in Figure 4, the resistors labeled RBI help bias up pass capacitors the higher the output pulse current without
the LM4780 off the half-supply node at the emitter of the supply droop increasing the peak output power. This will also
2N3904. But due to the input and output coupling capacitors increase the headroom of the amplifier and reduce THD.
in the circuit, along with the negative feedback, there are two
different values of RBI, namely 10kΩ and 200kΩ. These SIGNAL-TO-NOISE RATIO
resistors bring up the inputs at the same rate resulting in a In the measurement of the signal-to-noise ratio, misinterpre-
popless turn-on. Adjusting these resistors values slightly tations of the numbers actually measured are common. One
may reduce pops resulting from power supplies that ramp amplifier may sound much quieter than another, but due to
extremely quick or exhibit overshoot during system turn-on. improper testing techniques, they appear equal in measure-
ments. This is often the case when comparing integrated
PROPER SELECTION OF EXTERNAL COMPONENTS circuit designs to discrete amplifier designs. Discrete transis-
Proper selection of external components is required to meet tor amps often “run out of gain” at high frequencies and
the design targets of an application. The choice of external therefore have small bandwidths to noise as indicated below.
component values that will affect gain and low frequency
response are discussed below.
The gain of each amplifier is set by resistors Rf and Ri for the
non-inverting configuration shown in Figure 1. The gain is
found by Equation (6) below:
AV = 1 + Rf / Ri (V/V) (6)
For best noise performance, lower values of resistors are
used. A value of 1kΩ is commonly used for Ri and then
setting the value of Rf for the desired gain. For the LM4780
the gain should be set no lower than 10V/V and no higher

17 www.national.com
LM4780
Application Information (Continued) PHYSICAL IC MOUNTING CONSIDERATIONS
Mounting of the package to a heat sink must be done such
that there is sufficient pressure from the mounting screws to
insure good contact with the heat sink for efficient heat flow.
Over tightening the mounting screws will cause the package
to warp reducing contact area with the heat sink. Less
contact with the heat sink will increase the thermal resis-
tance from the package case to the heat sink (θCS) resulting
in higher operating die temperatures and possible unwanted
thermal shut down activation. Extreme over tightening of the
mounting screws will cause severe physical stress resulting
in cracked die and catastrophic IC failure. The recom-
20058699 mended mounting screw size is M3 with a maximum torque
of 50 N-cm. Additionally, it is best to use washers under the
Integrated circuits have additional open loop gain allowing screws to distribute the force over a wider area or a screw
additional feedback loop gain in order to lower harmonic with a wide flat head. To further distribute the mounting force
distortion and improve frequency response. It is this addi- a solid mounting bar in front of the package and secured in
tional bandwidth that can lead to erroneous signal-to-noise place with the two mounting screws may be used. Other
measurements if not considered during the measurement mounting options include a spring clip. If the package is
process. In the typical example above, the difference in secured with pressure on the front of the package the maxi-
bandwidth appears small on a log scale but the factor of 10in mum pressure on the molded plastic should not exceed
bandwidth, (200kHz to 2MHz) can result in a 10dB theoreti- 150N/mm2.
cal difference in the signal-to-noise ratio (white noise is Additionally, if the mounting screws are used to force the
proportional to the square root of the bandwidth in a system). package into correct alignment with the heat sink, package
In comparing audio amplifiers it is necessary to measure the stress will be increased. This increase in package stress will
magnitude of noise in the audible bandwidth by using a result in reduced contact area with the heat sink increasing
“weighting” filter (Note 16). A “weighting” filter alters the die operating temperature and possible catastrophic IC fail-
frequency response in order to compensate for the average ure.
human ear’s sensitivity to the frequency spectra. The weight-
ing filters at the same time provide the bandwidth limiting as LAYOUT, GROUND LOOPS AND STABILITY
discussed in the previous paragraph. The LM4780 is designed to be stable when operated at a
In addition to noise filtering, differing meter types give differ- closed-loop gain of 10 or greater, but as with any other
ent noise readings. Meter responses include: high-current amplifier, the LM4780 can be made to oscillate
1. RMS reading, under certain conditions. These oscillations usually involve
2. average responding, printed circuit board layout or output/input coupling issues.
3. peak reading, and When designing a layout, it is important to return the load
ground, the output compensation ground, and the low level
4. quasi peak reading. (feedback and input) grounds to the circuit board common
Although theoretical noise analysis is derived using true ground point through separate paths. Otherwise, large cur-
RMS based calculations, most actual measurements are rents flowing along a ground conductor will generate volt-
taken with ARM (Average Responding Meter) test equip- ages on the conductor which can effectively act as signals at
ment. the input, resulting in high frequency oscillation or excessive
Typical signal-to-noise figures are listed for an A-weighted distortion. It is advisable to keep the output compensation
filter which is commonly used in the measurement of noise. components and the 0.1µF supply decoupling capacitors as
The shape of all weighting filters is similar, with the peak of close as possible to the LM4780 to reduce the effects of PCB
the curve usually occurring in the 3kHz–7kHz region. trace resistance and inductance. For the same reason, the
ground return paths should be as short as possible.
LEAD INDUCTANCE In general, with fast, high-current circuitry, all sorts of prob-
Power op amps are sensitive to inductance in the output lems can arise from improper grounding which again can be
leads, particularly with heavy capacitive loading. Feedback avoided by returning all grounds separately to a common
to the input should be taken directly from the output terminal, point. Without isolating the ground signals and returning the
minimizing common inductance with the load. grounds to a common point, ground loops may occur.
Lead inductance can also cause voltage surges on the sup- “Ground Loop” is the term used to describe situations occur-
plies. With long leads to the power supply, energy is stored in ring in ground systems where a difference in potential exists
the lead inductance when the output is shorted. This energy between two ground points. Ideally a ground is a ground, but
can be dumped back into the supply bypass capacitors when unfortunately, in order for this to be true, ground conductors
the short is removed. The magnitude of this transient is with zero resistance are necessary. Since real world ground
reduced by increasing the size of the bypass capacitor near leads possess finite resistance, currents running through
the IC. With at least a 20µF local bypass, these voltage them will cause finite voltage drops to exist. If two ground
surges are important only if the lead length exceeds a couple return lines tie into the same path at different points there will
feet ( > 1µH lead inductance). Twisting together the supply be a voltage drop between them. The first figure below
and ground leads minimizes the effect. shows a common ground example where the positive input
ground and the load ground are returned to the supply
ground point via the same wire. The addition of the finite wire
resistance, R2, results in a voltage difference between the
two points as shown below.

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 LM4780
Application Information (Continued) device to the closest ground spot. As a final rule, make all
ground returns low resistance and low inductance by using
large wire and wide traces.
Occasionally, current in the output leads (which function as
antennas) can be coupled through the air to the amplifier
input, resulting in high-frequency oscillation. This normally
happens when the source impedance is high or the input
leads are long. The problem can be eliminated by placing a
small capacitor, CC, (on the order of 50pF to 500pF) across
the LM4780 input terminals. Refer to the External Compo-
nents Description section relating to component interaction
with Cf.

REACTIVE LOADING
It is hard for most power amplifiers to drive highly capacitive
loads very effectively and normally results in oscillations or
ringing on the square wave response. If the output of the
LM4780 is connected directly to a capacitor with no series
resistance, the square wave response will exhibit ringing if
the capacitance is greater than about 0.2µF. If highly capaci-
tive loads are expected due to long speaker cables, a
method commonly employed to protect amplifiers from low
impedances at high frequencies is to couple to the load
through a 10Ω resistor in parallel with a 0.7µH inductor. The
inductor-resistor combination as shown in the Figure 5 iso-
lates the feedback amplifier from the load by providing high
output impedance at high frequencies thus allowing the 10Ω
20058698
resistor to decouple the capacitive load and reduce the Q of
The load current IL will be much larger than input bias current the series resonant circuit. The LR combination also pro-
II, thus V1 will follow the output voltage directly, i.e. in phase. vides low output impedance at low frequencies thus shorting
Therefore the voltage appearing at the non-inverting input is out the 10Ω resistor and allowing the amplifier to drive the
effectively positive feedback and the circuit may oscillate. If series RC load (large capacitive load due to long speaker
there was only one device to worry about then the values of cables) directly.
R1 and R2 would probably be small enough to be ignored;
however, several devices normally comprise a total system. INVERTING AMPLIFIER APPLICATION
Any ground return of a separate device, whose output is in The inverting amplifier configuration may be used instead of
phase, can feedback in a similar manner and cause insta- the more common non-inverting amplifier configuration
bilities. Out of phase ground loops also are troublesome, shown in Figure 1. The inverting amplifier can have better
causing unexpected gain and phase errors. THD+N performance and eliminates the need for a large
The solution to most ground loop problems is to always use capacitor (Ci) reducing cost and space requirements. The
a single-point ground system, although this is sometimes values show in Figure 6 are only one example of an amplifier
impractical. The third figure above is an example of a single- with a gain of 20V/V (Gain = -Rf/Ri). For different resistor
point ground system. values, the value of RB should be eqaul to the parallel
combination of Rf and Ri.
The single-point ground concept should be applied rigor-
ously to all components and all circuits when possible. Vio- If the DC blocking input capacitor (CIN) is used as shown, the
lations of single-point grounding are most common among lower -3dB point is found using Equation (8) as discussed in
printed circuit board designs, since the circuit is surrounded the Proper Selection of External Components section.
by large ground areas which invite the temptation to run a

19 www.national.com
LM4780
Application Information (Continued)

20058621

FIGURE 6. Inverting Amplifier Application Circuit

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 LM4780
Application Information (Continued)

200586F3

FIGURE 7. Reference PCB Schematic

21 www.national.com
LM4780
Application Information (Continued)

LM4780 REFERENCE BOARD ARTWORK

200586D9 200586D8
Composite Layer Silk Layer

200586D7 200586E0
Top Layer Bottom Layer

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 LM4780
Application Information (Continued)

BILL OF MATERIALS FOR REFERENCE PCB

Symbol Value Tolerance Type/Description Comment
RIN1, RIN2 15kΩ 5% 1/4 Watt
RB1, RB2 1kΩ 1% 1/4 Watt
RF1, RF2 20kΩ 1% 1/4 Watt
Ri1, Ri2 1kΩ 1% 1/4 Watt
RSN1, RSN2, 2.7Ω 5% 1/4 Watt
RG 2.7Ω 5% 1/4 Watt
RM 10kΩ 5% 1/4 Watt
CIN1, CIN2 1µF 10% Metallized Polyester Film
Ci1, Ci2, 68µF 20% Electrolytic Radial / 50V
CSN1, CSN2 0.1µF 20% Monolithic Ceramic
CN1, CN2 15pF 20% Monolithic Ceramic
CS1, CS2, CS3 0.1µF 20% Monolithic Ceramic
CS4, CS5, CS6 10µF 20% Electrolytic Radial / 50V
CS7, CS8 1,000µF 20% Electrolytic Radial / 50V
S1 SPDT (on-on) Switch
J1, J2 Non-Switched PC Mount RCA
Jack
J4, J7, J8 PCB Banana Jack - BLACK
J3, J5, J6, J9 PCB Banana Jack - RED
U1 27 lead TO-220 Power Socket
with push lever release or
LM4780 IC

23 www.national.com
 LM4780 Overture™ Audio Power Amplifier Series
Stereo 60W, Mono 120W Audio Power Amplifier with Mute
Physical Dimensions inches (millimeters)
unless otherwise noted

Non-Isolated TO-220 27-Lead Package

Order Number LM4780TA
NS Package Number TA27A

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DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT AND GENERAL
COUNSEL OF NATIONAL SEMICONDUCTOR CORPORATION. As used herein:
1. Life support devices or systems are devices or 2. A critical component is any component of a life
systems which, (a) are intended for surgical implant support device or system whose failure to perform
into the body, or (b) support or sustain life, and can be reasonably expected to cause the failure of
whose failure to perform when properly used in the life support device or system, or to affect its
accordance with instructions for use provided in the safety or effectiveness.
labeling, can be reasonably expected to result in a
significant injury to the user.
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This datasheet has been download from:

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Datasheets for electronics components.

 National Semiconductor was acquired by Texas Instruments.
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This file is the datasheet for the following electronic components:

LM4780 - http://www.ti.com/product/lm4780?HQS=TI-null-null-dscatalog-df-pf-null-wwe

LM4780TA - http://www.ti.com/product/lm4780ta?HQS=TI-null-null-dscatalog-df-pf-null-wwe