IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 36, NO.

4, JULY/AUGUST 2000 1069

An Air-Gap-Flux-Oriented Vector Controller

for Stable Operation of Bearingless
Induction Motors
Takahiro Suzuki, Student Member, IEEE, Akira Chiba, Senior Member, IEEE, M. Azizur Rahman, Fellow, IEEE,
and Tadashi Fukao, Fellow, IEEE

Abstract—A bearingless induction machine has combined the strong flux distribution of a revolving magnetic field in
characteristics of an induction motor and magnetic bearings. It is the air gap between the stator and rotor [6], [7]. Thus, the
known that the magnetic suspension of the rotor becomes unstable information of the instantaneous orientation and amplitude of
at over load operation, particularly in transient conditions.
A novel air-gap-flux-oriented vector control scheme has been the revolving magnetic field is required in a controller of the
proposed to operate the bearingless induction motor during the bearingless motors. If there is an error in the field orientation,
high torque acceleration period. It has been found that there is an for example, the direction of a generated radial force has twice
optimal flux orientation for complete decoupling in radial force the angle error with respect to the radial force reference. If
generation. Test results in a laboratory bearingless induction correct amplitude and orientation of the revolving magnetic
motor validates the performance efficacy of proposed controller
at overload conditions. field are obtained in the controller, then the generated radial
force corresponds to the radial force reference in both direction
Index Terms—Bearingless motor, magnetic bearing, vector and amplitude. Thus, the vector control of the bearingless
control.
induction motor is necessary.
A direct field-oriented controller for induction-type bearing-
I. INTRODUCTION less motors was proposed [8]. On the other hand, an indirect
field-oriented controller has been proposed by the authors [9],
M AGNETIC bearings have been used in machine tools,
turbo-molecular pumps, compressors, blowers, com-
pact generators, and flywheels [1]–[4]. However, conventional
[10]. Successful operations in loaded conditions as well as tran-
sient conditions have been shown up to the rated values. It has
magnetic bearings have significant dimensions, and require a been experimentally learned that the magnetic suspension be-
number of windings as well as many single-phase inverters. comes stable with a slight increase in the reference value of rotor
resistance [10]. The increase in the results in the fact that
Bearingless motors, i.e., a hybrid of electrical motor and
an air-gap flux vector , rather than a rotor flux vector , is
magnetic bearings, have been expected to reduce dimensions,
aligned to a flux reference which is generated in the con-
number of inverters and associated cost. In induction-type
troller as a feedforward flux reference. The increase in is
bearingless motors, it was shown that only 0.0056 times of
not effective in overload operation in transient conditions. The
a motor VA is required for radial force control windings [5].
magnetic suspension becomes unstable during high torque ac-
This result indicates possible cost reduction in a bearingless
celeration because of significant variations in with respect
induction motor.
to . This fact suggests an application of air-gap-flux-oriented
In the radial positioning of the bearingless motors, radial
vector controller for the bearingless induction motor [11], [12].
forces are generated based on the feedback signals of radial
In this paper, an air-gap-flux-oriented vector controller has
displacement sensors detecting the movements of the rotor
been built. Successful operation at twice the rated torque current
shafts. The radial forces are generated taking advantage of
is shown experimentally. In addition, a universal field-oriented
controller is also applied to realize perfect decoupling of radial
Paper IPCSD 99–108, presented at the 1999 Industry Applications Society forces in two perpendicular axes. It is found that better perfor-
Annual Meeting, Phoenix, AZ, October 3–7, and approved for publication in
the IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS by the Electric Machines mance can be obtained with a slight lead angle in flux vector
Committee of the IEEE Industry Applications Society. Manuscript submitted rather than . The reasons for this fact are discussed.
for review June 1, 1999 and released for publication February 24, 2000. This
work was supported by the Ministry of Education, Science and Culture through
a Grant-in-Aid for Scientific Research. II. SYSTEM CONFIGURATION
T. Suzuki and A. Chiba are with the Department of Electrical Engi-
neering, Science University of Tokyo, Chiba 278-8510, Japan (e-mail:
A. Principle of Radial Force Generation
j7399621@ed.noda.sut.ac.jp; chiba@ee.noda.sut.ac.jp). The basic winding configurations of an induction-type bear-
M. A. Rahman is with the Faculty of Engineering and Applied Science,
Memorial University of Newfoundland, St. John’s, NF A1B 3X5 Canada ingless motor are shown in Fig. 1. Two sets of three-phase wind-
(e-mail: rahman@engr.mun.ca). ings are wound in the same stator slots. One is the four-pole
T. Fukao is with the Department of Electrical and Electronic Engi- windings for the production of motoring torque. It is called the
neering, Tokyo Insititute of Technology, Tokyo 152-8552, Japan (e-mail:
tfukao@ee.titech.ac.jp). motor winding. The other is the two-pole windings for con-
Publisher Item Identifier S 0093-9994(00)04775-7. trolling the rotor radial position in the air gap. This is referred
0093–9994/00$10.00 © 2000 IEEE
1070 IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 36, NO. 4, JULY/AUGUST 2000

bearingless induction motor. For this reason, it is very important

to choose the exact air-gap-flux orientation in a bearingless
induction motor.

B. Controller System Block Diagram

Fig. 3 shows the block diagram of a conventional field-ori-
ented control system of the bearingless induction motor aligned
with the rotor flux linkage [9]. It is to be noted that the two-pole
MMF wave traveling at the same velocity in space as the
four-pole wave requires the synchronous electrical frequency
of the two-pole current be one-half of the four-pole current.
The shaft speed is detected and speed errors are calculated. The
error is amplified in a proportional–integral (PI) controller. A
torque component current command is generated. A flux
current component is given as a constant. The amplitude
of line current motor , the phase lead angle command ,
Fig. 1. Principle of radial force generation.
and the slip frequency command are calculated from
and . Note that the slip frequency command is proportional
to the rotor time constant. If a reference command value
of the rotor resistance is the exactly the same as the actual
rotor resistance, then the rotor flux linkage is kept in a constant
amplitude. In addition, the speed of the rotor flux linkage
becomes . It is to be noted that is the sum of and
the measured rotor speed . Thus, the sinusoidal functions
and indicate the components of the rotor flux
linkage in and axes, respectively. The triangular functions
are shown in the modulation block of the radial position
controller of Fig. 3. These signals are supposed to indicate the
instantaneous flux components having a contribution in radial
force generation. The signal is hereafter referred to as
the controller flux reference .
In a radial position controller, radial positions and are de-
tected by displacement sensors. These displacements are com-
pared with the reference values. The errors are amplified by pro-
Fig. 2. Interference between  axis and  axis. portional-integral-derivative (PID) controllers to generate the
radial force commands and . In the modulation block of
as the radial force control winding. The four-pole flux and Fig. 3, based on radial force commands and sinusoidal func-
two-pole flux are generated by the winding currents and tions, radial force winding current commands and are gen-
in the and turns of stator windings, respectively. and erated so that the actual radial forces follow the radial force com-
are the mutually perpendicular rotor position control axes, re- mands.
spectively.
Under no-load balanced condition, if the rotor needs positive C. Acceleration Test Results with a Conventional System
radial force along the axis, one has to feed the radial force It has been known that the bearingless induction motor with
control windings current as shown in Fig. 1. It is observed rotor-flux-orientated vector controller can not maintain stable
from Fig. 1 that the flux density in the upper air gap is increased suspension and smooth operation at overload operating condi-
because both the and fluxes are in the same direction. On tions.
the other hand, the flux density in the lower air gap is decreased Fig. 4 shows the results of an acceleration test with a conven-
because and are in opposite directions. A positive radial tional controller with rotor-flux orientation in overloaded con-
force is produced in the -axis direction only. The opposite ditions. The reference value of the rotor resistance is set to
radial force can also be produced by the reverse current. The ra- an optimal value , i.e., a slightly higher than actual value
dial force in the -axis direction can also be produced using the . If the speed command is increased as a step function,
same principle by another radial force control windings perpen- then the rotor shaft speed is increased as a ramp function
dicular to . when the torque current command is limited to twice the rated
However, at loaded conditions of a bearingless induction value. The radial positions and of the rotor shaft have signif-
motor, there exists a phase difference between the and icant fluctuations. The rotor shaft is not successfully suspended
fluxes. A shifted radial force is produced as shown in Fig. 2. as the clearance of the touch down bearings is 100 m. The
It means that the radial force has and components is the detected flux waveform through an integrator connected
which cause interference in the radial force control for the to the terminals of a search coil wound around a stator tooth.
SUZUKI et al.: AN AIR-GAP-FLUX-ORIENTED VECTOR CONTROLLER 1071

Fig. 3. Block diagram of conventional bearingless induction motor.

The flux waveform and are not in phase during accel-

eration. The has a phase lead angle with respect to the .
Up to the rated current, the radial positions were stable with
a slight increase in the rotor resistance reference as shown in
[10]. However, under overloaded condition the radial positions
are unstable. It is considered that large influence of sharp tran-
sitional flux fluctuations makes this phase difference, because
of twice the rated value of torque. This phase angle difference
between the air-gap flux and occurs in spite of the slight in-
crease in the reference value of rotor resistance.

III. AN AIR-GAP-FLUX-ORIENTED VECTOR CONTROLLER

A. Analytical Model for Proposed Air-Gap-Flux-Oriented
Vector Controller
In an indirect field-oriented vector-controlled cage-type
three-phase induction motor, equations between the voltage,
current, and impedance are based on per phase equivalent
circuit. It is transformed from the – – axis to the – axis
using the transformation and, finally, to the –
axis using the transformation.
The voltage and current equations in matrix form are given in
(1), shown at the bottom of this page, where Fig. 4. Acceleration test results at overloaded condition with conventional
stator resistance; rotor-flux orientation.
rotor resistance;
stator self-inductance; nents of the stator; and subscript indicates components of the
rotor self-inductance; rotor.
mutual inductance; One can write from (1)
differential operator ( );
synchronous speed;
(2)
rotor shaft speed;
subscript indicates components of the axis; subscript in-
dicates components of the axis; subscript indicates compo- (3)

(1)
1072 IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 36, NO. 4, JULY/AUGUST 2000

Fig. 5. System configuration of proposed air-gap-flux-oriented vector controller.

The air-gap flux is given as By dividing the right-hand side of equation (11) by , one
obtains
(4)
(5)
(13)
Stator current and slip speed are determined as follows.
When it is assumed that an air-gap flux is perfectly aligned
with the axis, (5) reduces to Similarly, using (2), (3), (7), and (9) one can obtain the fol-
lowing expression:
(6)
(14)
and using (5) and (6), one can write
By dividing both sides of (14) by , the expression for ,
(7) can be obtained as

From (4), one can easily obtain (15)

(8)
The air-gap-flux-oriented vector controller scheme is based
on (13) and (15). The three-phase reference currents are pro-
In (3), slip speed is defined as
duced through the – axis transformation and transfor-
(9) mation which are identified in block of Fig. 5.

Again, (3) can be rearranged as B. Proposed Complete Drive System Configuration

Fig. 6 shows the bearingless induction motor system config-
(10) uration with the air-gap-flux-oriented vector controller. This
bearingless motor is four-pole machine. The slip frequency
From (7), (8), and (10), one can obtain the expression for command is redefined as in Fig. 6. From (13),
as

(11) (13-a)

where where

or (12)
SUZUKI et al.: AN AIR-GAP-FLUX-ORIENTED VECTOR CONTROLLER 1073

Fig. 6. Bearingless motor configuration with proposed controller.

From (15),

(15-a)

The terms and of (15-a) are set equal to

1, because motor current is assumed constant at 5 A. Thus,
one can approximate (15-a) as

(15-b)

This new system is implemented in software using a digital

signal processor (DSP) board. The and are calculated
values at the latest sampling periods. The torque current compo-
nent is also generated from a PI controller with speed errors.
The three-phase current commands are obtained from the
transformation, the – transformation and the transfor-
Fig. 7. Acceleration test at overload condition with the new air-gap-flux
mation as given in the following: orientation.

is increased as a step function, then the rotor shaft speed

is increased as a ramp function when the torque current com-
mand is also limited to twice the rated value. The rotating shaft
is successfully suspended with magnetic force during acceler-
ation. It is experimentally shown that the magnetic suspension
becomes stable with the new air-gap-flux-oriented vector con-
(16)
troller.
The sinusoidal functions and indicate flux
directions. The radial position control current command is gen- IV. STATIONARY LOAD TEST RESULTS
erated through modulation using the air-gap-flux reference. The The flux reference vector in a vector controller can be
phase shift angle is rotating at the same speed with the ro- easily shifted. Turn ratio “ ” is defined in references [11], [12].
tating air-gap field. The flux reference vector can shift to the rotor flux , the
air-gap flux , and the stator flux or among these flux vec-
C. Acceleration Test Results with the Proposed Controller tors as shown in Table I with the machine parameters. Thus, a
Fig. 7 shows the results of an acceleration test with the pro- flux vector between the air-gap flux and the stator flux can be
posed controller at overload conditions. If the speed command set to flux reference vector in a controller.
1074 IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 36, NO. 4, JULY/AUGUST 2000

Fig. 8. System configuration with turn ratio “a:”

TABLE I For the rated torque reference of Motor 2, is set to 5.0

CHANGES OF THE FLUX REFERENCE VECTOR WITH TURN RATIO “a” A for stationary load of Motor 1. The radial force references
, are measured at this condition with variation of turn
ratio “ ”. The radial force references , are expressed in
per-unit value using the following:

(17)

(18)

Test results are shown in Fig. 9 with turn ratio “ ” and ra-
dial force references. From this figure, it is shown that the ra-
dial force reference of -axis component does not vary with the
turn ratio “ ” However, the -axis component varies with the
Fig. 8 shows the system configuration of a vector controller turn ratio “ ” That is to say that the coupling between the two
with turn ratio “ .” This controller was originally proposed in perpendicular axes causes interference in the radial force refer-
[11], however, phase lead and lag block position has been opti- ences. When turn ratio “ ” is set to 1.04, decouple of the radial
mized. When the turn ratio “ ” is equal to 1, is equal to . force can be realized.
This system at “ ” will correspond to the air-gap-flux-ori-
ented vector controller shown in Fig. 6.
V. DISCUSSION OF PHASE SHIFT OF THE FLUX
It is shown by experiment and simulation that the air-gap-
flux-oriented vector controller is effective to make the mag- REFERENCE VECTOR
netic suspension stable in acceleration tests. However, it is quite The radial force is produced by using active unbalance of
possible that the air-gap-flux-oriented vector controller can not flux density of the air gap between the rotor and stator. There-
completely decouple to radial force components. Thus, the sta- fore, it is assumed that the magnetic suspension becomes stable
tionary load tests will be required to indicate the most suitable with the air-gap-flux-oriented vector controller. However, it is
flux reference vector for decoupling the radial forces. shown experimentally in Fig. 9 that successful decooupling of
The experimental machine is constructed from two units; one the radial forces takes place at turn ratio “ ” , but not
is called Motor 1, and the other is called Motor 2. Motor 1 is at the usual value of “ ” . Digital simulations in Matlab
driven as a motor and Motor 2 is driven as a generator. There- software carried out to determine how the flux reference vector
fore, Motor 1 is loaded with stationary load via Motor 2. move with variations of turn ratio “ .” Fig. 10 shows results of
The radial force references , are measured at no-load digital simulation. The standard value of phase lead angle of the
condition. These radial force references are considered as a stan- air-gap flux is based on flux reference generated in a con-
dard value. troller. It is evident from Fig. 10 that the phase angle of is
SUZUKI et al.: AN AIR-GAP-FLUX-ORIENTED VECTOR CONTROLLER 1075

Fig. 9. Turn ratio “a” versus radial force reference.

Fig. 11. Rotor resistance versus phase lead angle of .

Fig. 12. Leakage factor k versus phase lead angle of .

Fig. 10. Turn ratio “a” versus phase lead angle of . inductance is larger than that of the rotor. Therefore, it can be
considered that stator leakage inductance is larger than rotor
delayed when “ ” is set to a value larger than 1.0. Since the turn leakage inductance in this experimental bearingless machine.
ratio “ ” is larger than 1.0, is shifted toward . The expression for slip frequency of the universal field-
The simulation results shown in Fig. 11 illustrate the variation oriented controller from [11] and Fig. 8 can be written as
of the phase lead angle of the air-gap flux vector as a function
of rotor resistance value. The rotor resistance is expressed in
per-unit value based on actual value of the machine. (21)
It is seen that the phase angle of is advanced by the in-
crease of rotor resistance due to temperature rise of the wind-
ings. Thus, the fact of can be explained. The turn ratio When the rotor resistance is relatively increased by more
“ ” is set to a value larger than 1.0 in order to delay the phase than the reference value, then the denominator of (21) becomes
angle of which is advanced by 2.4 degree with the increase smaller, and is increased. Hence, the phase angle of is
of rotor resistance. However, there may be other causes like sat- advanced.
uration and leakage inductance variations for this phase shift. On the other hand, the rotor leakage inductance is relatively
In a conventional induction motor it is assumed that stator set to a value larger than the actual value, because the reference
leakage inductance and the rotor leakage inductance are equal value of the stator leakage is equal to that of the rotor leakage
in value. However, in all practical bearingless induction motors, inductance. Then, the is increased, the denominator of equa-
these stator and rotor leakage inductances are not necessarily tion (21) is decreased, and is increased. Consequently, the
equal in values. At least, the sum of the stator leakage induc- phase angle of will also be advanced.
tance and rotor leakage inductance has a set of fixed value, only
distribution of leakage components can be changed. The effect VI. CONCLUSION
of distribution of the leakage inductance is simulated in Fig. 12.
At overloaded conditions, the magnetic suspension of the
The leakage factor is defined as
bearingless induction motor becomes unstable. A new con-
(19) troller with air-gap-flux vector control has been proposed. It has
(20) been experimentally validated that stable magnetic suspension
where indicates stator leakage inductance, and indicates can be realized with the proposed controller, even in overload
rotor leakage inductance. conditions. It is noted that the phase angle of the flux reference
It is found from Fig. 12 that the phase angle of is delayed vector is delayed in stationary load conditions. It is found by
when is smaller than 0.5; that means that the stator leakage simulation that this phase shift of the flux reference vector is
inductance is smaller than that of rotor. The phase angle of caused by changes of rotor resistance value and the distribution
is advanced when is larger than 0.5, that is, the stator leakage of machine leakage inductance.
1076 IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 36, NO. 4, JULY/AUGUST 2000

ACKNOWLEDGMENT M. Azizur Rahman (S’67–M’68–SM’73–F’88) was

born in Santahar, Bangladesh, in 1941. He received
The authors would like to thank K. Yoshida who was a grad- the B.Sc.Eng. (Electrical) degree from Bangladesh
uate student at the Science University of Tokyo. University of Engineering and Technology (BUET),
Dhaka, Bangladesh, the M.A.Sc. degree from the
University of Toronto, Toronto, ON, Canada, and
REFERENCES the Ph.D. degree from Carleton University, Ottawa,
ON, Canada, in 1962, 1965, and 1968, respectively.
[1] G. Schweitzer, H. Bleuler, and A. Traxler, Active Magnetic Bear- In 1962, he joined the Department of Electrical En-
ings. Zurich, Switzerland: ETH Zurich Hochschulverlag AG, 1994. gineering, BUET, as a Lecture. He became an Assis-
[2] R. R. Agahi and U. Schroder, “Industrial high speed turbogenerator
tant Professor in 1969, Associate Professor in 1972,
system for energy recovery,” in Proc. 5th Int. Symp. Magnetic Bearings,
1996, pp. 381–387. and Professor in 1975. From 1963 to 1964, he was a Resident Consultant with
[3] M. Closs, P. Buhler, and G. Schweitzer, “Miniature active magnetic Dhaka Electric Supply. In 1968, he joined the Canadian General Electric Com-
bearing for very high rotational speeds,” in Proc. 6th Int. Symp. Mag- pany., Peterborough, ON, Canada, as a Research Scientist. He was a Visiting
netic Bearings, Boston, MA, Aug. 1998, pp. 62–67. Research Fellow at the Technische Hogeschool Eindhoven, Eindhoven, The
[4] M. Ohsawa, K. Yoshida, H. Ninomiya, T. Furuya, and E. Marui, “High- Netherlands, in 1973 and 1975, a Nuffield Fellow at Imperial College, London,
temperature blower for molten carbonate fuel cell supported by magnetic U.K., from 1974 to 1975, and a Visiting Fellow at the University of Toronto
bearings,” in Proc. 6th Inte. Symp. Magnetic Bearings, Boston, MA, in 1975 and 1984–1985. From 1975 to 1976, he was with Teshmont Consul-
Aug. 1998, pp. 32–41. tants, Winnipeg, Canada. In September 1976, he joined the Memorial Univer-
[5] E. Ito, A. Chiba, and T. Fukao, “A measurement of VA requirement in sity of Newfoundland, St. John’s, Canada, where he is currently a Professor
a induction type bearingless motor,” in Proc. 4th Int. Symp. Magnetic of Engineering and University Research Professor. From 1978 to 1979, he was
Suspension Technology, Gifu, Japan, Oct. 30, 1997, pp. 125–137. a Resident Consulting Engineer with General Electric Company, Schenectady,
[6] A. Chiba, K. Chida, and T. Fukao, “Principles and characteristics of a re- NY. During 1980–1981 and 1988, he was a Resident Consultant with the New-
luctance motor with windings of magnetic bearings,” in Proc. Int. Power foundland and Labrador Hydro. He was a Visiting Professor at Nanyang Tech-
Electronics Conf. (IPEC-Tokyo), Tokyo, Japan, Apr. 1990, pp. 919–926. nological University (NTU), Singapore, from 1991 to 1992, a Centennial Vis-
[7] A. Chiba, M. A. Rahman, and T. Fukao, “Radial force in a bearingless iting Professor at Tokyo Institute of Technology in 1992, a Visiting Professor
reluctance motor,” IEEE Trans. Magn., vol. 27, pp. 786–790, Mar. 1991. at the Science University of Tokyo in 1999, and will be a Visiting Professor
[8] R. Schob and J. Bichsel, “Vector control of the bearingless motor,” in at NTU until August 2000. He has more than 10 years of concurrent indus-
Proc. 4th Int. Symp. Magnetic Bearings, Zurich, Switzerland, 1994, pp. trial and consulting experience. His current interests include machines, power
327–332. electronics, power systems, digital protection, and intelligent controls. He has
[9] R. Furuichi, Y. Aikawa, K. Shimada, Y. Takamoto, A. Chiba, and T. authored more than 400 published technical papers. He is the holder of eight
Fukao, “A stable operation of induction type bearingless motor under patents.
loaded conditions,” IEEE Trans. Ind. Applicat., vol. 33, pp. 919–924,
Dr. Rahman is a Registered Professional Engineer in the Provinces of New-
July/Aug. 1997.
[10] A. Chiba, K. Yoshida, and T. Fukao, “Transient response of revolving foundland and Ontario, Canada, a member of the Institute of Electrical Engi-
magnetic field in induction type bearingless motors with secondary re- neers of Japan, a Fellow of the Institution of Electrical Engineers, U.K., a Life
sistance variations,” in Proc. 6th Int. Symp. Magnetic Bearings, Boston, Fellow of the Institution of Engineers, Bangladesh, and a Fellow of the Engi-
MA, Aug. 1998, pp. 461–475. neering Institute of Canada.
[11] R. W. De Doncker and D. W. Novotny, “The universal field oriented con-
troller,” IEEE Trans. Ind. Applicat., vol. 30, pp. 92–99, Jan./Feb. 1994.
[12] D. W. Novotny and T. A. Lipo, Vector Control and Dynamics of AC
Drives. New York: Oxford, 1996. Tadashi Fukao (M’85–SM’92–F’94) was born in
Shizuoka Prefecture, Japan, in 1940. He received
the B.S., M.S., and Ph.D. degrees in electrical
engineering from Tokyo Institute of Technology,
Tokyo, Japan, in 1964, 1966, and 1969, respectively.
Takahiro Suzuki (S’99) was born in Tokyo, Japan, From 1968 to 1977 and from 1977 to 1986, he was
in 1976. He received the B.E. degree in electrical a Research Associate and an Associate Professor, re-
engineering in 1999 from the Science University of spectively, at Tokyo Institute of Technology. Since
Tokyo, Chiba, Japan, where he is currently working 1986, he has been a Professor in the Department of
toward the M.E. degree. Electrical and Electronic Engineering. He is engaged
Mr. Suzuki is a member of the Institute of Elec- in research on static var compensators using multi-
trical Engineers of Japan. level inverters and high-speed motor and bearingless motor drive systems. He
was Vice-President of the Industry Applications Society of the Institute of Elec-
trical Engineers of Japan (IEEJ) from 1991 to 1992, and was its President from
1994 to 1995. He was an Editorial Director of the IEEJ from 1997 to 1998 and
Vice-President from 1998 to 1999. He also served as the Chairperson of the
Japanese National Committee on IEC TC22 (Power Electronics) from 1990 to
1997.

Akira Chiba (S’84–M’858–SM’97) was born in

Tokyo, Japan, in 1960. He received the B.S., M.S.,
and Ph.D. degrees in electrical engineering from
Tokyo Institute of Technology, Tokyo, Japan, in
1983, 1985, and 1988, respectively.
In 1988, he joined the Science University of
Tokyo, Chiba, Japan, as a Research Associate in the
Department of Electrical Engineering, Faculty of
Science and Technology. He was a Research Lecturer
from 1992 to 1993 and a Senior Lecturer from 1993
to 1997. He has been an Associate Professor since
1997. In 1990, he was a Natural Science and Engineering Research Council
of Canada International Post-Doctoral Fellow at the Memorial University of
Newfoundland, St. John’s, NF, Canada. He has been studying super-high-speed
synchronous reluctance motors, switched reluctance drives, and bearingless
drives.
Prof. Chiba is a member of the Institute of Electrical Engineers of Japan.