ANALOG FILTERS

FREQUENCY TRANSFORMATIONS

SECTION 5-5: FREQUENCY TRANSFORMATIONS

Until now, only filters using the lowpass configuration have been examined. In this
section, transforming the lowpass prototype into the other configurations: highpass,
bandpass, bandreject (notch) and allpass will be discussed .

Lowpass to Highpass

The lowpass prototype is converted to highpass filter by scaling by 1/s in the transfer
function. In practice, this amounts to capacitors becoming inductors with a value 1/C, and
inductors becoming capacitors with a value of 1/L for passive designs. For active designs,
resistors become capacitors with a value of 1/R, and capacitors become resistors with a
value of 1/C. This applies only to frequency setting resistor, not those only used to set
gain.

Another way to look at the transformation is to investigate the transformation in the s

plane. The complex pole pairs of the lowpass prototype are made up of a real part, α, and
an imaginary part, β. The normalized highpass poles are the given by:
α
αHP = Eq. 5-43
α2 + β2
and:
β
βHP = Eq. 5-44
α2 + β2
A simple pole, α0, is transformed to:
1
αω,HP = Eq. 5-45
α0
Lowpass zeros, ωz,lp, are transformed by:
1
ωZ,HP = Eq. 5-46
ωZ,LP

In addition, a number of zeros equal to the number of poles are added at the origin.

After the normalized lowpass prototype poles and zeros are converted to highpass, they
are then denormalized in the same way as the lowpass, that is, by frequency and
impedance.

As an example a 3 pole 1dB Chebyshev lowpass filter will be converted to a highpass

filter.

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From the design tables of the last section:

αLP1 = .2257
βLP1 = .8822
αLP2 = .4513
This will transform to:
αHP1= .2722
βHP1= 1.0639
αHP2= 2.2158
Which then becomes:

F01= 1.0982
α= .4958
Q= 2.0173

F02= 2.2158

A worked out example of this transformation will appear in a later section.

A highpass filter can be considered to be a lowpass filter turned on its side. Instead of a
flat response at DC, there is a rising response of n × (20dB/decade), due to the zeros at
the origin, where n is the number of poles. At the corner frequency a response of
n × (–20dB/decade) due to the poles is added to the above rising response. This results in
a flat response beyond the corner frequency.

Lowpass to Bandpass

Transformation to the bandpass response is a little more complicated. Bandpass filters

can be classified as either wideband or narrowband, depending on the separation of the
poles. If the corner frequencies of the bandpass are widely separated (by more than 2
octaves), the filter is wideband and is made up of separate lowpass and highpass sections,
which will be cascaded. The assumption made is that with the widely separated poles,
interaction between them is minimal. This condition does not hold in the case of a
narrowband bandpass filter, where the separation is less than 2 octaves. We will be
covering the narrowband case in this discussion.

As in the highpass transformation, start with the complex pole pairs of the lowpass
prototype, α and β. The pole pairs are known to be complex conjugates. This implies
symmetry around DC (0Hz.). The process of transformation to the bandpass case is one of
mirroring the response around DC of the lowpass prototype to the same response around
the new center frequency F0.

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This clearly implies that the number of poles and zeros is doubled when the bandpass
transformation is done. As in the lowpass case, the poles and zeros below the real axis are
ignored. So an nth order lowpass prototype transforms into an nth order bandpass, even
though the filter order will be 2n. An nth order bandpass filter will consist of n sections,
versus n/2 sections for the lowpass prototype. It may be convenient to think of the
response as n poles up and n poles down.

The value of QBP is determined by:

F0
QBP = Eq. 5-47
BW
where BW is the bandwidth at some level, typically –3dB.

A transformation algorithm was defined by Geffe ( Reference 16) for converting lowpass
poles into equivalent bandpass poles.

Given the pole locations of the lowpass prototype:

-α ± jβ Eq. 5-48
and the values of F0 and QBP, the following calculations will result in two sets of values
for Q and frequencies, FH and FL, which define a pair of bandpass filter sections.

C = α2 + β 2 Eq. 5-49
2α
D= Q Eq. 5-50
BP
C
E= +4 Eq. 5-51
QBP2
G = √ E2 - 4 D2 Eq. 5-52

Q=
√ E2 +DG 2 Eq. 5-53
Observe that the Q of each section will be the same.

The pole frequencies are determined by:

αQ Eq. 5-54
M=
QBP
W = M + √ M2 - 1 Eq. 5-55
FBP1 = F0 Eq. 5-56
W
FBP2 = W F0 Eq. 5-57

Each pole pair transformation will also result in 2 zeros that will be located at the origin.

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A normalized lowpass real pole with a magnitude of α0 is transformed into a bandpass

section where:
QBP
Q= α Eq. 5-58
0
and the frequency is F0.

Each single pole transformation will also result in a zero at the origin.

Elliptical function lowpass prototypes contain zeros as well as poles. In transforming the
filter the zeros must be transformed as well. Given the lowpass zeros at ± jωZ , the
bandpass zeros are obtained as follows:
αQ Eq. 5-59
M=
QBP
W = M + √ M2 - 1 Eq. 5-60
F0
FBP1 = Eq. 5-61
W
FBP2 = W F0 Eq. 5-62

Since the gain of a bandpass filter peaks at FBP instead of F0, an adjustment in the
amplitude function is required to normalize the response of the aggregate filter. The gain
of the individual filter section is given by:

√ ( )
2
F0 F Eq. 5-63
AR = A0 1 + Q2 - BP
FBP F0

where:
A0 = gain a filter center frequency
AR = filter section gain at resonance
F0 = filter center frequency
FBP = filter section resonant frequency.

Again using a 3 pole 1dB Chebychev as an example:

αLP1 = .2257
βLP1 = .8822
αLP2 = .4513
A 3 dB bandwidth of 0.5Hz. with a center frequency of 1Hz. is arbitrarily assigned. Then:

QBP = 2

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FREQUENCY TRANSFORMATIONS

Going through the calculations for the pole pair the intermediate results are:

C = 0.829217 D = 0.2257
E = 4.207034 G = 4.098611
M = 1.01894 W = 1.214489
and:
FBP1 = 0.823391 FBP2 = 1.214489
QBP1 = QBP2 = 9.029157
And for the single pole:
FBP3 = 1 QBP3 = 4.431642

Again a full example will be worked out in a later section.

Lowpass to Bandreject (Notch)

As in the bandpass case, a bandreject filter can be either wideband or narrowband,

depending on whether or not the poles are separated by 2 octaves or more. To avoid
confusion, the following convention will be adopted. If the filter is wideband, it will be
referred to as a bandreject filter. A narrowband filter will be referred to as a notch filter.

One way to build a notch filter is to construct it as a bandpass filter whose output is
subtracted from the input (1 – BP). Another way is with cascaded lowpass and highpass
sections, especially for the bandreject (wideband) case. In this case, the sections are in
parallel, and the output is the difference.

Just as the bandpass case is a direct transformation of the lowpass prototype, where DC is
transformed to F0, the notch filter can be first transformed to the highpass case, and then
DC, which is now a zero, is transformed to F0.

A more general approach would be to convert the poles directly. A notch transformation
results in two pairs of complex poles and a pair of second order imaginary zeros from
each lowpass pole pair.

First, the value of QBR is determined by:

F0 Eq. 5-64
QBR =
BW
where BW is the bandwidth at – 3dB.

Given the pole locations of the lowpass prototype

-α ± jβ Eq. 5-65

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and the values of F0 and QBR, the following calculations will result in two sets of values
for Q and frequencies, FH and FL, which define a pair of notch filter sections.
C = α 2 + β2
Eq. 5-66
α
D= Eq. 5-67
QBRC
β
E= Eq. 5-68
QBRC
F = E2 - 4 D2 + 4 Eq. 5-69

G=
√ F +
2 √
F2 + D2 E2
4
Eq. 5-70

H= DE Eq. 5-71
G
1
2√
K= (D + H)2 + (E + G)2 Eq. 5-72
K
Q= Eq. 5-73
D+H

The pole frequencies are given by:

F
FBR1 = K0 Eq. 5-74

FBR2 = K F0 Eq. 5-75

FZ = F0 Eq. 5-76

F0 = √ FBR1*FBR2 Eq. 5-77

where F0 is the notch frequency and the geometric mean of FBR1 and FBR2.

A simple real pole, α0, transforms to a single section having a Q given by:
Q = QBR α0 Eq. 5-78
with a frequency FBR = F0. There will also be transmission zero at F0.

In some instances, such as the elimination of the power line frequency (hum) from low
level sensor measurements, a notch filter for a specific frequency may be designed.

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Assuming that an attenuation of A dB is required over a bandwidth of B, then the

required Q is determined by:

ω0
Q= Eq. 5-79
B √10.1 A - 1

A 3 pole 1 dB Chebychev is again used as an example:

αLP1 = .2257
βLP1 = .8822
αLP2 = .4513

A 3dB bandwidth of 0.1 Hz. with a center frequency of 1Hz. is arbitrarily assigned.
Then:
QBR = 10

Going through the calculations for the pole pair yields the intermediate results:

C = 0.829217 D = 0.027218
E = 0.106389 F = 4.079171
G = 2.019696 H = 0.001434
K = 1.063139
and
FBR1 = 0.94061 FBR2 = 1.063139
QBR1 = QBR2 = 37.10499

and for the single pole

FBP3 = 1 QBP3 = 4.431642

Once again a full example will be worked out in a later section.

Lowpass to Allpass

The transformation from lowpass to allpass involves adding a zero in the right hand side
of the s plane corresponding to each pole in the left hand side.

In general, however, the allpass filter is usually not designed in this manner. The main
purpose of the allpass filter is to equalize the delay of another filter. Many modulation
schemes in communications use some form or another of quadrature modulation, which
processes both the amplitude and phase of the signal.

Allpass filters add delay to flatten the delay curve without changing the amplitude. In
most cases a closed form of the equalizer is not available. Instead the amplitude filter is
designed and the delay calculated or measured. Then graphical means or computer
programs are used to figure out the required sections of equalization.

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Each section of the equalizer gives twice the delay of the lowpass prototype due to the
interaction of the zeros. A rough estimate of the required number of sections is given by:

n = 2 ∆BW ∆T + 1

Where ∆BW is the bandwidth of interest in hertz and ∆T is the delay distortion over ∆BW
in seconds.

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SECTION 5-6: FILTER REALIZATIONS

Now that it has been decided what to build, it now must be decided how to build it. That
means that it is necessary to decide which of the filter topologies to use. Filter design is a
two step process where it is determined what is to be built (the filter transfer function)
and then how to build it (the topology used for the circuit).

In general, filters are built out of one-pole sections for real poles, and two-pole sections
for pole pairs. While you can build a filter out of three-pole, or higher order sections, the
interaction between the sections increases, and therefore, component sensitivities go up.

It is better to use buffers to isolate the various sections. In addition, it is assumed that all
filter sections are driven from a low impedance source. Any source impedance can be
modeled as being in series with the filter input.

In all of the design equation figures the following convention will be used:

H = circuit gain in the passband or at resonance

F0 = cutoff or resonant frequency in Hertz
ω0 = cutoff or resonant frequency in radians/sec.
Q = circuit “quality factor”. Indicates circuit peaking.
α = 1/Q = damping ratio

It is unfortunate that the symbol α is used for damping ratio. It is not the same as the α
that is used to denote pole locations (α ± jβ). The same issue occurs for Q. It is used for
the circuit quality factor and also the component quality factor, which are not the same
thing.

The circuit Q is the amount of peaking in the circuit. This is a function of the angle of the
pole to the origin in the s plane. The component Q is the amount of losses in what should
be lossless reactances. These losses are the parasitics of the components; dissipation
factor, leakage resistance, ESR (equivalent series resistance), etc. in capacitors and series
resistance and parasitic capacitances in inductors.

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Single Pole RC

The simplest filter building block is the passive RC section. The single pole can be either
lowpass or highpass. Odd order filters will have a single pole section.

The basic form of the lowpass RC section is shown in Figure 5-37(A). It is assumed that
the load impedance is high (> ×10), so that there is no loading of the circuit. The load will
be in parallel with the shunt arm of the filter. If this is not the case, the section will have
to be buffered with an op amp. A lowpass filter can be transformed to a highpass filter by
exchanging the resistor and the capacitor. The basic form of the highpass filter is shown
in Figure 5-37(B). Again it is assumed that load impedance is high.

(A) (B)
LOWPASS HIGHPASS

Figure 5-37: Single pole sections

The pole can also be incorporated into an amplifier circuit. Figure 5-38(A) shows an
amplifier circuit with a capacitor in the feedback loop. This forms a lowpass filter since
as frequency is increased, the effective feedback impedance decreases, which causes the
gain to decrease.

-
-
+
+

(A) (B)
LOWPASS HIGHPASS

Figure 5-38: Single pole active filter blocks

Figure 5-38(B) shows a capacitor in series with the input resistor. This causes the signal
to be blocked at DC. As the frequency is increased from DC, the impedance of the
capacitor decreases and the gain of the circuit increases. This is a highpass filter.

The design equations for single pole filters appear in Figure 5-66.

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FILTER REALIZATIONS

Passive LC Section

While not strictly a function that uses op amps, passive filters form the basis of several
active filters topologies and are included here for completeness.

As in active filters, passive filters are built up of individual subsections. Figure 5-39
shows lowpass filter sections. The full section is the basic two pole section. Odd order
filters use one half section which is a single pole section. The m derived sections, shown
in Figure 5-40, are used in designs requiring transmission zeros as well as poles.

(A) (B)
HALF SECTION FULL SECTION

Figure 5-39: Passive filter blocks (lowpass)

(A) (B)
HALF SECTION FULL SECTION

Figure 5-40: Passive filter blocks (lowpass m-derived)

A lowpass filter can be transformed into a highpass (see Figures 5-41 and 5-42) by simply
replacing capacitors with inductors with reciprocal values and vise versa so:
1 Eq. 5-80
LHP =
CLP
and
1
CHP = Eq. 5-81
LLP
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 OP AMP APPLICATIONS

Transmission zeros are also reciprocated in the transformation so:

1
ω Z ,HP = ω Eq. 5-82
Z ,LP

(A) (B)
HALF SECTION FULL SECTION

Figure 5-41: Passive filter blocks (highpass)

(A) (B)
HALF SECTION FULL SECTION

Figure 5-42: Passive filter blocks (highpass m-derived)

The lowpass prototype is transformed to bandpass and bandreject filters as well by using
the table in Figure 5-43.

For a passive filter to operate, the source and load impedances must be specified. One
issue with designing passive filters is that in multipole filters each section is the load for
the preceding sections and also the source impedance for subsequent sections, so
component interaction is a major concern. Because of this, designers typically make use
of tables, such as in Williams's book (Reference 2).

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FILTER REALIZATIONS
LOWPASS BANDPASS CIRCUIT
BRANCH CONFIGURATION VALUES
L
C
1
C=
C
ω02 L

1
L=
ω02 C
L C
L
1
Ca =
ω02 La
La Ca
La
Lb
1
Cb Lb =
Cb ω02 Cb

1
C1 =
L1 C2 L1 C1 C2 ω02 L1
1
L2 L2 =
ω02 C2

HIGHPASS BANDREJECT CIRCUIT

BRANCH CONFIGURATION VALUES

Figure 5-43: Lowpass → bandpass and highpass → bandreject transformation

Integrator

Any time that you put a frequency-dependent impedance in a feedback network the
inverse frequency response is obtained. For example, if a capacitor, which has a
frequency dependent impedance that decreases with increasing frequency, is put in the
feedback network of an op amp, an integrator is formed, as in Figure 5-44.

Figure 5-44: Integrator

The integrator has high gain (i.e. the open loop gain of the op amp) at DC. An integrator
can also be thought of as a low pass filter with a cutoff frequency of 0Hz.

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General Impedance Converter

Figure 5-45 is the block diagram of a general impedance converter. The impedance of this
circuit is:
Z1 Z3 Z5 Eq. 5-83
Z=
Z2 Z4

By substituting one or two capacitors into appropriate locations (the other locations being
resistors), several impedances can be synthesized (see Reference 25).

One limitation of this configuration is that the lower end of the structure must be
grounded.

Z1

Z2
+
-
-
+

Z3

Z4

Z5

Figure 5-45: General impedance converter

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FILTER REALIZATIONS

Active Inductor

Substituting a capacitor for Z4 and resistors for Z1, Z2, Z3 & Z5 in the GIC results in an
impedance given by:

sC R1 R3 R5
Z11 = Eq. 5-84
R2

By inspection it can be shown that this is an inductor with a value of:

C R1 R3 R5
L= Eq. 5-85
R2
This is just one way to simulate an inductor as shown in Figure 5-46.

R1

R2
+
-
-

C R1 R3 R5
+

R3 L= R2

R5

Figure 5-46: Active inductor

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Frequency Dependent Negative Resistor (FDNR)

By substituting capacitors for two of the Z1, Z3 or Z5 elements, a structure known as a

frequency dependant negative resistance (FDNR) is generated. The impedance of this
structure is:
sC2 R2 R4
Z11 = Eq. 5-86
R5

This impedance, which is called a D element, has the value:

D = C2 R4 Eq. 5-87
assuming
C1 = C2 and R2 = R5. Eq. 5-88

The three possible versions of the FDNR are shown in Figure 5-47.

+
+ +
-
- - -
+ - -
+ +

(A) (B) (C)

Figure 5-47: Frequency dependent negative resistor blocks

There is theoretically no difference in these three blocks, and so they should be
interchangeable. In practice though there may be some differences. Circuit (a) is
sometimes preferred because it is the only block to provide a return path for the amplifier
bias currents.

For the FDNR filter (see Reference 24), the passive realization of the filter is used as the
basis of the design. As in the passive filter, the FDNR filter must then be denormalized
for frequency and impedance. This is typically done before the conversion by 1/s. First
take the denormalized passive prototype filter and transform the elements by 1/s. This
means that inductors, whose impedance is equal to sL, transform into a resistor with an
impedance of L. A resistor of value R becomes a capacitor with an impedance of R/s; and
a capacitor of impedance 1/sC transforms into a frequency dependent resistor, D, with an
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impedance of 1/s2C. The transformations involved with the FDNR configuration and the
GIC implementation of the D element are shown in Figure 5-48. We can apply this
transformation to lowpass, highpass, bandpass or notch filters, remembering that the
FDNR block must be restricted to shunt arms.

L 1
R C

R C
1

Figure 5-48: 1/s transformation

A worked out example of the FDNR filter is included in the next section.

A perceived advantage of the FDNR filter in some circles is that there are no op amps in
the direct signal path, which can add noise and/or distortion, however small, to the signal.
It is also relatively insensitive to component variation. These advantages of the FDNR
come at the expense of an increase in the number of components required.

Sallen-Key

The Sallen-Key configuration, also known as a voltage control voltage source (VCVS),
was first introduced in 1955 by R. P. Sallen and E. L. Key of MIT’s Lincoln Labs (see
Reference 14). It is one of the most widely used filter topologies and is shown in Figure
5-49. One reason for this popularity is that this configuration shows the least dependence
of filter performance on the performance of the op amp. This is due to the fact that the op
amp is configured as an amplifier, as opposed to an integrator, which minimizes the gain-
bandwidth requirements of the op amp. This infers that for a given op amp, you will be
able to design a higher frequency filter than with other topologies since the op amp gain
bandwidth product will not limit the performance of the filter as it would if it were
configured as an integrator. The signal phase through the filter is maintained
(noninverting configuration).

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Another advantage of this configuration is that the ratio of the largest resistor value to the
smallest resistor value and the ratio of the largest capacitor value to the smallest capacitor
value (component spread) are low, which is good for manufacturability. The frequency
and Q terms are somewhat independent, but they are very sensitive to the gain parameter.
The Sallen-Key is very Q-sensitive to element values, especially for high Q sections. The
design equations for the Sallen-Key low pass are shown in Figure 5-67.

R1 C1

IN OUT

R2

-
C2

R3

R4

Figure 5-49: Sallen-Key lowpass filter

There is a special case of the Sallen–Key lowpass filter. If the gain is set to 2, the
capacitor values, as well as the resistor values, will be the same.

While the Sallen–Key filter is widely used, a serious drawback is that the filter is not
easily tuned, due to interaction of the component values on F0 and Q.

To transform the low pass into the highpass we simply exchange the capacitors and the
resistors in the frequency determining network (i.e., not the op amp gain resistors). This is
shown in Figure 5-50 (opposite). The comments regarding sensitivity of the filter given
above for the low pass case apply to the high pass case as well. The design equations for
the Sallen-Key high pass are shown in Figure 5-68.

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The bandpass case of the Sallen–Key filter has a limitation (see Figure 5-51 below). The
value of Q will determine the gain of the filter, i.e., it can not be set independent, as in the
lowpass or highpass cases. The design equations for the Sallen-Key bandpass are shown
in Figure 5-69.
C1 R1

IN OUT

C2

-
R2

R3

R4

Figure 5-50: Sallen-Key highpass filter

OUT
R2

IN
R1
+

C1 -

C2 R3

R4

R5

Figure 5-51: Sallen-Key bandpass filter

A Sallen–Key notch filter may also be constructed, but it has a large number of
undesirable characteristics. The resonant frequency, or the notch frequency, can not be
adjusted easily due to component interaction. As in the bandpass case, the section gain is
fixed by the other design parameters, and there is a wide spread in component values,
especially capacitors. Because of this and the availability of easier to use circuits, it is not
covered here.

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Multiple Feedback

The multiple feedback filter uses an op amp as an integrator as shown in Figure 5-52
below. Therefore, the dependence of the transfer function on the op amp parameters is
greater than in the Sallen-Key realization. It is hard to generate high Q, high frequency
sections due to the limitations of the open loop gain of the op amp. A rule of thumb is
that the open loop gain of the op amp should be at least 20dB (×10) above the amplitude
response at the resonant (or cutoff) frequency, including the peaking caused by the Q of
the filter. The peaking due to Q will cause an amplitude, A0:
A0 = H Q Eq. 5-89
where H is the gain of the circuit. The multiple feedback filter will invert the phase of the
signal. This is equivalent to adding the resulting 180° phase shift to the phase shift of the
filter itself.

OUT

R4 C5

R1
IN -
R3
+
C2

Figure 5-52: Multiple feedback lowpass

The maximum to minimum component value ratios is higher in the multiple feedback
case than in the Sallen-Key realization. The design equations for the multiple feedback
lowpass are given in Figure 5-70.

Comments made about the multiple feedback low pass case apply to the highpass case as
well (see Figure 5-53 opposite). Note that we again swap resistors and capacitors to
convert the lowpass case to the highpass case. The design equations for the multiple
feedback highpass are given in Figure 5-71.

The design equations for the multiple feedback bandpass case (see Figure 5-54 opposite)
are given in Figure 5-72.

This circuit is widely used in low Q (< 20) applications. It allows some tuning of the
resonant frequency, F0, by making R2 variable. Q can be adjusted (with R5) as well, but
this will also change F0.

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FILTER REALIZATIONS

Tuning of F0 can be accomplished by monitoring the output of the filter with the
horizontal channel of an oscilloscope, with the input to the filter connected to the vertical
channel. The display will be a Lissajous pattern. This pattern will be an ellipse that will
collapse to a straight line at resonance, since the phase shift will be 180°. You could also
adjust the output for maximum output, which will also occur at resonance, but this is
usually not as precise, especially at lower values of Q where there is a less pronounced
peak.

OUT

C4 R5

C1 C3
IN

-
+

R2

Figure 5-53: Multiple feedback highpass

OUT

R5
C4

IN R1 C3

-
+

R2

Figure 5-54: Multiple feedback bandpass

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State Variable

The state-variable realization (see Reference 11) is shown in Figure 5-55, along with the
design equations in Figure 5-73. This configuration offers the most precise
implementation, at the expense of many more circuit elements. All three major
parameters (gain, Q & ω0) can be adjusted independently, and lowpass, highpass, and
bandpass outputs are available simultaneously. Note that the lowpass and highpass
outputs are inverted in phase while the bandpass output maintains the phase. The gain of
each of the outputs of the filter is also independently variable. With an added amplifier
section summing the low pass and highpass sections the notch function can also be
synthesized. By changing the ratio of the summed sections, lowpass notch, standard notch
and highpass notch functions can be realized. A standard notch may also be realized by
subtracting the bandpass output from the input with the added op amp section. An allpass
filter may also be built with the four amplifier configuration by subtracting the bandpass
output from the input. In this instance, the bandpass gain must equal 2.

IN R1 R2 LP OUT

C1 C2
R3 R4 R5

- - -
+ + +

BP OUT

R6
HP OUT
R7

Figure 5-55: State variable filter

Since all parameters of the state variable filter can be adjusted independently, component
spread can be minimized. Also, variations due to temperature and component tolerances
are minimized. The op amps used in the integrator sections will have the same limitations
on op amp gain-bandwidth as described in the multiple feedback section.

Tuning the resonant frequency of a state variable filter is accomplished by varying R4 and
R5. While you don’t have to tune both, if you are varying over a wide range it is generally
preferable. Holding R1 constant, tuning R2 sets the lowpass gain and tuning R3 sets the
highpass gain. Bandpass gain and Q are set by the ratio of R6 & R7.

Since the parameters of a state variable filter are independent and tunable, it is easy to add
electronic control of frequency, Q and ω0. This adjustment is accomplished by using an
analog multiplier, multiplying DACs (MDACs) or digital pots, as shown in one of the
examples in a later section. For the integrator sections adding the analog multiplier or
MDAC effectively increases the time constant by dividing the voltage driving the resistor,
5.72
 ANALOG FILTERS
FILTER REALIZATIONS

which, in turn, provides the charging current for the integrator capacitor. This in effect
raises the resistance and, in turn, the time constant. The Q and gain can be varied by
changing the ratio of the various feedback paths. A digital pot will accomplish the same
feat in a more direct manner, by directly changing the resistance value. The resultant
tunable filter offers a great deal of utility in measurement and control circuitry. A worked
out example is given in Section 8 of this chapter.

Biquadratic (Biquad)

A close cousin of the state variable filter is the biquad as shown in Figure 5-56. The name
of this circuit was first used by J. Tow in 1968 (Reference 11) and later by L. C. Thomas
in 1971 (see Reference 12). The name derives from the fact that the transfer function is a
quadratic function in both the numerator and the denominator. Hence the transfer
function is a biquadratic function. This circuit is a slight rearrangement of the state
variable circuit. One significant difference is that there is not a separate highpass output.
The bandpass output inverts the phase. There are two lowpass outputs, one in phase and
one out of phase. With the addition of a fourth amplifier section, highpass, notch
(lowpass, standard and highpass) and allpass filters can be realized. The design equations
for the biquad are given in Figure 5-74.

IN R1 R2

R3 R4 C2 R5 R6

C1 LP OUT
(OUT OF
- - PHASE)

+ +

-
LP OUT
+ (IN PHASE)

BP OUT

Figure 5-56: Biquad filter

Referring to Figure 5-74, the allpass case of the biquad, R8 = R9/2 and R7 = R9. This is
required to make the terms in the transfer function line up correctly. For the highpass
output, the input, bandpass and second lowpass outputs are summed. In this case the
constraints are that R1 = R2 = R3 and R7 = R8 = R9.

Like the state variable, the biquad filter is tunable. Adjusting R3 will adjust the Q.
Adjusting R4 will set the resonant frequency. Adjusting R1 will set the gain. Frequency
would generally be adjusted first followed by Q and then gain. Setting the parameters in
this manner minimizes the effects of component value interaction.

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 OP AMP APPLICATIONS

Dual Amplifier Bandpass (DAPB)

The Dual Amplifier bandpass filter structure is useful in designs requiring high Qs and
high frequencies. Its component sensitivity is small, and the element spread is low. A
useful feature of this circuit is that the Q and resonant frequency can be adjusted more or
less independently.

Referring to Figure 5-57 below, the resonant frequency can be adjusted by R2. R1 can
then be adjusted for Q. In this topology it is useful to use dual op amps. The match of the
two op amps will lower the sensitivity of Q to the amplifier parameters.

+ OUT
- R4

R5

C R3

R2

IN R1 -
+

Figure 5-57: Dual amplifier bandpass filter

It should be noted that the DABP has a gain of 2 at resonance. If lower gain is required,
resistor R1 may be split to form a voltage divider. This is reflected in the addendum to the
design equations of the DABP, Figure 5-75.

5.74
 ANALOG FILTERS
FILTER REALIZATIONS

Twin T Notch

The twin T is widely used as a general purpose notch circuit as shown in Figure 5-58. The
passive implementation of the twin T (i.e. with no feedback) has a major shortcoming of
having a Q that is fixed at 0.25. This issue can be rectified with the application of
positive feedback to the reference node. The amount of the signal feedback, set by the
R4/R5 ratio, will determine the value of Q of the circuit, which, in turn, determines the
notch depth. For maximum notch depth, the resistors R4 and R5 and the associated op
amp can be eliminated. In this case, the junction of C3 and R3 will be directly connected
to the output.
IN R1 R2
OUT
+

-
C3

R4

+
R3 -
R5
C1 C2

Figure 5-58: Twin-T notch filter

Tuning is not easily accomplished. Using standard 1% components a 60dB notch is as
good as can be expected, with 40 –50dB being more typical.

The design equations for the Twin T are given in Figure 5-76.

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 OP AMP APPLICATIONS

Bainter Notch

A simple notch filter is the Bainter circuit (see Reference 21). It is composed of simple
circuit blocks with two feedback loops as shown in Figure 5-59. Also, the component
sensitivity is very low.

This circuit has several interesting properties. The Q of the notch is not based on
component matching as it is in every other implementation, but is instead only dependant
on the gain of the amplifiers. Therefore, the notch depth will not drift with temperature,
aging and other environmental factors. The notch frequency may shift, but not the depth.
R4

C1 NOTCH
OUT

IN R1 R2 R3
- R5
+ +

- -

C2
R7

R8
R6

Figure 5-59: Bainter notch filter

Amplifier open loop gain of 104 will yield a Qz of > 200. It is capable of orthogonal
tuning with minimal interaction. R6 tunes Q and R1 tunes ωZ. Varying R3 sets the ratio of
ω0/ωZ produces lowpass notch (R4 > R3), notch (R4 = R3) or highpass notch (R4 < R3).

The design equations of the Bainter circuit are given in Figure 5-77.

5.76
 ANALOG FILTERS
FILTER REALIZATIONS

Boctor Notch

The Boctor circuits (see References 22, 23), while moderately complicated, uses only one
op amp. Due to the number of components, there is a great deal of latitude in component
selection. These circuits also offer low sensitivity and the ability to tune the various
parameters more or less independently.

R2 R1

R4

C2 R6 C1

IN

+ OUT

R3

R5

Figure 5-60: Boctor lowpass notch filter

There are two forms, a lowpass notch (Figure 5-60 above) and a highpass notch (Figure
5-61 below). For the lowpass case, the preferred order of adjustment is to tune ω0 with
R4, then Q0 with R2, next Qz with R3 and finally ωz with R1.

R4

R5
OUT

C2 R2
-
IN
+
C1

R1
R3

R6

Figure 5-61: Boctor highpass filter

5.77
 OP AMP APPLICATIONS

In order for the components to be realizable we must define a variable, k1, such that:

ω 02 Eq. 5-90
< k1 < 1
ωz2
The design equations are given in Figure 5-78 for the lowpass case and in Figure 5-79 for
the highpass case.

In the highpass case circuit gain is require and it applies only when
1
Q< Eq. 5-91
ω2
1 - Z2
ω0
but a highpass notch can be realized with one amplifier and only 2 capacitors, which can
be the same value. The pole and zero frequencies are completely independent of the
amplifier gain. The resistors can be trimmed so that even 5% capacitors can be used.

5.78
 ANALOG FILTERS
FILTER REALIZATIONS

"1 – Bandpass" Notch

As mentioned in the state variable and biquad sections, a notch filter can be built as
1 - BP. The bandpass section can be any of the all pole bandpass realizations discussed
above, or any others. Keep in mind whether the bandpass section is inverting as shown in
Figure 5-62 (such as the multiple feedback circuit) or non-inverting as shown in
Figure 5-63 (such as the Sallen-Key), since we want to subtract, not add, the bandpass
output from the input.

IN R/2 R

BANDPASS

R
- OUT

Figure 5-62: 1 − BP filter for inverting bandpass configurations

R R

-
IN R
BANDPASS +
OUT

Figure 5-63: 1 − BP filter for noninverting bandpass configurations

It should be noted that the gain of the bandpass amplifier must be taken into account in
determining the resistor values. Unity gain bandpass would yield equal values.

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 OP AMP APPLICATIONS

First Order Allpass

The general form of a first order allpass filter is shown in Figure 5-64. If the function is a
simple RC highpass (Figure 5-64A), the circuit will have a have a phase shift that goes
from -180° at 0Hz. and 0°at high frequency. It will be -90° at ω = 1/RC. The resistor may
be made variable to allow adjustment of the delay at a particular frequency.
IN R1 R1 OUT IN R1 R1 OUT

C -
R -
+
+

R
C

(A) (B)

Figure 5-64: First order allpass filters

If the function is changed to a lowpass function (Figure 5-64B), the filter is still a first
order allpass and the delay equations still hold, but the signal is inverted, changing from
0° at DC to −180° at high frequency.

Second Order Allpass

A second order allpass circuit shown in Figure 5-65 was first described by Delyiannis
(see Reference 17). The main attraction of this circuit is that it only requires one op amp.
Remember also that an allpass filter can also be realized as 1 – 2BP.
C

OUT

R2

C
R1
IN -
+

R3

R4

Figure 5-65: Second order allpass filter

We may use any of the all pole realizations discussed above to build the filter, but you
need to be aware of whether the BP inverts the phase or not. We must also be aware that
the gain of the BP section must be 2. To this end, the DABP structure is particularly
useful, since its gain is fixed at 2.

Figures 5-66 through 5-81 following summarize design equations for various active filter
realizations. In all cases, H, ωo, Q, and α are given, being taken from the design tables.

5.80
 ANALOG FILTERS
FILTER REALIZATIONS

SINGLE POLE

LOWPASS HIGHPASS
C
R
IN OUT IN OUT

R
C

VO 1 VO sC R
VIN
=
sC R + 1 VIN = sC R + 1

1 1
Fo = Fo =
2π R C 2π R C

C IN
C Rin Rf

Rf
- OUT
IN Rin
- OUT +

VO Rf 1 VO Rf sC R1
= - = - Rin sC R1 + 1
VIN Rin sC R2 + 1 VIN

Rf Rf
Ho = - Ho = - Rin
Rin

1 1
Fo = Fo =
2π Rf C 2π Rin C

Figure 5-66: Single pole filter design equations

5.81
 OP AMP APPLICATIONS

SALLEN-KEY LOWPASS
R1 C1
IN OUT

R2

-
C2

+H ω02 R3
s2 + α ω0 s + ω02 R4

1
H
VO R1 R2 C1 C2
=

[ ]
VIN
C2 C2 C1 1
s2 + s + + (1-H) +
R1 R2 R2 R1 R2 C1 C2

CHOOSE: C1 R3
THEN: k = 2 π FO C1 R3
R4 =
α2 (H-1)
m= + (H-1)
4
C2 = m C1
2
R1 =
αk
α
R2 =
2mk

Figure 5-67: Sallen-Key lowpass design equations

5.82
 ANALOG FILTERS
FILTER REALIZATIONS

SALLEN-KEY HIGHPASS
C1
R1
IN OUT

C2

-
R2

+H s2 R3
s2 + α ω0 s + ω02 R4

VO H s2

]
=

[
VIN C2 C1 C2
+ + (1-H) 1
s2 + s R2 R2 R1 +
C1 C2 R1 R2 C1 C2

CHOOSE: C1 R3
THEN: k = 2 π FO C1 R3
R4 =
(H-1)
C2 = C1

α + √ α2 + (H-1)
R1 =
4k

4 1
R2 = * k
α + √ α2 + (H-1)

Figure 5-68: Sallen-Key highpass design equations

5.83
 OP AMP APPLICATIONS

SALLEN-KEY BANDPASS
OUT

R2

R1 C1
IN +

-
C2 R3
+H ω0 s
R4
s2 + α ω0 s + ω02
R5

1
H s R1C2
VO

[ ]
=
VIN
s2 + s
C1 (C1 + C2) C2 C1
R3
+
R1
+
C1 C2
+
R2 R2 +
1
(
R1 + R2
R3 C1 C2 R1 R2 )
CHOOSE: C1 R4
THEN: k = 2 π FO C1 R4
R5 =
1 (H-1)
C2 = C1
2
2
R1 =
k
2
R2 =
3k
4
R3 =
k

(
H = 1 6.5 - 1
3 Q )

Figure 5-69: Sallen-Key bandpass design equations

5.84
 ANALOG FILTERS
FILTER REALIZATIONS

MULTIPLE FEEDBACK LOWPASS

OUT

R4 C5

R1 R3
IN -
+
C2
-H ω02
s2 + α ω0 s + ω02

1
-H
VO R3 R4 C2 C5
=
VIN
s2 + s
1 1
(
+
1
+
C2 R1 R3 R4
1
) +
1
R3 R4 C2 C5

CHOOSE: C5
THEN: k = 2 π FO C5
4
C2 = 2 ( H + 1 ) C5
α
R1 = α
2Hk
R3 = α
2 (H + 1) k
α
R4 =
2k

Figure 5-70: Multiple feedback lowpass design equations

5.85
 OP AMP APPLICATIONS

MULTIPLE FEEDBACK HIGHPASS

OUT

C4 R5

C1 C3

IN -
+

R2
-H s2
s2 + α ω0 s + ω02

C1
- s2
VO C4
=
VIN (C1 + C3 + C4) 1
s2 + s +
C3 C4 R5 R2 R5 C3 C4

CHOOSE: C1
THEN: k = 2 π FO C1

C3 = C1
C1
C4 =
H
α
R2 =
( ) 1
k 2+ H

H (2 + H )
1
R5 =
αk

Figure 5-71: Multiple feedback highpass design equations

5.86
 ANALOG FILTERS
FILTER REALIZATIONS

MULTIPLE FEEDBACK BANDPASS

OUT

C4 R5

R1
IN -

C3 +

R2

- H ω0 s
s2 + α ω0 s + ω02

1
-s
VO R1 C4
=
VIN
s2 + s
( C3 + C4 )
C3 C4 R5
+
1
R5 C3 C4 ( R11 + R21 )
CHOOSE: C3
THEN: k = 2 π FO C3

C4 = C3
1
R1 =
Hk
R2 = 1
( 2Q - H) k
2Q
R5 =
k

Figure 5-72: Multiple feedback bandpass design equations

5.87
 OP AMP APPLICATIONS

STATE VARIABLE (A)

IN R1 R2 LP OUT

R3 R4 C1 R5 C2

- - -
+ + +

BP OUT

R6
HP OUT
R7

R2
ALP (s = 0) = - CHOOSE R1:
R1
R2 = ALP R1
R3
AHP (s = ∞) = - R3 = AHP R1
R1

ω0 =
√ R2 R4 R3
R5 C1 C2 CHOOSE C:
2 π FO
√
AHP
R=
LET R4 = R5 = R, C1 = C2 = C C ALP

R6 + R7 CHOOSE R7:
R7
ABP (s = ω0) = R6 =
( ) ( )
1 1 1 1
R1 + +
R1 R2 R3 R7 √ R2 R3 Q 1 1 1
+ +
R1 R2 R3

Figure 5-73A: State variable design equations

5.88
 ANALOG FILTERS
FILTER REALIZATIONS

STATE VARIABLE (B)

FOR NOTCH:
R8 R10
HP OUT NOTCH OUT

R9
ωZ2 R9 R2
=
-
LP OUT
+
ωO2 R8 R3

CHOOSE R10:
CHOOSE AHP, ALP, ANOTCH = 1:
FOR ωZ = ωO: R8 = R9 = R10

FOR ωZ < ωO: R9 = R10

ω02
R8 = R10
ωZ2

FOR ωZ > ωO: R8 = R10

R8 R10 ω 2
R9 = ωZ2 R10
BP OUT NOTCH OUT 0

-
R9
+
INPUT

R11 CHOOSE ANOTCH = 1:

CHOOSE R10:
R8 = R9 = R11 = R10

Figure 5-73B: State variable design equations

5.89
 OP AMP APPLICATIONS

STATE VARIABLE (C)

ALLPASS
R8 R10
INPUT AP OUT

R9
-
BP OUT
+

H=1
R8 = R10
R9 = R8/2

Figure 7-73C: State variable design equations

5.90
 ANALOG FILTERS
FILTER REALIZATIONS

BIQUADRATIC (A)
R1 R2

IN

R3 R4 C2 R5 R6

C1
- - LP OUT
+ +

+ LP OUT

BP OUT
CHOOSE C, R2, R5 CHOOSE C, R5, R7
K= 2 π f0 C K= 2 π f0 C
C1 = C2 =C
C1 = C2 =C
R2 1
R1 = R1 = R2 = R3 =
H kα
1 1
R3 = R4 = 2
kα k R2
R6 = R5
1
R4 = 2
k R2
HIGHPASS
R7 R10
INPUT HP OUT

R8
BP OUT R7 = R8 = R9 = R
R9 R
R10 =
LP2 OUT H
-

Figure 5-74A: Biquad design equations

5.91
 OP AMP APPLICATIONS

BIQUADRATIC (B)

NOTCH
R7 R9
BP OUT NOTCH OUT

R8
INPUT
-

H=1
R7 = R8 = R9

ALLPASS
R7 R9
INPUT AP OUT

R8
BP OUT
-

H=1
R7 = R9
R8 = R7/2

Figure 5-74B: Biquad design equations

5.92
 ANALOG FILTERS
FILTER REALIZATIONS

DUAL AMPLIFIER BANDPASS

+
OUT
- R4

+H ω0 s R5

s + α ω0 s + ω02
2
C
R3

R2
-
R1
IN +

C 2 s
VO R1 C
=
VIN
s2 + s 1 + 1
R1 C R2 R3 C2
CHOOSE: C R4
THEN: R5 = R4
1
R=
2 π F0 C
R1 = Q R

R2 = R3 = R
R2
FOR GAINS LESS THAN 2 (GAIN = AV):
R1A
R1A = 2R1
IN
AV
C
R1B
R1A AV
R1B = 2 - AV

Figure 5-75: Dual amplifier bandpass design equations

5.93
 OP AMP APPLICATIONS

TWIN T NOTCH
R1 R2
IN
+ OUT
-
C3

R4

+
R3 -
R5
C1 C2

1 s2 + ω02
s2 +
V0 RC s2 + 4ω0(1-K)s + ω02
=
VIN
s2 +
1
RC
4 ( 1-
R5
R4 + R5 ) s+
1
RC

CHOOSE: C R’

k = 2 π F0 C R4 = (1 - K) R’
1 R5 = K R’
R=
k
R = R1 = R2 = 2 R3 1
C3 K= 1-
C = C1 = C2 = 4Q
2
for K = 1, eliminate R4 and R5
1
F0 = (i.e R5 > 0, Q > ∞)
2πRC
for R >> R4, eliminate buffer

Figure 5-76: Twin-T notch design equations

5.94
 ANALOG FILTERS
FILTER REALIZATIONS

BAINTER NOTCH
R4
OUT

C1
IN R1 R2 R3 - R5
+ +

- -

C2 R7

R8
R6

H ( s2 + ωz2)
ω
s2 + Q0 s + ω02
VOUT K2 * [S + R3 R5K1C1 C2 ]
2

VIN = (R5 + R6) K2

s2 + s+
R5 R6 C2 R4 R5 C1 C2
CHOOSE C1, R1,R7,K1, K2
C2 = C1 =C
k = 2 π FO C
K2
R4 =
R2 = K1* R1 2Qk

Z= ω( )
ωZ 2
0
2Q
R5 = R6 = k

K1 R8 = (K2 – 1) R7
R3 =
2ZQk

Figure 5-77: Bainter notch

5.95
 OP AMP APPLICATIONS

BOCTOR NOTCH
LOWPASS

R2 R1
R4

C2 R6 C1
IN

-
R3 OUT
+ ωz
+
H ( s2 2)

ω
s2 + Q0 s + ω02 R5

R1 + (R2||R4) + R6
s2 +
VOUT R1 (R2||R4) R6 C1 C2
= H
VIN
s2 + ( R6 1C2 + (R2||R4)
1
C2 ) s +
1
R4 R6 C1 C2

GIVEN ω0, ωZ, Q0

CHOOSE R6 R5 C1

R4 =
1
ω0 C1 2Q0
R3 = (R6
R1
+ 2 ) R5
C1
C2
R4 R6 R4
R2 = C2 = 4 Q02 C1
R4 + R6 R6

(
R1 = 1 R6 ωZ2 - 1 )
2

2 R4 ω0

Figure 5-78: Boctor notch, lowpass

5.96
 ANALOG FILTERS
FILTER REALIZATIONS

BOCTOR NOTCH
HIGHPASS (A)

H ( s2 + ωz2) R4
R5
ω OUT
s2 + Q0 s + ω02
C2 R2
-
IN
+
C1

R1
R3
1
Q< FZ2 R6
1-
F02

VOUT (1 + R5
R4 ) (s 2 +
1
R1 R2 C1 C2 )
=
VIN
s2 + [ 1
REQ1 C1(1-
R
R1 R2 )] s + R
R EQ1 EQ2
EQ1
1
REQ2 C1 C2
WHERE: REQ1 = R1 || R3 || R6
REQ2 = R2 + (R4 || R5)

GIVEN: FZ F0 H or FZ Q H

1 1
Q= F0 = F Z

√
1
√( 2
FZ2
F02
-1 ) 1-
2Q2

1
Y=
Q 1-( FZ2
F02
)

Figure 5-79A: Boctor highpass design equations

5.97
 OP AMP APPLICATIONS

BOCTOR NOTCH
HIGHPASS (B)

GIVEN: C, R2, R3
R4
C1 = C2 = C R5 OUT
1
REQ1 =
C Y 2π F0 IN C2 R2
2
REQ2 = Y REQ1
-

+
C1
R4 = REQ2 – R2 ( )
H
H-1
R1
R5 = (H-1) R4 R3

1 R6
R1 =
)2
(2π F0 R2 C2
R6 = REQ1

Figure 5-79-B: Boctor highpass design equations (continued)

5.98
 ANALOG FILTERS
FILTER REALIZATIONS

FIRST ORDER ALLPASS

R1 R1
IN OUT

1 C -
s-
VO RC +
VIN = 1
s+
RC R

PHASE SHIFT (φ) = - 2 TAN-1

(2RπCF )
2RC
GROUP DELAY =
( 2 π F R C)2 + 1
DELAY AT DC = 2 R C
GIVEN A PHASE SHIFT OF φ AT A FREQUENCY = F

R C = 2 π F TAN - ( φ2 )
R1 R1
IN OUT

-
R
+

DESIGN AS ABOVE EXCEPT

THE SIGN OF THE PHASE CHANGES C

Figure 5-80: First order allpass design equations

5.99
 OP AMP APPLICATIONS

SECOND ORDER ALLPASS

C
OUT

R2

R1 C
IN -
+
ω0
R3
( )+ω
s2 - s Q 0
2

s + s( ) + ω
2 ω 0 2
R4 0
Q

V0
=
( ) + R1 R21 C
s2 - s
2
R2 C 2

s + s( )
VIN 2 1
2 +
R2 C R1 R2 C 2

CHOOSE: C

k = 2 π F0 C
2Q
R2 =
k
1
R1 =
2kQ
R3 = R1
Q
R4 =
2

Figure 5-81: Second order allpass

5.100
 ANALOG FILTERS
PRACTICAL PROBLEMS IN FILTER IMPLEMENTATION

SECTION 5-7: PRACTICAL PROBLEMS IN FILTER

IMPLEMENTATION

In the previous sections filters were dealt with as mathematical functions. The filter
designs were assumed to have been implemented with "perfect" components. When the
filter is built with real-world components design tradeoffs must typically be made.

In building a filter with an order greater the two, multiple second and/or first order
sections are used. The frequencies and Qs of these sections must align precisely or the
overall response of the filter will be affected. For example, the antialiasing filter design
example in the next section is a 5th order Butterworth filter, made up of a second order
section with a frequency (Fo) = 1 and a Q = 1.618, a second order section with a
frequency (Fo) = 1 and a Q = 0.618, and a first order section with a frequency (Fo) = 1
(for a filter normalized to 1 rad/sec). If the Q or frequency response of any of the sections
is off slightly, the overall response will deviate from the desired response. It may be
close, but it won't be exact. As is typically the case with engineering, a decision must be
made as to what tradeoffs should be made. For instance, do we really need a particular
response exactly? Is there a problem if there is a little more ripple in the passband? Or if
the cutoff frequency is at a slightly different frequency? These are the types of questions
that face a designer, and will vary from design to design.

Passive Components (Resistors, Capacitors, Inductors)

Passive components are the first problem. When designing filters, the calculated values of
components will most likely not available commercially. Resistors, capacitors, and
inductors come in standard values. While custom values can be ordered, the practical
tolerance will probably still be ± 1% at best. An alternative is to build the required value
out of a series and/or parallel combination of standard values. This increases the cost and
size of the filter. Not only is the cost of components increased, so are the manufacturing
costs, both for loading and tuning the filter. Furthermore, success will be still limited by
the number of parts that are used, their tolerance, and their tracking, both over
temperature and time.

A more practical way is to use a circuit analysis program to determine the response using
standard values. The program can also evaluate the effects of component drift over
temperature. The values of the sensitive components are adjusted using parallel
combinations where needed, until the response is within the desired limits. Many of the
higher end filter CAD programs include this feature.

The resonant frequency and Q of a filter are typically determined by the component
values. Obviously, if the component value is drifting, the frequency and the Q of the filter
will drift which, in turn, will cause the frequency response to vary. This is especially true
in higher order filters.

5.101
 OP AMP APPLICATIONS

Higher order implies higher Q sections. Higher Q sections means that component values
are more critical, since the Q is typically set by the ratio of two or more components,
typically capacitors.

In addition to the initial tolerance of the components, you must also evaluate effects of
temperature/time drift. The temperature coefficients of the various components may be
different in both magnitude and sign. Capacitors, especially, are difficult in that not only
do they drift, but the temperature coefficient (TC) is also a function of temperature, as
shown in Figure 5-82. This represents the temperature coefficient of a (relatively) poor
film capacitor, which might be typical for a Polyester or Polycarbonate type. Linear TC in
film capacitors can be found in the polystyrene, polypropylene, and Teflon dielectrics. In
these types TC is on the order of 100-200ppm/°C, and if necessary, this can be
compensated with a complementary TC elsewhere in the circuit.
% CAPACITANCE CHANGE

–1

–2

-55 -25 0 25 50 75 100 125

TEMPERATURE (°C)

Figure 5-82: A poor film capacitor temperature coefficient

The lowest TC dielectrics are NPO (or COG) ceramic (±30ppm/°C), and polystyrene
(–120ppm/°C). Some capacitors, mainly the plastic film types, such as polystyrene and
polypropylene, also have a limited temperature range.

While there is infinite choice of the values of the passive components for building filters,
in practice there are physical limits. Capacitor values below 10pF and above 10µF are not
practical. Electrolytic capacitors should be avoided. Electrolytic capacitors are typically
very leaky. A further potential problem is if they are operated without a polarizing
voltage, they become non-linear when the AC voltage reverse biases them. Even with a
DC polarizing voltage, the AC signal can reduce the instantaneous voltage to 0V or
below. Large values of film capacitors are physically very large.

Resistor values of less than 100Ω should be avoided, as should values over 1MΩ. Very
low resistance values (under 100Ω) can require a great deal of drive current and dissipate
a great deal of power. Both of these should be avoided. And low values and very large
values of resistors may not be as readily available. Very large values tend to be more
prone to parasitics since smaller capacitances will couple more easily into larger
impedance levels. Noise also increases with the square root of the resistor value. Larger
value resistors also will cause larger offsets due to the effects of the amplifier bias
currents.
5.102
 ANALOG FILTERS
PRACTICAL PROBLEMS IN FILTER IMPLEMENTATION

Parasitic capacitances due to circuit layout and other sources affect the performance of the
circuit. They can form between two traces on a PC board (on the same side or opposite
side of the board), between leads of adjacent components, and just about everything else
you can (and in most cases can't) think of. These capacitances are usually small, so their
effect is greater at high impedance nodes. Thus, they can be controlled most of the time
by keeping the impedance of the circuits down. Remember that the effects of stray
capacitance are frequency dependent, being worse at high frequencies because the
impedance drops with increasing frequency.
Parasitics are not just associated with outside sources. They are also present in the
components themselves.
A capacitor is more than just a capacitor in most instances. A real capacitor has
inductance (from the leads and other sources) and resistance as shown in Figure 5-83.
This resistance shows up in the specifications as leakage and poor power factor.
Obviously, we would like capacitors with very low leakage and good power factor (see
Figure 5-84).
In general, it is best to use plastic film (preferably Teflon or polystyrene) or mica
capacitors and metal film resistors, both of moderate to low values in our filters.
IDEAL CAPACITOR

MOST GENERAL MODEL OF A REAL CAPACITOR

LEAKAGE CURRENT MODEL HIGH FREQUENCY MODEL

HIGH CURRENT MODEL DIELECTRIC ABSORPTION (DA) MODEL

Figure 5-83: Capacitor equivalent circuit

One way to reduce component parasitics is to use surface mounted devices. Not having
leads means that the lead inductance is reduced. Also, being physically smaller allows
more optimal placement. A disadvantage is that not all types of capacitors are available in
surface mount. Ceramic capacitors are popular surface mount types, and of these, the
NPO family has the best characteristics for filtering. Ceramic capacitors may also be
prone to microphonics. Microphonics occurs when the capacitor turns into a motion
sensor, similar to a strain gauge, and turns vibration into an electrical signal, which is a
form of noise.

5.103
 OP AMP APPLICATIONS

CAPACITOR COMPARISON CHART

TYPE TYPICAL DA ADVANTAGES DISADVANTAGES

Polystyrene 0.001% Inexpensive Damaged by temperature > +85°C

to Low DA Large
0.02% Good stability High inductance
(~120ppm/°C) Vendors limited

Polypropylene 0.001% Inexpensive Damaged by temperature > +105°C

to Low DA Large
0.02% Stable (~200ppm/°C) High inductance
Wide range of values

Teflon 0.003% Low DA available Expensive

to Good stability Large
0.02% Operational above +125 °C High inductance
Wide range of values

Polycarbonate 0.1% Good stability Large

Low cost DA limits to 8-bit applications
Wide temperature range High inductance
Wide range of values

Polyester 0.3% Moderate stability Large

to Low cost DA limits to 8-bit applications
0.5% Wide temperature range High inductance (conventional)
Low inductance (stacked
film)

NP0 Ceramic <0.1% Small case size DA generally low (may not be
Inexpensive, many vendors specified)
Good stability (30ppm/°C) Low maximum values (≤ 10nF)
1% values available
Low inductance (chip)

Monolithic >0.2% Low inductance (chip) Poor stability

Ceramic Wide range of values Poor DA
(High K) High voltage coefficient

Mica >0.003% Low loss at HF Quite large

Low inductance Low maximum values (≤ 10nF)
Good stability Expensive
1% values available

Aluminum Very high Large values High leakage

Electrolytic High currents Usually polarized
High voltages Poor stability, accuracy
Small size Inductive

Tantalum Very high Small size High leakage

Electrolytic Large values Usually polarized
Medium inductance Expensive
Poor stability, accuracy
Figure 5-84: Capacitor comparison chart

5.104
 ANALOG FILTERS
PRACTICAL PROBLEMS IN FILTER IMPLEMENTATION

RESISTOR COMPARISON CHART

TYPE ADVANTAGES DISADVANTAGES

DISCRETE Carbon Lowest Cost Poor Tolerance (5%)

Composition High Power/Small Case Size Poor Temperature Coefficient
Wide Range of Values (1500 ppm/°C)

Wirewound Excellent Tolerance (0.01%) Reactance is a Problem

Excellent TC (1ppm/°C) Large Case Size
High Power Most Expensive

Metal Film Good Tolerance (0.1%) Must be Stabilized with Burn-In

Good TC (<1 to 100ppm/°C) Low Power
Moderate Cost
Wide Range of Values
Low Voltage Coefficient

Bulk Metal or Excellent Tolerance (to 0.005%) Low Power

Metal Foil Excellent TC (to <1ppm/°C) Very Expensive
Low Reactance
Low Voltage Coefficient

High Megohm Very High Values (108 to 1014Ω) High Voltage Coefficient
Only Choice for Some Circuits (200ppm/V)
Fragile Glass Case (Needs Special
Handling)
Expensive

NETWORKS Thick Film Low Cost Fair Matching (0.1%)

High Power Poor TC (>100ppm/°C)
Laser-Trimmable Poor Tracking TC (10ppm/°C)
Readily Available

Thin Film Good Matching (<0.01%) Often Large Geometry

Good TC (<100ppm/°C) Limited Values and
Good Tracking TC (2ppm/°C) Configurations
Moderate Cost
Laser-Trimmable
Low Capacitance
Suitable for Hybrid IC Substrate

Figure 5-85: Resistor comparison chart

Resistors also have parasitic inductances due to leads and parasitic capacitance. The
various qualities of resistors are compared in Figure 5-85.

5.105
 OP AMP APPLICATIONS

Limitations of Active Elements (Op Amps) in Filters

The active element of the filter will also have a pronounced effect on the response.
In developing the various topologies (Multiple Feedback, Sallen-Key, State Variable,
etc.), the active element was always modeled as a "perfect" operational amplifier. That is
to say it has:
1) infinite gain
2) infinite input impedance
3) zero output impedance

none of which vary with frequency. While amplifiers have improved a great deal over the
years, this model has not yet been realized.

The most important limitation of the amplifier has to due with its gain variation with
frequency. All amplifiers are band limited. This is due mainly to the physical limitations
of the devices with which the amplifier is constructed. Negative feedback theory tells us
that the response of an amplifier must be first order (– 6dB per octave) when the gain falls
to unity in order to be stable. To accomplish this, a real pole is usually introduced in the
amplifier so the gain rolls off to <1 by the time the phase shift reaches 180° (plus some
phase margin, hopefully). This roll off is equivalent to that of a single pole filter. So in
simplistic terms, the transfer function of the amplifier is added to the transfer function of
the filter to give a composite function. How much the frequency dependent nature of the
op amp affects the filter is dependent on which topology is used as well as the ratio of the
filter frequency to the amplifier bandwidth.

The Sallen-Key configuration, for instance, is the least dependent on the frequency
response of the amplifier. All that is required is for the amplifier response to be flat to just
past the frequency where the attenuation of the filter is below the minimum attenuation
required. This is because the amplifier is used as a gain block. Beyond cutoff, the
attenuation of the filter is reduced by the rolloff of the gain of the op amp. This is because
the output of the amplifier is phase shifted, which results in incomplete nulling when fed
back to the input. There is also an issue with the output impedance of the amplifier rising
with frequency as the open loop gain rolls off. This causes the filter to lose attenuation.

The state variable configuration uses the op amps in two modes, as amplifiers and as
integrators. As amplifiers, the constraint on frequency response is basically the same as
for the Sallen-Key, which is flat out to the minimum attenuation frequency. As an
integrator, however, more is required. A good rule of thumb is that the open loop gain of
the amplifier must be greater than 10 times the closed loop gain (including peaking from
the Q of the circuit). This should be taken as the absolute minimum requirement. What
this means is that there must be 20dB loop gain, minimum. Therefore, an op amp with
10MHz unity gain bandwidth is the minimum required to make a 1MHz integrator. What
happens is that the effective Q of the circuit increases as loop gain decreases. This
phenomenon is called Q enhancement. The mechanism for Q enhancement is similar to
that of slew rate limitation. Without sufficient loop gain, the op amp virtual ground is no
longer at ground. In other words, the op amp is no longer behaving as an op amp. Because
of this, the integrator no longer behaves like an integrator.
5.106
 ANALOG FILTERS
PRACTICAL PROBLEMS IN FILTER IMPLEMENTATION

The multiple feedback configuration also places heavy constraints on the active element.
Q enhancement is a problem in this topology as well. As the loop gain falls, the Q of the
circuit increases, and the parameters of the filter change. The same rule of thumb as used
for the integrator also applies to the multiple feedback topology (loop gain should be at
least 20dB). The filter gain must also be factored into this equation.

In the FDNR realization, the requirements for the op amps are not as clear. To make the
circuit work, we assume that the op amps will be able to force the input terminals to be
the same voltage. This implies that the loop gain be a minimum of 20dB at the resonant
frequency.

Also it is generally considered to be advantageous to have the two op amps in each leg
matched. This is easily accomplished using dual op amps. It is also a good idea to have
low bias current devices for the op amps, so FET input op amps should be used, all other
things being equal.

In addition to the frequency dependent limitations of the op amp, several others of its
parameters may be important to the filter designer.

One is input impedance. We assume in the "perfect" model that the input impedance is
infinite. This is required so that the input of the op amp does not load the network around
it. This means that we probably want to use FET amplifiers with high impedance circuits.

There is also a small frequency dependent term to the input impedance, since the effective
impedance is the real input impedance multiplied by the loop gain. This usually is not a
major source of error, since the network impedance of a high frequency filter should be
low.

Distortion resulting from Input Capacitance Modulation

Another subtle effect can be noticed with FET input amps. The input capacitance of a
FET changes with the applied voltage. When the amplifier is used in the inverting
configuration, such as with the multiple feedback configuration, the applied voltage is
held to 0V. Therefore there is no capacitance modulation. However, when the amplifier is
used in the non-inverting configuration, such as in the Sallen-Key circuit, this form of
distortion can exist.

There are two ways to address this issue. The first is to keep the equivalent impedance
low. The second is to balance the impedance seen by the inputs. This is accomplished by
adding a network into the feedback leg of the amplifier which is equal to the equivalent
input impedance. Note that this will only work for a unity gain application.

5.107
 OP AMP APPLICATIONS

As an example, which is taken from the OP176 data sheet, a 1kHz highpass Sallen-Key
filter is shown (Figure 5-86). Figure 5-87 shows the distortion for the uncompensated
version (curve A1) as well as with the compensation (curve A2). Also shown is the same
circuit with the impedances scaled up by a factor of 10 (B1 uncompensated, B2
compensated). Note that the compensation improves the distortion, but not as much as
having low impedance to start with.
R1
IN OUT
C1 11kΩ
0.01µF
C2 +VS
0.01µF
+
R2 OP176
22kΩ
–
–VS

ZCOMP

ZCOMP R2

(HIGHPASS) C2 C1

R1

Figure 5-86: Compensation for input capacitance voltage modulation

Figure 5-87: Distortion due to input capacitance modulation

Similarly, the op amp output impedance affects the response of the filter. The output
impedance of the amplifier is divided by the loop gain, therefore the output impedance
will rise with increasing frequency. This may have an effect with high frequency filters if
the output impedance of the stage driving the filter becomes a significant portion of the
network impedance.

5.108
 ANALOG FILTERS
PRACTICAL PROBLEMS IN FILTER IMPLEMENTATION

The fall of loop gain with frequency can also affect the distortion of the op amp, since
there is less loop gain available for correction. In the multiple feedback configuration the
feedback loop is also frequency dependent, which may further reduce the feedback
correction, resulting in increased distortion. This effect is counteracted somewhat by the
reduction of distortion components in the filter network (assuming a lowpass or bandpass
filter).

All of the discussion so far is based on using classical voltage feedback op amps. Current
feedback, or transimpedance, op amps offer improved high frequency response, but are
unusable in any topologies discussed except the Sallen-Key. The problem is that
capacitance in the feedback loop of a current feedback amplifier usually causes it to
become unstable. Also, most current feedback amplifiers will only drive a small
capacitive load. Therefore, it is difficult to build classical integrators using current
feedback amplifiers. Some current feedback op amps have an external pin that may be
used to configure them as a very good integrator, but this configuration does not lend
itself to classical active filter designs.

Current feedback integrators tend to be non-inverting, which is not acceptable in the state
variable configuration. Also, the bandwidth of a current feedback amplifier is set by its
feedback resistor, which would make the Multiple Feedback topology difficult to
implement. Another limitation of the current feedback amplifier in the Multiple Feedback
configuration is the low input impedance of the inverting terminal. This would result in
loading of the filter network. Sallen-Key filters are possible with current feedback
amplifiers, since the amplifier is used as a non-inverting gain block. New topologies that
capitalize on the current feedback amplifiers superior high frequency performance and
compensate for its limitations will have to be developed.

The last thing that you need to be aware of is exceeding the dynamic range of the
amplifier. Qs over 0.707 will cause peaking in the response of the filter (see Figures 5-5
through 5-7). For high Q's, this could cause overload of the input or output stages of the
amplifier with a large input. Note that relatively small values of Q can cause significant
peaking. The Q times the gain of the circuit must stay under the loop gain (plus some
margin, again, 20dB is a good starting point). This holds for multiple amplifier topologies
as well. Be aware of internal node levels, as well as input and output levels. As an
amplifier overloads, its effective Q decreases, so the transfer function will appear to
change even if the output appears undistorted. This shows up as the transfer function
changing with increasing input level.

We have been dealing mostly with lowpass filters in our discussions, but the same
principles are valid for highpass, bandpass, and band reject as well. In general, things like
Q enhancement and limited gain/bandwidth will not affect highpass filters, since the
resonant frequency will hopefully be low in relation to the cutoff frequency of the op
amp. Remember, though, that the highpass filter will have a low pass section, by default,
at the cutoff frequency of the amplifier. Bandpass and band reject (notch) filters will be
affected, especially since both tend to have high values of Q.

5.109
 OP AMP APPLICATIONS

The general effect of the op amp's frequency response on the filter Q is shown in Figure
5-88.

dB A(s)
Q ACTUAL

Q THEORETICAL

f
Figure 5-88: Q enhancement
As an example of the Q enhancement phenomenon, consider the Spice simulation of a
10kHz bandpass Multiple Feedback filter with Q = 10 and gain = 1, using a good high
frequency amplifier (the AD847) as the active device. The circuit diagram is shown in
Figure 5-89. The open loop gain of the AD847 is greater than 70dB at 10kHz as shown in
Figure 5-91(A). This is well over the 20dB minimum, so the filter works as designed as
shown in Figure 5-90.

We now replace the AD847 with an OP-90. The OP-90 is a DC precision amplifier and
so has a limited bandwidth. In fact, its open loop gain is less than 10dB at 10kHz (see
Figure 5-91(B)). This is not to imply that the AD847 is in all cases better than the OP-90.
It is a case of misapplying the OP-90.

1000pF
316kΩ
IN +VS

–
15.8Ω 1000pF
OUT
AD847
+
(OP-90)
–VS

Q = 10
H = 1
F0 = 10kHz

Figure 5-89: 1kHz multiple feedback bandpass filter

From the output for the OP-90, also shown in Figure 5-90, we see that the magnitude of
the output has been reduced, and the center frequency has shifted downward.

5.110
 ANALOG FILTERS
PRACTICAL PROBLEMS IN FILTER IMPLEMENTATION

+20

+10

0
GAIN
(dB)
–10

–20

–30

–40
0.1 1.0 10 100 1000

FREQUENCY (kHz)

Figure 5-90: Effects of "Q enhancement"

Figure 5-91: AD847 and OP-90 Bode plots

5.111
 OP AMP APPLICATIONS

NOTES:

5.112
 ANALOG FILTERS
DESIGN EXAMPLES

SECTION 5-8: DESIGN EXAMPLES

Several examples will now be worked out to demonstrate the concepts previously
discussed

Antialias Filter

As an example, passive and active antialiasing filters will now be designed based upon a
common set of specifications. The active filter will be designed in four ways: Sallen-
Key, Multiple Feedback, State Variable, and Frequency Dependent Negative Resistance
(FDNR).

The specifications for the filter are given as follows:

1) The cutoff frequency will be 8kHz.

2) The stopband attenuation will be 72dB. This corresponds to a 12 bit
system.
3) Nyquist frequency of 50kSPS.
4) The Butterworth filter response is chosen in order to give the best
compromise between attenuation and phase response.

6.25
RESPONSE (dB)

–50
4
5
– 72 dB

–90
0.1 0.2 0.4 0.8 1.1 2.0 4.0 8.0 10

FREQUENCY (Hz)

Figure 5-92: Determining filter order

Consulting the Butterworth response curves (Figure 5-14, reproduced above in
Figure 5-92), we see that for a frequency ratio of 6.25 (50kSPS / 8kSPS), that a filter
order of 5 is required.

5.113
 OP AMP APPLICATIONS

Now consulting the Butterworth design table (Figure 5-25), the normalized poles of a 5th
order Butterworth filter are:

STAGE Fo α .
1 1.000 1.618
2 1.000 0.618
3 1.000 ------

The last stage is a real (single) pole, thus the lack of an alpha value. It should be noted
that this is not necessarily the order of implementation in hardware. In general, you would
typically put the real pole last and put the second order sections in order of decreasing
alpha (increasing Q) as we have done here. This will avoid peaking due to high Q
sections possibly overloading internal nodes. Another feature of putting the single pole at
the end is to bandlimit the noise of the op amps. This is especially true if the single pole is
implemented as a passive filter.

For the passive design, we will choose the zero input impedance configuration. While
"classic" passive filters are typically double terminated, that is with termination on both
source and load ends, we are concerned with voltage transfer not power transfer so the
source termination will not be used. From the design table (see Reference 2, p. 313), we
find the normalized values for the filter (see Figure 5-93).

IN 1.5451H 1.3820H 0.3090H OUT

1.6844F 0.8944F 1Ω

Figure 5-93: Normalized passive filter implementation

These values are normalized for a 1 rad/s filter with a 1Ω termination. To scale the filter
we divide all reactive elements by the desired cutoff frequency, 8kHz (= 50265 rad/sec, =
2π 8×103). This is commonly referred to as the frequency scale factor (FSF). We also
need to scale the impedance.

For this example, an arbitrary value of 1000Ω is chosen. To scale the impedance, we
multiply all resistor and inductor values and divide all capacitor values by this magnitude,
which is commonly referred to as the impedance scaling factor (Z).

5.114
 ANALOG FILTERS
DESIGN EXAMPLES

After scaling, the circuit looks like Figure 5-94.

IN 30.7mH 27.5mH 6.15mH OUT

1kΩ
0.033µF 0.018µF

Figure 5-94: Passive filter implementation

For the Sallen-Key active filter, we use the design equations shown in Figure 5-49. The
values for C1 in each section are arbitrarily chosen to give reasonable resistor values. The
implementation is shown in Figure 5-95.
IN 2.49kΩ 0.01µF 6.49kΩ 0.01µF 2kΩ OUT
2kΩ

0.01µF
2.49kΩ 6.49kΩ

+ +
-
- -
0.0062µF 910pF
+

Figure 5-95: Sallen-Key implementation

The exact values have been rounded to the nearest standard value. For most active
realization to work correctly, it is required to have a zero-impedance driver, and a return
path for DC due to the bias current of the op amp. Both of these criteria are
approximately met when you use an op amp to drive the filter.

In the above example the single pole has been built as an active circuit. It would have
been just as correct to configure it as a passive RC filter. The advantage to the active
section is lower output impedance, which may be an advantage in some applications,
notably driving an ADC input that uses a switched capacitor structure.

This type of input is common on sigma delta ADCs as well as many other CMOS type of
converters. It also eliminates the loading effects of the input impedance of the following
stage on the passive section.

5.115
 OP AMP APPLICATIONS

Figure 5-96 shows a multiple feedback realization of our filter. It was designed using the
equations in Figure 5-52. In this case, the last section is a passive RC circuit.

OUT

2kΩ
0.01µF
1.62kΩ 1.24kΩ 0.005µF
0.01µF
IN
1.62kΩ 806Ω -
1.24kΩ 619Ω -

+ +

0.03µF
0.1µF

Figure 5-96: Multiple feedback implementation

An optional buffer could be added after the passive section, if desired. This would give
many of the advantages outlined above, except for bandlimiting the noise of the output
amp. By using one of the above two filter realizations, we have both an inverting and a
non-inverting design.

The state variable filter, shown in Figure 5-97, was designed with the equations in Figure
5-55. Again, we have rounded the resistor values to the nearest standard 1% value.
10kΩ 10kΩ

IN
10kΩ 20kΩ 0.01µF 20kΩ 0.01µF
- - -

+ + +

8.45kΩ
10kΩ

10kΩ OUT
+
10kΩ 2kΩ
-
10kΩ 20kΩ 0.01µF 20kΩ 0.01µF
- - - 0.01µF

+ + +

38.3kΩ

10kΩ

Figure 5-97: State variable implementation

Obviously this filter implementation has many more parts than either the Sallen-Key or
the multiple feedback. The rational for using this circuit is that stability is improved and
the individual parameters are independently adjustable.

5.116
 ANALOG FILTERS
DESIGN EXAMPLES

The Frequency Dependent Negative Resistance (FDNR) realization of this filter is shown
in Figure 5-98.

IN 3.09kΩ 2.74kΩ 619Ω OUT

0.01µF 0.01µF
0.01µF

2kΩ 2kΩ
+ +

- - - -

+ +
2kΩ 2kΩ

3.4kΩ 1.78kΩ

0.01µF 0.01µF

Figure 5.98: FDNR Implementation

In the conversion process from passive to FDNR, the D element is normalized for a
capacitance of 1F. We then scale the filter to a more reasonable value (0.01µF in this
case).

In all of the above implementations standard values were used instead of the calculated
values. Any variation from the ideal values will cause a shift in the filter response
characteristic, but often the effects are minimal. The computer can be used to evaluate
these variations on the overall performance and determine if they are acceptable.

To examine the effect of using standard values, take the Sallen-Key implementation.
Figure 5-99 shows the response of each of the 3 sections of the filter. While the Sallen-
Key was the filter used, the results from any of the other implementations will give
similar results.

Figure 5-100 then shows the effect of using standard values instead of calculated values.
Notice that the general shape of the filter remains the same, just slightly shifted in
frequency. This investigation was done only for the standard value of the resistors. To
understand the total effect of component tolerance the same type of calculations would
have to be done for the tolerance of all the components and also for their temperature and
aging effects.

In active filter applications using op amps, the dc accuracy of the amplifier is often
critical to optimal filter performance. The amplifier's offset voltage will be passed by the
lowpass filter and may be amplified to produce excessive output offset. For low
frequency applications requiring large value resistors, bias currents flowing through these
resistors will also generate an output offset voltage.

5.117
 OP AMP APPLICATIONS

In addition, at higher frequencies, an op amp's dynamics must be carefully considered.

Here, slewrate, bandwidth, and open loop gain play a major role in op amp selection. The
slewrate must be fast as well as symmetrical to minimize distortion.
5

α = 0.618
0

SINGLE POLE

–10
RESPONSE (dB)

α = 1.618

TOTAL FILTER

–20

–30
2.0 3.0 10 20
FREQUENCY (kHz)

Figure 5-99: Individual section response

5
A

B
0

E
D
RESPONSE (dB)

F
C
–10

H
G

–20

–30
2.0 3.0 FREQUENCY (kHz) 10 20

A= α = 0.618 B= α = 0.618 C= α = 1.618 D= α = 1.618

REAL VALUES CALC. VALUES REAL VALUES CALC. VALUES
SINGLE POLE SINGLE POLE TOTAL FILTER TOTAL FILTER
E= F= G= H=
REAL VALUES CALC. VALUES REAL VALUES CALC. VALUES

Figure 5-100: Effect of using standard value resistors

5.118
 ANALOG FILTERS
DESIGN EXAMPLES

Transformations

In the next example the transformation process will be investigated:

As mentioned earlier, filter theory is based on a low pass prototype, which is then
manipulated into the other forms. In these examples the prototype that will be used is a
1kHz, 3 pole, 0.5dB Chebyshev filter. A Chebyshev was chosen because it would show
more clearly if the responses were not correct, a Butterworth would probably be too
forgiving in this instance. A 3 pole filter was chosen so that a pole pair and a single pole
would be transformed.

The pole locations for the LP prototype were taken from Figure 5-30. They are:

STAGE α β FO α
.

1 0.2683 0.8753 1.0688 0.5861

2 0.5366 0.6265

The first stage is the pole pair and the second stage is the single pole. Note the
unfortunate convention of using α for 2 entirely separate parameters. The α and β on the
left are the pole locations in the s-plane. These are the values that are used in the
transformation algorithms. The α on the right is 1/Q, which is what the design equations
for the physical filters want to see.

The Sallen-Key topology will be used to build the filter. The design equations in Figure
5-67 (pole pair) and Figure 5-66 (single pole) where then used to design the filter. The
schematic is shown in Figure 5-101.
0.1µF
5.08kΩ 2.54kΩ
IN OUT

5.08kΩ
0.1µF

-
8.59nF

Figure 5-101: Lowpass prototype

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Using the equation string described in Section 5, the filter is now transformed into a
highpass filter. The results of the transformation are:

STAGE α β FO α
.
1 0.3201 1.0443 0.9356 0.5861
2 1.8636 1.596

A word of caution is warranted here. Since the convention of describing a Chebyshev

filter is to quote the end of the error band instead of the 3dB frequency, the F0 must be
divided (for highpass) by the ratio of ripple band to 3dB bandwidth (Table 1, Section 4).

The Sallen-Key topology will again be used to build the filter. The design equations in
Figure 5-68 (pole pair) and Figure 5-66 (single pole) where then used to design the filter.
The schematic is shown in Figure 5-102.
0.01µF 0.01µF
4.99kΩ
IN OUT

0.01µF
9.97kΩ

58kΩ

Figure 5-102: Highpass transformation

Figure 5-103 shows the response of the lowpass prototype and the highpass
transformation. Note that they are symmetric around the cutoff frequency of 1kHz. Also
note that the errorband is at 1kHz, not the −3dB point, which is characteristic of
Chebyshev filters.

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DESIGN EXAMPLES
5

RESPONSE (dB) –5

–10

–15

–20

–25
0.1 0.3 1.0 3.0 10

FREQUENCY (kHz)

Figure 5-103: Lowpass and highpass response

The lowpass prototype is now converted to a bandpass filter. The equation string outlined
in Section 5-5 is used for the transformation. Each pole of the prototype filter will
transform into a pole pair. Therefore the 3 pole prototype, when transformed, will have 6
poles (3 pole pairs). In addition, there will be 6 zeros at the origin.

Part of the transformation process is to specify the 3 dB bandwidth of the resultant filter.
In this case this bandwidth will be set to 500 Hz. The results of the transformation yield:

STAGE F0 Q A0
.
1 804.5 7.63 3.49
2 1243 7.63 3.49
3 1000 3.73 1

The reason for the gain requirement for the first 2 stages is that their center frequencies
will be attenuated relative to the center frequency of the total filter. Since the resultant Q's
are moderate (less than 20) the Multiple Feedback topology will be chosen. Figure 5-72
was then used to design the filter sections.

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Figure 5-104 is the schematic of the filter and Figure 5-105 shows the filter response.
OUT

0.01µF 0.01µF
301kΩ 196kΩ 118kΩ
0.01µF
IN 43.2kΩ 28kΩ 59kΩ
- - -

0.01µF
+ + +
0.01µF 0.01µF
1.33kΩ 866Ω 2.21kΩ

Figure 5-104: Bandpass transformation

–5
RESPONSE (dB)

–10

–15

–20
0.3 1.0 3.0
FREQUENCY (kHz)

Figure 5-105: Bandpass filter response

Note that again there is symmetry around the center frequency. Also the 500Hz
bandwidth is not 250Hz either side of the center frequency (arithmetic symmetry). Instead
the symmetry is geometric, which means that for any 2 frequencies (F1 & F2) of equal
amplitude are related by:
F0 = √ F1* F2

Lastly the prototype will be transformed into a bandreject filter. For this the equation
string in Section 5-5 is used. Again, each pole of the prototype filter will transform into a
pole pair. Therefore, the 3 pole prototype, when transformed, will have 6 poles (3 pole
pairs).

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DESIGN EXAMPLES

As in the bandpass case, part of the transformation process is to specify the 3dB
bandwidth of the resultant filter. Again in this case this bandwidth will be set to 500Hz.
The results of the transformation yield:

STAGE F0 Q F0Z
.
1 763.7 6.54 1000
2 1309 6.54 1000
3 1000 1.07 1000

Note that there are 3 cases of notch filters required. There is a standard notch (F0 = FZ,
section 3), a lowpass notch (F0 < FZ, section 1) and a highpass notch (F0 > FZ, section 2).
Since there is a requirement for all 3 types of notches, the Bainter Notch is used to build
the filter. The filter is designed using Figure 5-77. The gain factors K1 & K2 are
arbitrarily set to 1. Figure 5-106 is the schematic of the filter.

1.58kΩ 931Ω

IN 0.01µF 0.01µF
10kΩ 10kΩ 10kΩ 10kΩ

- -
- - 274kΩ - -
+ +
+ + + +
931Ω 1.58kΩ 158kΩ
0.01µF 0.01µF

274kΩ 158kΩ

1.21kΩ
10kΩ 10kΩ

0.01µF -
- -
+
OUT
1.21kΩ
+ +
210kΩ

0.01µF
210kΩ

Figure 5-106: Bandreject transformation

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The response of the filter is shown in Figure 5-107 and in detail in Figure 5-108. Again,
note the symmetry around the center frequency. Again the frequencies have geometric
symmetry.

–20
RESPONSE (dB)

–40

–60

–80

–100
0.1 0.3 1.0 3.0 10

FREQUENCY (kHz)

Figure 5-107: Bandreject response

0
RESPONSE (dB)

–5

–10

–15

–20
0.1 0.3 1.0 3.0 10

FREQUENCY (kHz)

Figure 5-108: Bandreject response (detail)

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DESIGN EXAMPLES

CD Reconstruction Filter

This design was done for a magazine article describing a high quality outboard D/A
converter for use with digital audio sources (se Reference 26).

A reconstruction filter is required on the output of a D/A converter because, despite the
name, the output of a D/A converter is not really an analog voltage but instead a series of
steps. The converter will put out a discrete voltage, which it will then hold until the next
sample is asserted. The filter's job is to remove the high frequency components,
smoothing out the waveform. This is why the filter is sometimes referred to as a
smoothing filter. This also serves to eliminate the aliases of the conversion process. The
"standard" in the audio industry is to use a 3rd order Bessel function as the reconstruction
filter. The reason to use a Bessel filter is that it has the best phase response. This helps to
preserve the phase relationship of the individual tones in the music. The price for this
phase "goodness" is that the amplitude discrimination is not as good as some other filter
types. If we assume that we are using 8× oversampling of the 48kSPS data stream in the
D/A converter then the aliases will appear at 364kHz (8 × 48k – 20k). The digital filter
that is used in the interpolation process will eliminate the frequencies between 20kHz and
364kHz. If we assume that the bandedge is 30kHz, then we have a frequency ratio of
approximately 12 (364 ÷ 30). We use 30kHz as the band edge, rather than 20kHz to
minimize the rolloff due to the filter in the passband. In fact, the complete design for this
filter includes a shelving filter to compensate for the passband rolloff. Extrapolating from
Figure 5-20, a 3rd order Bessel will only provide on the order of 55dB attenuation at 12 ×
Fo. This is only about 9 bit accuracy.

By designing the filter as 7th order, and by designing it as a linear phase with equiripple
error of 0.05°, we can increase the stopband attenuation to about 120dB at 12 × Fo. This
is close to the 20 bit system that we are hoping for.

The filter will be designed as a FDNR type. This is an arbitrary decision. Reasons to
choose this topology are it’s low sensitivities to component tolerances and the fact that
the op amps are in the shunt arms rather than in the direct signal path.

The first step is to find the passive prototype. To do this, use the charts in Williams's book.
We then get the circuit shown in Figure 5-109A. Next perform a translation in the s-plane.
This gives the circuit shown in Figure 5-109B. This filter is scaled for a frequency of 1Hz.
and an impedance level of 1Ω. The D structure of the converted filter is replaced by a GIC
structure that can be physically realized. The filter is then denormalize by frequency
(30kHz) and impedance (arbitrarily chosen to be 1kΩ). This gives a frequency-scaling
factor (FS) of 1.884 ×105 (= 2π (3 ×104)). Next arbitrarily choose a value of 1nF for the
capacitor. This gives an impedance-scaling factor (Z) of 5305 (= (COLD/CNEW )/ FSF).

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IN 1.4988 0.8422 0.6441 0.1911 OUT

1
1.0071 0.7421 0.4791

Figure 5-109A: CD Reconstruction filter – passive prototype

IN 1.4988 0.8422 0.6441 0.1911 OUT

1.0071 0.7421 0.4791 1

Figure 5-109B: CD Reconstruction filter – transformation in s-plane

IN 1.4988 0.8422 0.6441 0.1911 OUT

1 1 1
+ + +

- - -

1 1 1 1

1
- - -

+
1 + +
1

1.0071 0.7421 0.4791

1 1 1

Figure 5-109C: CD Reconstruction filter – normalized FDNR

IN 13kΩ||21kΩ 4.87kΩ||61.9kΩ 4.42kΩ||15.8kΩ 1.24kΩ||5.90kΩ OUT

5.36kΩ AD712 5.36kΩ AD712 5.36kΩ

AD712
+ + +

- - -

5.36kΩ 5.36kΩ 5.36kΩ

1nF

1nF 1nF
- - -

+ + +
1nF

AD712 AD712 AD712

5.62kΩ||137kΩ 5.36kΩ||15.8kΩ 3.16kΩ||13.7kΩ

1nF 1nF 1nF

Figure 5-109D: CD Reconstruction filter – final filter

5.126
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DESIGN EXAMPLES

Then multiply the resistor values by Z. This results in the resistors that had the
normalized value of 1Ω will now have a value of 5.305kΩ. For the sake of simplicity
adopt the standard value of 5.36kΩ. Working backwards, this will cause the cutoff
frequency to change to 29.693kHz. This slight shift of the cutoff frequency will be
acceptable.

The frequency scaling factor is then recalculated with the new center frequency and this
value is used to denormalize the rest of the resistors. The design flow is illustrated in
Figure 5-109. The final schematic is shown it Figure 5-109D.

The performance of the filter is shown in Figure 5-110(A-D).

(A) FREQUENCY RESPONSE (B) LINEARITY

(C) SIGNAL TO NOISE RATIO (D) THD + N

Figure 5-110: CD filter performance

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Digitally Programable State Variable Filter

One of the attractive features of the state variable filter is that the parameters (gain, cutoff
frequency and "Q") can be individually adjusted. This attribute can be exploited to allow
control of these parameters.

To start, the filter is reconfigured slightly. The resistor divider that determines Q (R6 &
R7 of Figure 5-54) is changed to an inverting configuration. The new filter schematic is
shown in Figure 5-111. Then the resistors R1, R2, R3 & R4 (of Figure 5-111) are
replaced by CMOS multiplying DACs. Note that R5 is implemented as the feedback
resistor implemented in the DAC. The schematic of this circuit is shown in Figure 5-112.
R7

IN R1 R5 C1 C2
R8 R3 R4

- R6 - - - LP OUT
+ + + +

AD825 AD825 AD825 AD825

BP OUT

R2

Figure 5-111: Redrawn state variable filter

The AD7528 is an 8 bit dual MDAC. The AD825 is a high speed FET input opamp.
Using these components the frequency range can be varied from around 550Hz to around
150kHz (Figure 5-113). The Q can be varied from approximately 0.5 to over 12.5 (Figure
5-114). The gain of circuit can be varied from 0dB to –48dB (Figure 5-115).

The operation of the DACs in controlling the parameters can be best thought of as the
DACs changing the effective resistance of the resistors. This relationship is:

DAC EQUIVALENT RESISTANCE 256 * DAC RESISTANCE

DAC CODE (DECIMAL)

This, in effect, varies the resistance from 11kΩ to 2.8MΩ for the AD7528.

5.128
 ANALOG FILTERS
DESIGN EXAMPLES
IN R1 R2
LP OUT
VIN RREF R3
DAC A
HP OUT

LEVEL CONTROL
CS
WR IOUT
AD7528 – –

D0 … D7
+ +
AD825
AD825
...

R5 R6
C2
1kΩ
1kΩ
...

C1
Q CONTROL

DAC A VIN RREF DAC A VIN RREF

D0 … D7
CS CS
WR AD7528 IOUT WR AD7528 IOUT
DAC A IOUT – –
AD7528
CS
+ +
WR D0 … D7 D0 … D7
VIN RREF AD825 AD825
... ...

FREQUENCY CONTROL
BP OUT

Figure 5-112: Digitally controlled state variable filter

–20
AMPLITUDE (dB)

–40

–60

–80
0.1 0.3 1.0 3.0 10 30 100 300 1k
FREQUENCY (kHz)

Figure 5-113: Frequency response vs. DAC control word

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50

40

30
AMPLITUDE (dB)

20

10

–10

–20
0.1 0.3 1.0 3.0 10
FREQUENCY (kHz)

Figure 5-114: Q Variation vs. DAC control word

10

–10
AMPLITUDE (dB)

–20

–40

–60

–80
0.1 0.3 1.0 3.0 10
FREQUENCY (kHz)

Figure 5-115: Gain variation vs. DAC control word

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DESIGN EXAMPLES

One limitation of this design is that the frequency is dependent on the ladder resistance of
the DAC. This particular parameter is not controlled. DACs are trimmed so that the ratios
of the resistors, not their absolute values, are controlled. In the case of the AD7528, the
typical value is 11kΩ. It is specified as 8kΩ min. and 15kΩ max. A simple modification
of the circuit can eliminate this issue. The cost is 2 more op amps (Figure 5-116). In this
case, the effective resistor value is set by the fixed resistors rather than the DAC's
resistance. Since there are 2 integrators the extra inversions caused by the added op amps
cancel.
IN
R C OUT

VIN RREF
DAC A
CS
WR AD7528-4 IOUT
– –
CONTROL
BUS AD825 AD825
D0 … D7
+ +
...

Figure 5-116: Improved digitally variable integrator

As a side note, the multiplying DACs could be replaced by analog multipliers. In this case
the control would obviously be an analog rather than a digital signal. We also could just
as easily have used a digital pot in place of the MDACs. The difference is that instead of
increasing the effective resistance, the value of the pot would be the maximum.

60 Hz Notch Filter

A very common problem in instrumentation is that of interference of the telemetry that is

to be measured. One of the primary sources of this interference is the power line. This is
particularly true of high impedance circuits. Another path for this noise is ground loops.
One possible solution is to use a notch filter to remove the 60Hz. component. Since this is
a single frequency interference, the Twin-T circuit will be used.

Since the maximum attenuation is desired and the minimum notch width is desired, the
maximum Q of the circuit is desired. This means the maximum amount of positive
feedback is used (R5 open and R4 shorted). Due to the high impedance of the network, a
FET input op amp is used.

The filter is designed using Figure 5-78. The schematic is shown in Figure 5-117 and the
response in Figure 5-118.

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IN 26.5kΩ 26.5kΩ
OUT
+

-
0.2µF AD712

13.5Ω
13.5kΩ

0.1µF 0.1µF

Figure 5-117: 60 Hz Twin-T notch filter

–10
AMPLITUDE (dB)

–20

–30

–40

–50
50 60 70
FREQUENCY (Hz)

Figure 5-118: 60 Hz notch response

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CHAPTER REFERENCES

CHAPTER REFERENCES:
1. A. I. Zverev, Handbook of Filter Synthesis, John Wiley, 1967.

2. A. B. Williams, Electronic Filter Design Handbook, McGraw-Hill, 1981, ISBN: 0-07-070430-9.

3. M. E. Van Valkenburg, Analog Filter Design, Holt, Rinehart & Winston, 1982

4. M. E. Van Valkenburg, Introduction to Modern Network Synthesis, John Wiley and Sons, 1960.

5. Zverev and H. J. Blinchikoff, Filtering in the Time and Frequency Domain, John Wiley and Sons,
1976.

6. S. Franco, Design with Operational Amplifiers and Analog Integrated Circuits, McGraw-Hill
1988, ISBN: 0-07-021799-8.

7. W. Cauer, Synthesis of Linear Communications Networks, McGraw-Hill, New York, 1958.

8. Budak, Passive and Active Network Analysis and Synthesis, Houghton Mifflin Company, Boston,
1974.

9. L. P. Huelsman and P. E. Allen, Introduction to the Theory and Design of Active Filters, McGraw
Hill, 1980, ISBN: 0-07-030854-3.

10. R. W. Daniels, Approximation Methods for Electronic Filter Design, McGraw-Hill, New York,
1974.

11. J. Tow, "Active RC Filters – a State-space Realization," Proc. IEEE, 1968, vol.56, pp. 1137-1139.

12. L.C. Thomas, "The Biquad: Part I – Some Practical Design Considerations,"
IEEE Trans. Circuits and Systems, Vol. CAS-18, 1971, pp. 350-357.

13. L.C. Thomas, "The Biquad: Part I – A Multipurpose Active Filtering System,"
IEEE Trans. Circuits and Systems, Vol. CAS-18, 1971, pp. 358-361.

14. R. P. Sallen and E. L. Key, "A Practical Method of Designing RC Active Filters,"
IRE Trans. Circuit Theory, Vol. CT-2, 1955, pp. 74-85.

15. P. R. Geffe, "How to Build High-Quality Filters out of Low-Quality Parts," Electronics, Nov. 1976,
pp. 111-113.

16. P. R. Geffe, "Designers Guide to Active Bandpass Filters," EDN, Apr. 5 1974, pp. 46-52.

17. T. Delyiannis, "High-Q Factor Circuit with Reduced Sensitivity," Electronics Letters, 4, Dec. 1968,
pp. 577.

18. J. J. Friend, "A Single Operational-Amplifier Biquadratic Filter Section,"

IEEE ISCT Digest Technical Papers, 1970 pp. 189.

19. L. Storch, "Synthesis of Constant-Time-Delay Ladder Networks Using Bessel Polynomials,"

Proceedings of IRE, Vol. 42, 1954, pp. 1666-1675.

20. K. W. Henderson and W. H. Kautz, "Transient Response of Conventional Filters,"

IRE Trans. Circuit Theory, Vol. CT-5, 1958, pp. 333-347.

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21. J. R. Bainter, "Active Filter Has Stable Notch and Response Can be Regulated," Electronics, Oct. 2
1975, pp.115-117.

22. S. A. Boctor, "Single Amplifier Functionally Tunable Low-Pass Notch Filter,"

IEEE Trans. Circuits and Systems, Vol. CAS-22, 1975, pp. 875-881.

23. S. A. Boctor, "A Novel second-order canonical RC-Active Realization of High-Pass-Notch Filter,"
Proc. 1974 IEEE Int. Symp. Circuits and Systems, pp. 640-644.

24. L. T. Burton, "Network Transfer Function Using the Concept of Frequency Dependant Negative
Resistance," IEEE Trans. Circuit Theory, Vol. CT-16, 1969, pp. 406-408.

25. L. T. Burton and D. Trefleaven, "Active Filter Design Using General Impedance Converters," EDN,
Feb. 1973, pp.68-75.

26. H. Zumbahlen, "A New Outboard DAC, Part 2," Audio Electronics, Jan. 1997, pp. 26-32, 42.

27. M. Williamsen, "Notch-Filter Design," Audio Electronics, Jan. 2000, pp. 10-17.

28. W. Jung, "Bootstrapped IC Substrate Lowers Distortion in JFET Op Amps," Analog Devices AN232.

29. H. Zumbahlen, "Passive and Active Filtering," Analog Devices AN281.

30. P. Toomey & W. Hunt, "AD7528 Dual 8-Bit CMOS DAC," Analog Devices AN318.

31. W. Slattery, "8th Order Programmable Lowpass Analog Filter Using 12-Bit DACs,"
Analog Devices AN209.

32. CMOS DAC Application Guide, Analog Devices.

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