~ AUDIO AMPLIFIER

SYSTEMS

TRANSISTOR CIRCUITS
INTEGRATED CIRCUITS
LOUDSPEAKERS

Printed In Th e Netherlands 3BA01 (j)9-'70 9399 103 06901

 Audio Amplifier Systems

l
j
 Audio Amplifier Systems
transistor circuits - integrated circuits - loudspeakers

M. D. Hull, C. Eng., A.M.I.E.R.E.

PUBLICATIONS DEPARTMENT
ELECTRONIC COMPONENTS AND MATERIALS DIVISION
 List of Contents

© N. V. Philips' Gloeilampenfabriek en 1 Germanium and Silicon Transistors . . . . . . . . . . . . . . . . . . .

EINDHOVEN - The N etherlands

2 Low-level Audio Amplifiers and Control Circuits. . . 2
Se pt ember 1970 2. 1 D es ign Require me n ts fo r Au d io Pre-a mpli fiers 2
2. 1. 1 Input Fac ili t ies a nd Co ntrols . 2
2. 1.2 Frequ e ncy Ba nd wid th 7
2. 1.3 Signa l/ No ise R a ti o . . . . . 9
2. 1.4 D is to rtio n . . . . . . . . . 9
2.2 T ra n sistors fo r S ma ll-s igna l Applica tio ns . JO
2.2. 1 Ga in Co nside ra ti o ns 10
2.2. 2 Survey of T ra nsisto rs . . . . . . II
2.3 Lo w-l eve l Ampli fie r C irc uit s . . . . . . 15
2.3. l C irc uit I : Bas ic A .F . Vo ltage Ampli fie r. 15
2.3.2 Ci rcuit 2 : Low-di sto rtio n , Hi gh-o utput vo ltage A mpli fie r 21
2.3 .3 C ircui t 3: Bu ffe r A mplifie r . . . 22
2.3 .4 C ircui t 4: Micro ph o ne Ampli fier . . . . 24
2 .3.5 C ircui t 5: Mixe r Amplifie r . . . . . . . 26
2 .3.6 C irc uit 6 : M agnetic Pick -up Pre-am pl ifie r 27
2 .3.7 C irc uit 7 : Pre-am p lifie r fo r Pic k -up, T a pe a nd Ra di o In p uts . 30
2.3.8 C ircuit 8 : U nive rsa l Pre-a mpl ifie r . . . . 33
2.4 Co ntrol C ircuits . . . . . . . . . . . . . . 43
2.4.1 C irc uit 9 : Input Sw itc hing Arra nge me nts 43
2.4.2 C irc ui t I0 : Ste reo Ba la nce Co ntro l 45
2 .4.3 C irc uit 11 : Ste reo Ba la nce Me te r . . . . 47
2.4.4 C irc uit 12: A cti ve T o ne C o nt ro l 48
2 .4.5 C irc uit 13: So und So urce Width ( Dim e ns io n) Co nt ro l 50
2 .4.6 C irc ui t 14 : Lo w-Pass/ High-Pass Filte r 52
2.4. 7 C irc uit 15: Noise a nd Rumbl e Fi lte r . 53
2.4.8 C ircuit 16 : Prese nce Co nt ro l . . . . . 55

3 Power Amplifiers . . . . . . . 57
3. 1 Pe rfo rm a nce C ha ra cte ristics 57
3. I. I Ge nera l . . . . 57
3. 1.2 Di sto rti o n . . . 57
3.1. 3 Dyna mi c R ange 58
3. 1.4 D a mpin g Fac to r 59
The publication of this do cument does not 3.1. 5 . Power Ba nd wi d th 60
3.1 .6 O utput Powe r R a tin gs 60
imply a licence under any patent
 3.2 Output Stage Arrangements . . . . . . . . . . 61
3.2. 1 General. . . . . . . . . . . . . . . . 61 5 Loudspeaker Systems . . . . . . . . 169
3.2 .2 Principles of Complementary Output Stages 61
3.2.3 Quasi-Complementary Symmetry. . . . . 65 5.1 Principles of Loudspeaker Selection 169
3.3 Survey of Transistors for Power Amplifiers . . . . 65 5.1.1 Survey of Loudspeakers . 169
3.3.1 Transistors for Battery-operated Equipment 65 5.1 .2 Power Requirements . 175
3.3.2 Silicon Driver and Power Transistors 70 5.1.3 Methods of Mounting . . 181
3.4 Practical Layo ut of Audio Circuitry 70 5.2 Sealed Enclosures for High Fidelity Systems 182
3.4.1 General Principles 70 5.2 . l Principles of Design . . . . . . . . 182
3.4.2 Earthing . . . 71 5.2.2 Enclosure Volume and Frequency Respo nse 186
3.4.3 Stray Fields . . 73
3.4.4 Power Supplies. 73 Appendix - Heat-sink Design and Calculations 192
3.5 Economical Transistor Power Amplifiers 74
3.5. l Circuit 17: One Watt Stereo Pick-up Amplifier 74 Index . . . 197
3.5.2 Circuit 18: Two Watt Stereo Pick-up/ Radio Amplifier 76
3.5.3 Circuit 19: Four Watt Mono Car Radio Amplifier . 80
3.5.4 Circuit 20: Five/Ten Watt Low-cost Amplifier. . . . 83
3.6 High Fidelity Power Amplifiers . . . . . . . . . . . . . 86
3.6. l Circuit 21: 8 Watt Hi-Fi Amplifier using AD 161 / 162 Output Tran-
sistors . . . . . . . . . . . . . . . . . . . . . . . . 86
3.6.2 Circuit 22 : 15 Watt Hi-Fi Amplifier using BD 181 Output Tran-
sistors . . . . . . . . . . . . . . . . . . . . . . . . 90
3.6.3 Circuit 23: 20 Watt Hi-Fi Amplifier using BDI81 Output Tran-
sistors . . . . . . . . . . . . . . . . . . . . . . . . 97
3.6.4 Circuit 24: 25 Watt Hi-Fi Amplifier using BDl82 Output Tran-
sistors . . . . . . . . . . . . . . . . . . . . . . . . I 00
3.6.5 Circuit 25: 40 Watt Hi-Fi Amplifier using BD182 Output Tran-
sistors . . . . . . . . . . . . . . . . . . . . . . . 1J9
3.6.6 Circuit 26 : Overdrive Indicator for 25 Wand 40 W Amplifiers . . 133

4 Integrated Circuit Amplifiers . . . . . . . . . . . . . 135

4. 1 Application of Integrated Circuits to Audio Amplifiers 135

4 .2 Practical Circuits . . . . . . . . . . . . . . . . 13 6
4.2.1 Circuit 27 : One Watt Transformerless Class-B Audio Amplifier
for Record Players using the TAA300 Integrated Circuit . . . 136
4.2.2 Circuit 28: Two Watt Pick-up Amplifier using the TAA320 and
the high voltage Class-A Transistor BD 115 . . . . . . . . . J 50
4.2.3 Circuit 29: Four Watt Pick-up Amplifier using the TAA320 and
two high voltage BD 115 Output Transistors. · . . . . . . . . 155
4.2.4 Circuit 30: Four Watt Transformerless Car Radio Amplifier
using the TAA435 a nd AD 161 / 162 Output Transistors . . . . 157
1
4.2.5 Circuit 31: Recording and Playback Amplifier using the TAA3 l 0
Integrated Circuit . . . . . . . . . . . . . . . . . . . 161
 Foreword

Sound recording and reproduction became firmly established with the

invention of the gramophone in 1887. Ever since then , improvements in
the quality of the reproduced sound have taken place. The invention
of the electron tube and the moving coil loudspeaker revolutionized not
only the quality of the sound but also the shape and size of the equipment.
Now another revolution has taken place with the development of solid-
state devices. The use of the transistor and integrated circuit in audio
The circuits described in this book are proven designs and we regrel that equipment has brought about a radical change once again in the appear-
we cannot undertake to answer queries from home-constructors arising ance of the equipment. Never before have so many watts been packed
from the information presented. into so small a volume.
The development of small attractive units, blending subtly with modern
decor in the home, has not been entirely without some drawbacks.
Jn harmony with smaller amplifiers there must be smaller loudspeaker
enclosures and , to achieve this size reduction without impairing the sound
quality, the requirements imposed on the components concerned have
It should be stressed that the circuits have been constructed using com-
become more and more stringent.
ponents of our own manufacture throughout. No guarantee can therefor e
For instance, reducing the volume of the speaker enclosure results in a
be given that the published results will be obtained if other components are
lower efficiency for the same speaker. To obtain the same acoustic
used.
output, the electrical power to the loudspeaker system has to be increased.
This, in turn, means higher power ratings for the amplifier components
and power supply.
To ensure that our components keep abreast of the requirements for
modern audio equipment design , our Application Laboratories are
engaged in a continuous development programme. A comprehensive
range of miniature components is readily available for small-sized
equipment, in which transistors play the major role.
There are transistors for every audio application , ranging from low
noise to high power, for use in pre-amplifier, power amplifier and tape
recorder circuits. To demonstrate the differences in their properties we
have selected over 30 circuits representative of general and sophisticated
requirements . They cater for a wide range of tastes from the inexpensive
Whilst e very care has been tak en to ensure t!te accura cy of t!te information presented , and simple design, to the more costly high fidelity circuits.
no liability therefor ca11 be assumed. Jn addition to the circuits of complete amplifiers, there is a wide range
of circuits for individual stages, such as mixers, tone controls and filters.
These are intended to be used in combination with one another, like
building bricks, provided that a little discretion is used in their inter-
connection. All the designs are for construction on printed-wiring boards. Acknowledgements
Five of the amplifiers described use integrated circuits. The great
simplicity with which these can be employed makes them an attractive
Acknowledgements are made to members of the Staff of the Semicon-
proposition, not only for present-day designs, but for the equipment
ductor Laboratories, particularly W. A. M. Peters, G. W. J. van de Pol ,
of tomorrow.
J. J. Sips and J. Thi:irig, for their valuable assistance. Also to J. Vandekerck-
Since an audio amplifier system comprises all the equipment necessary
hove of the Loudspeaker Development Laboratory for carrying out a
to amplify the electrical signals from the input transducer and produce
large number of measurements on loudspeakers at the author's request.
a replica of the original sound , the loudspeaker forms a vital part of any
Some of the information included in this book has been taken from
audio amplifier system. To obtain the best performance from an audio
recently published material by Philips Electron Devices, Toronto ;
amplifier it is essential that the correct load conditions are applied a~d
Mullard Ltd. , London ; and Valvo GmbH, Hamburg, to whom thanks
we have therefore included information on loudspeakers for use with
are due.
the amplifiers, together with basic details of the design of sealed enclosures.
Finally, the author acknowledges the valuable assistanc.e of L. J. M.
Finally, an Appendix on heat-sink design and calculations has been
van Boldrik for many helpful suggestions and without whose tireless
included, since it is most important that adequate provision is made for
enthusiasm this book might never have been written.
the dissipation of heat in the power amplifier stages.
M.D.H .
 1 Germanium and Silicon Transistors

Nearly all transistors are made from either silicon or germanium. In the
past germanium was much easier to refine than silicon and, since silicon
presented many manufacturing problems, the early transistors were
made from germanium. The problems with silicon have largely been
overcome by the development of the planar process a nd both silicon
and germanium transistors are now produced concurrently. There are
applications for which each is the more suitable.
The operating junction temperature of germanium transistors is
limited to about 100 °C, whereas silicon devices can operate at temper-
atures in excess of 175 °C. However, germanium transi stors still have the
advantage of low values of forward base/emitter voltage and knee voltage
and, since these represent voltage losses in the circuit, germanium devices
are particularly suitable for the lower-voltage type of audio output stage.
The leakage currents of silicon transistors are far smaller than those
of germa nium transistors due to the extremely high intrinsic resistivity
of silicon (3 x 10 5 n cm at 27 °C) compared with germanium (47 n cm
at 27 °C). This advantage, combined with higher permissible junction
temperatures and high voltage ratings of silicon transistors, makes them
ideal for a wide variety of audio applications, especially at high output
The BEO MASTE R 3000 shown here is an elegant example of a high-fidelity stereo tuner- powers where high voltages are necessary.
amplifier, which uses small-signal transistors mentioned in Chapter 2. ( By co urtesy of The advantages of using either silicon or germanium transistors for
Bang & Olufsen A / S, Denmark .) specific applications are discussed for low-level audio amplifiers in
Chapter 2, and for power a mplifiers, including portable equipment,
in Chapter 3.

1
 used. Table 2.1 gives detail s of some of the recording characteristics
2 Low-level Audio Amplifiers and Control Circuits in past and present use. An example of a pre-amplifier offering a choice
of di sc equalization characteristics is given in Circuit 6.
2.1 Design Requirements for Audio Pre-amplifiers l n addition to di sc records, facilities for tape replay are often requ ired.
Tape recorded outputs are of two kinds: a h.igh-l evel pre-corrected output
2.1.l INPUT FACILITI ES AND CONTROLS of the order of 250 m V, or a low-level output direct from a playback head.
For the li stener the most common sou rce of programme material is The low-level output may be of the order of 300 to 500 µ V and will
undoubtedly the radio , followed by disc and tape in that order. The
microphone is normally used only in connection wit h a tape recorder,
apart from public address applications.
Radio tuners, specially designed to feed a low-level aud io signal to Table 2.1. Past and Present Disc Recording Characteristics.
high quality amplifiers are now becoming popular. Tuner outputs vary, BSI , AES, RIAA
but it is normal to expect a signal level of between 150 m V and 250 mV. frequency early Decca H.M .V. American
JEC
( H z)
Provision of an input to accept a tuner signal is therefore desirable and
this input should have a high impedance normally from l 00 kQ to 500 kD .
78 I 33 78 45 I N.A.B . 78 I 33/45
Crystal pick-ups working directly into a pre-amplifier require a high 20 - 18 I - 15 - 15. 5 - 18.6
input impedance, otherwise there will be considerable loss of bass. 30
50 - II - 14 - 12 - 26
I
- 16
-
-
14.8
14.0
I --17.8
17.0
Since they may produce as much as 1 V output, provision to handl e
60 I - 13.I - 16. 1
this signal has to be made at the pre-amplifier input, and extra precautions 70 I - 12 .3 - 15 .3
must be taken to prevent damage to the input transistor if the pick-up 80 - 7 - 13 - 8 - 15 - 11. 6 - 14.5
cartridge is dropped on the record, when a voltage of the order of I00 Y 100 - 5 - 12 - 7 - 18 - 8.75 - 10.2 - 13.1
I
may be generated. An input impedance of the order of 0.5 to 1 MD 150 - 7.6 - 10.2
200 I - II - 5.8 - 8.3
would be considered normal for crystal pick-up inputs and a pre-amplifier 400 0 0 - 2.3 - 3.8
would be designed to accept, a signal level from about 250 mV. 500 0 - 3 0 - 2.5 - 1.75 - 1.5 - 2.6
Magnetic pick-ups, very suitable for high fide lity applications, have a 700 0 0 0 - I - 0.7 - 1.2
lower voltage output than crystal pick-ups a nd conseq uently a high gain
IOOO 0 0 0 + 1 + 1.3 0 0
amplifier including frequency correction is essential. Assuming that the 1500 0 I + o.7 + 1.4
pre-amplifier unit must deliver 300 mY to fully drive the power output 2000 0 +2 0 + 3 + 4. 2 + 1.4 + 2.6
stages, a gain of aro und I00 would be required, since an output voltage 3000 0 I +4 +6 + 2.8 + 4.7
of only 3 mV could be expected from the pick-up. This leads to compli- 4000 + 1 +6 0 + 8.5 + 4.2 + 6.6
cations in the design of the pre-amp li fier and precautions have to be 5000 +2 +6 0 +9 10. 2 + 5.5 + 8.2
6000 + 3.5 + 6.7 + 9.6
taken to prevent feedback and in stability. Care must be taken to avoid 7000 0 + 11.5 + 13 + 7.7 + 10.7
I +9
hum pick-up, and earth loops should be kept as small as possible. An 8000 0 + 8.7 + 11.9
input impedance of around 50 H2 is required for a magnetic pick-up. 10000 -t 6 + 11.5 0 + 11. 5 + 16 + 10.5 + 13.7
For the reproduction of disc recordings the pre-amplifier circuit should 12000 + 11.9 + 15 .3
14000 + 13.2 + 16.6
provide fac ilities for equalization of the recording characteristic. Practi- 16000 I -t- 14.3 + 17.7
cally all present-day domestic disc recordings are in accordance with 18000 + 15. 3 + 18.7
the KI. A.A . characteristic but, to enable older recordings to be played, 20000 + 16.2 + 19.6
a switch providing a choice of eq ualization characteristics is so metimes I I I I

2 3
-
depend , amongst other things, on the tape speed. An input impedance
/
of the order of 10 kQ wou ld be normal for an inp ut direct from a magnetic v
head . /
~

M icrophone inp uts are not very common un less publi c address /
/
applications arise. The crystal microphone works under simi lar conditions ~"'
to the crystal p ick-u p, but it has a frequency response incomparable to v
1,,
the dynamic micropho ne which is favo ured for high fide lity work. /
v
The dy namic microphone, li ke the magnetic pick-up, produces on ly a low /

signal voltage of the order of 3 to 4 mV, and requires an amplifier input v

I/
impedance of between say, 20 kQ and 50 kQ. -1
'
I/
1Z~t3 9~
150
loudness level I
lphonsl ~ -2
intens ity 0 20 40 60 BO 100 120
le vel / 120 loudness level ( phon:;)
(dBi - 110
...... 100 Fig . 2.2. Effec ti ve loudness as a ji111ction of loudness level.
100
....... ~ -_,, 90
/ BO
t--.
t--. ._(; _,,, 70
tive loudness is suhstantially logarithmic above abo ut 40 phons and semi-
lo' ,.....__ - +- L / 60
-~" t--.::
~--......
~

........ ~
~

~
v
v
I/
L / 40
50
logarithmic below that level. Fig. 2.2 shows how the effective loudness
50
'-- ..... ~
/ 30 is related to the loudness level. By taking the loudness at various fre-
,........_ / 20
!'-.. v quencies for a given intensity from Fig. 2.1 and correcting for the modi-
- ....- 10
'
"- --
,__
i--.,._
/
/
0
fied logarithmic response of the ear, as shown in Fig. 2.2, a curve can
be p lotted show ing the relative effective loudness as a function of fre-
quency. Fig. 2.3 shows the resu lt and clearly ill ustrates how a reduction
in volu me causes a considerable drop in bass.
-so
10 10' 10' f !Hz ) 10
-
Fig. 2.1 . Fletcher-Munson equal-loudness contours.

To avoid interaction between the various signal inputs it is desirable

that inputs not in use are shorted to earth. An input se lector switch which
/
v )', BOphon

also performs this earthing funct ion is therefore a usefu l faci lity. ,
4 0 & 60
Other faci li ties which must be provided in addition to the input selector,
, I / , phon
i.;
are the volume and tone controls. From 10 dB to 20 dB boost and cut D 7
of both bass and treble is normally provided , but values much in excess p~gn ·' / J
60
of that are seldom required si nce the basic use of the controls is to correct !J phon / phon
40

the reproduced sound image, not disto rt it further. Where stereo ap-
plications are consi dered, a ba lance control is also usually fitted to
achieve aural equality of both channels in the listen in g room .
0.1
10 10'
I f {Hz )

Fig. 2. 1 shows the Fletcher-Munson equal- loudness conto urs. The effec- Fig. 2.3. Response of the ear as a fun ction of loudness level.

4 5
At low volume levels, therefore , reali stic sound reproduction requires phase cross-talk up to about 20 % can be introduced, the two channels
bass boostin g a nd to avoid the li ste ner having to reset the bass contro l becom in g further and further separated as the control is advanced.
each time the vo lume is adjusted , a physiological vo lume control is some- At about 24 % cross-ta lk the so und pict ure breaks up and becomes un-
times used. This automatically increases the bass at low vo lume leve ls pleasant. The apparent width of the " sound stage" can thus be varied
to mak e up for the insensitivity of the ear in that part of the spectrum wit h this control.
and, because it fo llows the Fletcher-M un son contours, it is sometimes Finall y, a number of useful filters may be employed. Rumble and
known as a contour contro l. The atten uat ion curves of a typical contour scratch fi lters can be used to make up for the deficiencies of turn-tables
contro l are shown in Fig. 2.4. a nd di scs respectively, the incorporation of a scratch fi lter having the
additional advantage that it may be used to remove the noise from radio
30
_ld B
tuner signals. For greater refinement low-pass and high-pass fi lters may
dB
- - 60 be incorporated in the reproduction cha in . The variable cross-over types
--so "
I I"
' normally provide a selection of frequencies at which roll-off commences
20
- 40 ' a nd a fixed rate of attenuation is m aintained : the var iable slope fi lters,
,\. o n the other hand , start to ro ll off at fixed frequencies and the rate of
" '\.\.
"\ atten uation can be varied in steps. An example of a variab le cross-over
I\\.
10 ,__ !30
' "'
r-..._ 1'. ~
fi lter having an atte nuatio n of 8 dB/octave is given in Circu it 14.
......... ~
1'-.l'N v -60
15

dB JI\
I 1
10
I
-10 /1
10 10' 10 3 f(Hz) I
I
/t'\.
Fig. 2.4. Frequency chara cteristic of a physiological vo lume control ( col/fo ur co11tro!). / 2
'
"' \
~
Artificially boosting the mid-range at a se lected frequency between /.~
~
3 ....
2000 a nd 3000 Hz produces the effect of bringing the vocali st or musical
instrument nearer to the li stener, and is known as presence. It is also a
valu able faci lity for correcting mid-range absorption in imperfect li sten ing
conditio ns. A presence contro l is somet imes fitted on a mplifiers and 5
10 10 2 10 4 ff Hz ) 10 5
gives a lift of around 6 dB at a selected freque ncy, normally between
2000 and 3000 Hz. Fig. 2.5 shows a typical mod ified frequency response. Fig. 2.5. A 111plit11de/freq ue11cy characteris1ic of a presence co11trol.
Curve 1: maximum; Curve 2: half;· Curve 3: mi11im11m.
Some stereo installations have a sound source width (or dimension)
control. This is a cont inu o usly variable contro l which a t one end of its
travel provides 100 % in-phase cross-talk betwee.n channels and results
in mono. T he sound source then appears mid-way between the speak- 2. 1.2 FR EQUENCY B AN DW I DTH
ers provid ed both speakers are in phase and ass uming acc urate Fig. 2.6 shows the frequency range of a number of musical instruments.
balancin g. When the control is turned in the opposite direction anti - But to li mit the freq uency bandwidth of an amp li fier to pass only those

6 7
 I 2. 1.3 SIGNAL/ NOISE RATIO
tympani
I -- - Thermal noise arises in any resistive component which is not at absolute
bass drum -- - ~-
ze ro temperature. The frequency spectrum of this thermal noise is in-
snare drum
finite . When the term signal / noise ratio is applied to sound reproduction,
I 14in. cymbals the noise components in each pa rt of the spectrum do not a ll sound
I _,_ equa lly loud as shown in Fig. 2.1. Noise is measu red via a network
bass violin
which has a frequency response approximately the inverse of the Fletcher-
~ -- Munson contours, usually the 70 phon curve, and the signal/ noi se
piano ra ti o determined in this way is known as the weighted signa l/ noi se ratio.
I violin ... --- Hum components may also be lumped with the noise and the difference
between the weighted and unweighted signal/ noise ratios will be even more
I
bass tuba
__ ,_ noticeable, since the weighting reduces the hum component by about
IO dB.
trombone
It is suggested that a signal/ noise ratio of 70 dB would be acceptable
french horn ,_
to a very critical listener in a very quiet room. Modern transistors, and
trumpet especially carbon film resistors, used in the low-level input circuits
I ma ke possible the achievement of signal/ noise ratios even better than
bass saxophone -- this. For example, the unweighted signal/ noise ratio of the Un iversal
bassoon - Pre-amplifier in Circuit 8 is > 90 dB on the magnetic pick -up input
bass clarinet -- position. The excellence of the signal /noise ratio will be apparent when it is
I
I
clarinet --- rea lised that the frequency bandwidth of this pre-amplifier is from I0 Hz
to 35000 H z, within 1 dB of the 1000 H z res ponse.
soprano saxophone --
j:_
L.!I~ --
2.1.4 D ISTORTION

Di stortion in all its forms should be as low as possible. It has been sug-
piccolo ---- gested that harmonic distortion below l % is undetectable by the ear, but
with modern semiconductors entailing transformerless configurations it is
male speech fairly easy to arrange for the harmonic distortion to be only 0.1 %.
female speech What is probably more important is the level of the intermodulation pro-
I ducts a ri sing from non-linearity. A very small percentage of intermodu-
20 so 100 200 500 1000 2000 5000 10000 20000
la tion products can be easily detected by the ear and this objectionable
frequency IHzl
Fig. 2.6. Frequency ranges of musical instruments. form of distortion occurs mainly in the output stage. This is dealt with in
greater detail under Power Amplifiers in Chapter 3.
frequencies is neither desirable, nor necessary . The transient handling It is particularly important that distortion arising in the input stages
capabilities of an amp lifier are directly related to its bandwidth, since a is kept very low. Further ampl ification will soon raise the level of any
step-function waveform is rich in high order harmonics. Many high distortion beyond a n acceptable limit. High-gain low-noise transistors
fide1ity amplifiers have bandwidths (- 1 dB relative to the response may be used in input stages, with heavy negative feedback to obtain
at l 000 Hz) of from 15 Hz to well over 30 000 Hz. linear operating conditions.

8 9
2.2 Transistors for Small-signal Applications other words, the stage gain is affected. The gain could be much higher
if the leak age current l c JJo wa~ less.
2.2.l GAIN CONSIDERATIONS

At the working point the parameters of the tran sistors, i.e. gain, lea kage
Rl
current, etc., have definite values under specified conditions. Temperature
is one of these conditions and it can cause a shift in the working point.
To prevent a shift occuring it is necessary to stabilize the values of direct +

current representing the working point and in the case of germanium

transistors it is usual to consider the circuit configuration of Fig. 2.7,
where conventional voltage biasing is shown. The effect of leakage
current on the stage gain under these conditions will now be considered. Fig. 2.9. Co11 ve11tio11a/ curren t-biased transistor.

Silicon planar transistors, such as the BCl49 and BCl59, have negligi-
ble leakage current - just a few pico-amperes (I pA = 10- 3 nA =
10- 6 µ A), a nd if silicon transistors are substituted for the germanium
R2 R; t ransistor transistors that have been hitherto considered, the bleeder resistors R 1
a nd R 2 can have high values and the stage gain will be increased con-
siderably.
The thermal stabilization of the stage can be simplified, however,
since with a small gain (hfe) spread it is possible to use current biasing
Fig. 2.7. Co 11ve11tio11al voltage- Fig. 2.8. Equi valent circuit of as shown in Fig. 2.9. Two resistors and one capacitor are omitted and
biased transistor. voltage-biased transistor.
a very hi gh gain is achieved. For small-signal a pplications, therefore,
sili co n transistors are preferred .

The current I flowing through resistors R 1 and R 2 is made much

larger (e.g. 10 tim es greater than the sum of the base current JJJ and the 2.2.2 SURVEY OF TRANSISTORS FOR SMALL-SIGNAL APPLICATIONS

leakage current I cJJo). The bleeder current is determined by the supply Th e availability of more or less identical tran sistors in both n- p-n and
voltage VJJ divided by the sum of the bleeder resistors, R 1 +
R 2 • Since p-n-p versions affords the widest possible flexibility in circuit design to
the leakage current of germanium transi sto rs is of the order of 10 to 60 µ A meet the requirements of direct-coupled stages, phase-splitting circuits,
at Tumb = 45 °C, the resi stance values of R 1 and R 2 will be low. etc. However, in order to reduce the number of transistors and so si m-
Assuming that the internal resistance of the battery can be neglected , plify the des igner's tas k, the properties of the transistors for different
compared with the resistance values of R 1 and R 2 , Fig. 2.7 m ay be re- circ uit functions have been combined a nd a small transistor range having
drawn as shown in Fig. 2.8 with the bleeder resistors in parallel with wide application has been achieved. This range covers not only the low-
each other. Their effective resi stance is therefore very low and this, in noise input stage req uirements of tape recorders a nd high fidelity ampli-
turn, shunts the input resistance of the transitor. At a norm a l d.c. fiers , but also the requirements of all the interm ediate stages before the
curr nt of about I mA the input resistance of the tran sistor is also low. driver stage. The main characteristics of the transistors are given in
Therefore only a part of the total current 15 flows into the transistor. In Tables 2.2 and 2.3.

JO Il
Table 2.2. N-P-N Transistors for Small-signal Applications. 4

envelope F
Ve Es VeEO f eM F typ ldB I

T0- 18 1
) Lock-fit 2
)
(V) (V) (A) (dB)
3)
BCI07(A,B) BC l47(A,B) 4 ) 50 45 0.2 125-500 2
BC I08(A,B,C) BC l48(A,B,C) 30 20 0.2 125-900 2
BCJ09(B,C) BC l49(B,C) 30 20 0.2 240-900 1. 2
2

' --...
,
typ
-
Table 2.3. P-N-P Transistors for Small- signal Applications.

envelope Vo:x 1 ! VeEO l eM F typ

1 2 (V) (V) (A) (d B) 0
T0-18 Lock-fit o.i fl kHz) 100
) )
'°
BC l 77 5 ) BC l 57 6 ) 50 45 0.2 75-260 2 ( a) BCJ09 and B C l49
BC l78(A, B) BC l 58(A,B) 30 25 0.2 75-500 2
BC l 79(A, B) BCl59(A,B) 25 20 0.2 125-500 I

1
) T 1 max = 175 cc, R"• l - a = 500 cC/ W
2
) T 1 max = 125 cc, Rr1, l - a = 400 cC/ W
3
) le 8 o < 15 µA at Vea = 20 V a nd T1 = 150 cc .
4
) leao < 5 µA at Ve 8 = 20 V a nd T1 = 125 cc.
5
) - lcao < 10 µA at - Vea = 20 V and T 1 = 150 cc. i..
6
) - l ci;o < 4 µA at - Vc 8 = 20 V and T1 = 125 cc. F
7
) VBE = Iv ld B I

Jn the a bove Tab les the letters A, B and C in brackets indicate the gain sp read where
T1 = 25 cc, l e = 2 mA , VcE = 5 V and f = I kH z, as fo ll ows: A = 125-260, B =
240-500, C = 450-900. The tra nsistors to which these letters appl y ma y be ordered
with an A, B o r C onl y.

..........
-....._
The noi se values given in the Tables do not sufficiently emphasize ,
the outstanding qualities of the transistors li sted . Transistor noi se depends --
t yp

on the current and vo ltage sett ings, the source resistance a nd the fre-
quency. Fig. 2.10 shows how the noi se figure F varies with frequency. 0
o.i f(kHz l 100
The low-noise wide-band transisto rs BCI09/ 149 and BCl59/ 179 feature '°
stable low noise even at very low frequencies, as can be seen. Figs. 2.11 (b) BC/59 and BCl 79
and 2. 12 show the constant noi se contours for the BC 159 transistor at
two'frequencies , l kHz and 10 kHz, the source resistance and the current Fig. 2.10. No ise fig ures of the B CJ09/149 and B C l 59/179 tra11sistors at l e= 0.2 mA ,
setting varying. VeE = 5 V, B = 200 Hz and T 1 = 25 cc.

12 13
 1B9t. O\
10'
2.3 Low-level Amplifier Circuits
- le
Im A I In the low-level amplifiers and contro l circuits which fo ll ow, all have a
high input impedance and a low output impedance and , with o ne ex-
10
ceptio n, a uniform supply voltage has been used , a value of l 8 volts
I
- ~
~

"-
bein g chosen. The circuits can, therefore, be connected together in a ny
o rder. A variation of ± 15 % in the supply vo ltage is possib le without
"' "'
~

a ny a ppreciable change in the performance of the circuits.

' F= ...__

" I'-.
10dB

1 6d 8
2. 3. I CIR CU IT I - BASI C A.F. VOLTAGE AMPLIFIER
3d8 -
Fo ur amplifi ers a re described in this Sectio n. They are derived from
I
I 1.7518 the basic voltage ampl ifier shown in Fig. 2.13 . Two d .c. feedb ack loops
10

Fig. 2.11. Constant noise col/fours of the B C 159 transistor at 1 kHz, rh e source resistance
and current selling l'ary ing.

10 1 1 l 59 t.0 2

- le
lmAJ

10
Fig. 2.1 3. Basic a.f vollage amplifier. S ee Table 2.4 f or component values.
y

--., "
Tab le 2.4. Component Va lues f or Basic Amplifier fo r different Voltage Gains
1~ci 8
' '\
c irc uit ga in gain gain gain
40 dB
6d8 f- refe ren ce 10 dB 20 dB 30 dB
I
3dB._ RI 4.7 kO 1. 5 kO 1. 5 kO I kO
R2 12 kO 15 kO 56 kO 180 kO
\ 1.7518
RJ 1. 8 kO 2.2 kO 2.2 kO 2.2 kO
-
10 10 2
R4 470 n 560 n 330 n 680 n
I R5 1. 2 kO 470 n 270 n 220 n
Fig. 2. 12. Constanr noise contours of rhe BC 159 rransisror at 10 kHz, rh e source resis- c - - - 10 pF
rance and current selling l'ary ing.

14 15
to achieve stabilization against temperature variations are used: one fro m
the emitter of the seco nd transistor to the base of the first, and the other
from the collector of the seco nd transistor to the emitter of the first.
0.2
The component values for circ uits with voltage gains of JO, 20, 30 and
40 dB are given in Table 2.4. The vo ltages at three points in the circu it,
/
together with the input and output impedances Z;,, and Z 0 " " are given
in Table 2.5.
d tot
1%1 I/
/ h
12 .SkHy ' '//
0 .1
Tab le 2.5. Voltages and Impedan ces for Basic Amplifier 1kHz -
~ ~ /'Z'oHz
voltage or gain gain gain gain _b::::::== ~ / "
impedance 10 dB 20 dB 30 dB 40 dB

VI 3.4 v 0.97 v 0.4 v 0. 15 v

V2 J0.8 v 9.3 v 9.3 v 9.7 v
VJ 5.6 v 3.55 v 2.3 v 3.4 v 0
O V0 (V)
Zi,, 145 kD 140 kD 135 kD 110 kD
Zout 63 n 140 n 260n 100n ( a) distortion

Figs. 2.1 4 to 2.17 show for the four circuits the total distortion as a
funct ion of the output voltage at three freque ncies: 40 Hz, 1000 Hz and
12 500 Hz, and the noise vo ltage at the output as a function of the gener- 10
ator resistance at the inp ut. The total distortion for all fo ur amplifiers
remains below 0. 1 %for outp ut voltages up to 1 V at 1000 Hz, and below I
,,
1 %for o utput voltages up to 3 V. The noise voltage referred to the input 7.5
/
/
in all four amplifiers is less than 1 µ V. The frequency response (-3 dB /
points) of all amplifiers is from < 20 Hz to > 20000 Hz. ./

......-
..... ---
....
2.5

0
, 10 10' 10'

(b) noise

Fig. 2.14. Total distortion and noise voltage at output of JO dB amplifier.

16 17
 1H9406
o . 5 ------------------------==~
1
0.2

/
/
II IIJ
7 / d lo t
/ 1%1 II
f-----+--+-+--l----+--1-- -1+-l----t--+--+--+---+--+-_____,
/
7
/ 7 J J'
:{'HJ,,,<'o1Hz
_I//
12.Sk Hj . /
/
_,,_,v-+-+--+--+--+--+--+---l---lf----l
1

0 .1
,,, v
v
v ./
I/
0.251--+--+--+-+-
1 1
~t.-:::: v ~
I/ 12. 5kH ~~kH,I/AoHz
~- ~
.-.-

0 o '---'--'--~-.i._--'---'--~-.i._--''---'--...J....-.J..__'---'---'
0 V0 IVI 0 1.5 7.5

(a) d istortion ( a) distortion

aC==±=±±:i::±±tlt:==±=±=±::±±±±±i=:=±=±=±:J::l±i±J
, 10 10 2
10 10' R9 lkfil

( b) noise ( b) noise

Fig. 2. 15. Total distortion and noise voltage at 0111p111 of20 dB amplifier . Fig. 2. 16. Tola/ distortion a11d noise voltage al 011tp111 of30 dB amplifier.

18 19
 2.3.2. CIR CU IT 2 - LOW- DISTORTION , HIGH -OUTPUT VOLTAGE AMPLIFIER

Fig. 2. 18 shows the circuit of an amplifier having a total distortion of

0. 11 % at I000 Hz and maximum output. The amplifier has a voltage
gain of 20 dB and a maximum o utput vo ltage of JO V. To achieve this
I/
high outp ut voltage with low d istortion, the supply vo ltage has been
I
J
raised from 18 V to 45 Y. The frequency response (- 3 dB points) is
1.5
I from < 20 Hz to > 20000 Hz, and the input and output impedances
J
I/ are 140 kQ and 200 Q respectively. The variation of total distortion
j
wit h output vo ltage is given for three frequencies in Fig. 2. 19.
/
12.5kHz If
y ,A t-

0.5
v
/
,, "
1kHz ..I. V 4QHz
v """
0
0
- i..-

-
~
,,.
v

( a) distortion

Fig. 2. 18. Circuit of /ow-distortion high -output amplifier.

200
/
VN 0.4
{µ VI
/
/
150 I
/ I
/ 0. 3
/ /
,, I

100
/
/
0.2
v
v /
JV /

50
/ / ,,
v
c..- 12. SkHz/ 4 0 Hz _,,,,,. 1k Hz
0 .1 ,,.
- -- - --
~ ~ /
~

10 '
_,__
10
0
( b) noise 0 5 10 V0 (V) 15

Fig. 2. 19. Total distortion of th e amplifier shown in Fig. 2. 18. Note the high output
Fig. 2.17. Total distortion and noise voltage at outp11t of 40 dB amplifier. voltage.

20 21
2.3.3 CIRCUIT 3 - BUFFER AMPLIFIER

A buffer amplifier circuit is shown in Fig. 2.20. Its first stage works in
the common-emitter configuration with a large amount of feedback, 0 .5 7l'9 1.15

while the second stage is an emitter follower. The circuit provides a high
input impedance of 3.6 MO and a low output impedance of 250 Q.
I
Voltage gain is unity, the frequency response (-3 dB points) is from
< 20 Hz to > 20 000 Hz. Fig. 2.21 shows the total distortion and noise I;
voltage at the output. I '1
0.1 5
/ '/_I
12 .SkH; / '/ ~kHz
/
v
v / "40 Hz

"" ~ ~
- - ::::::: r:::::: v
0
0 V0 (V)

(a) distortion

Fig. 2.20. Btdfer amplifier circuit.

~o~-2~=1::=:1:::t:t::t:±J±,to_=,==±=::t::±::tt±±:J±===±:::±::±::±:±±±8
R 9 1Mfil 10

(b) noise

Fig. 2.21. Total distortion and noise voltage at output of buffer amplifier.

22 23
 2.3.4 CIRCU IT 4 - M IC ROPHON E AMPLIFIER

Fig. 2.22 shows t he circuit of a microphone a mpli fier with a voltage

gain adjustable between 13 dB a nd 40 d B. It has only 0. 15 % d istortion
for a gain of 13 d B, and 0.75 % for a gai n of 40 dB, with an o utp ut
voltage of 2 V. T he total distortion for t he limi ting values of voltage - - 40Hz
l- - t - - - + --1 - - 1k H z
gain is given in F ig. 2.23 . The values of noi se voltage correspond to d tot
(O/o) --- 12.SkHz

those of t he JO d B a nd 40 dB amplifiers given in Figs. 2. 14 and 2. 17

respectively. T he input and outp ut impedances and the frequency
response (- 3 d B po ints) are shown in Table 2.6. O.SJ---+----l--+--+-+-_._-l---l---+--l---l-- . l . - - ' ----11--...J

- ==-
o ~~~--'-~-"--'--'---'---'--'-....J...-.1,_-'---'-~
0

( a) Vo ltage gain = 13 dB .

l!>L.15

-
- - 40Hz
Fig. 2.22. Microphone amplifier circuit. 1kHz
- -- 12.SkH z
-~
v
.;~

v
O.S
~
_,,, ~ v
Table 2.6. Impedance and Frequency R esponse of Microphone Amplifier J--~
v
v
/,. . v."-. '
impedance ga in ga in /

or frequ ency 13 dB 40 dB

z,,, 145 kO 120 kO

Z OUI 47 n 120 n
( b) Voltage gain = 40 dB .
f~ow e r < 20 Hz 20 Hz
fupp e r » 20 kHz 20 kH z
Fig. 2.23. Total distortion of microphone amplifier.

24 25

l
2.3 .5 CIRCUIT 5 - MIXER AMPLIFIER 2.3.6 CIRCU IT 6 - MAGNETIC PICK-UP PR E-AMPLIFIER
The circuit of a mixer amplifier is given in Fig. 2.24. Here two inputs The pre-amplifier circuit shown in Fig. 2.26 has a high input impedance
are fed to separate transistors which have a common collector load and so permits a magnetic pick-up of any inductance to be used without
resistor. An emitter follower stage ensures a low output impedance change in the upper a udi o response. A cho ice of five equali zation charac-
of 70 Q and the input impedance is 2.5 MO. At both inputs the voltage teristics is offered for which component values a re given in Table 2.7.
gain is unity.
Fig. 2.25 shows the total distortion of the mixer when a signal is
applied to one input with the other input short-circuited. The distortion
is 0.5 % for 2 V output, a nd drops to 0.1 % fo r outputs less than 0.5 Y.
To avoid exceeding 0.5 % distortion , the voltage of each input sho uld be
restricted to 1 V, otherwise overloading can occur.

Fig. 2.26. Magn etic pick-up pre-amplifier circuit.

Fig. 2.24. Mixer amplifier circuit with two inputs.

0.5

n
II Tab le 2.7. Component Va lues for Magnetic Pick-up Pre-amplifier

I , 'I circuit equa li zation characteristic

units
reference
/) I I I 2 I 3 I 4 I 5

0.2 5
;," R 56 56 56 47 47 kO
/17 Ci
C2
12
-
5.6
-
6.8
3.9
6.8
1.5
6.8
2.2
nF
nF
12 .SkH_y ~40Hz C3 25 25 1.5 3.2 5 µF
i...-- ~ ~ 1kHz

i..,..:: ~

0
For characteristics 1 and 2 the electrolytic capacitor of 250 µF in the
0
emitter ci rc uit of the second transistor must be connected to earth.
Fig. 2.25. Total distortio n of mixer amplifier. Thi s is shown by the dotted line in Fig. 2.26.

26 27
 The five eq.µalization characteristics are shown in Figs. 2.27 and 2.28:

- Characteristic I correspond s with the old European recording charac-

teristic (cross-over frequency 250 Hz) as used before the introduction
of the micro-groove.
20
' ..... ~
dB I'\

- Characteristic 2 corresponds with the recording characteristic which

was applied in the U .S.A. up to 1940 and used in Europe up to about
10
'
,1 ' 2

1950 (cross-over frequency 500 Hz). "' I\..

- Characteristic 3 represents the N .A.R .T.B. recording characteristic

'
used in the U.S.A . up to about 1960.

Characteristic 4 shou ld be used for eq ualization of records made in - 10

Germany between 1952 and 1955. It has time constants of 3180 µs,
3 18 µsand 50 µs.
-20
10 10' 10' 10 4 fl Hzl
- Characteristic 5 is the present international standard with time constants
of 3 180 µs, 318 µs and 75 µs, su itable for mono and stereo records. Fig. 2.27. Equalization characteristics I and 2 (see text).

The discrepancies between these equalization characteristics and the

required characteristics can be neglected. For example, equali zation
characteristic 5 shows up a difference of maximum - 0.5 dB at 30 Hz and
maximum + 0.7 dB at 15 kHz, i.e., it is within ± 1 dB of the R .l.A.A .
requirement.
The corresponding voltage gain at 1000 Hz, a nd the input output
20
5
impedances for the different equ ali zation characteristics are given in - 4
3
Table 2.8 . The distortion at l kHz and 4 V amounts to 0.25 % and is dB

below 0. 1 % for output voltages below 1.5 V. The noise voltage at the out- ·"
put is 22 µ V, measured with a generator source resistance of I kQ at
10
'"" °'"' 3
5
the input. 4 ~

Thi s pre-a mplifier is suitable for feeding into the " radio " or "a uxiliary "
input on most amplifiers. ~

"''-
Table 2.8. Voltage Gains and Impedances f or Magnetic Pick-up Pre-amplifie r
"' '
- 10
ga in or equali zation characteristic
impeda nce I 2 3 4 5
units
"
""
'" 4
ga in 30 30 25 27 26 dB -20
3"'-' s
10 10' 10' 10 4 f (Hz)
' Z;n 250 250 250 250 250 kD
Z o ut 160 160 190 240 240 n Fig. 2.28. Equalization characteristics 3, 4 and 5 (see text) .

28 29
I
2.3.7 CIRCUIT 1 - PRE-AMPLIFIER FOR PICK-UP , TAPE AND RADIO INPUTS The total distortion of the pre-amplifier plotted as a function of the
The circuit of an equalizing pre-amplifier for use with magnetic and output voltage, is shown in 2.32.
ceramic pick-up cartridges, radio, and tape playback heads is given
in Fig. 2.29.
20
22okn ,...
,---,--{=J--,----------O+l8V
dB

2.SµF I +
1ekn
164VI
, 10
'' '\.
+ + '
IOµF 2.SµF
116VI 164VI
.........
volume
10 kn I'...
ce ramic O--C::J--t--r--__, log.
~--Ooutput

-10
' ... ~

I'\.
'\.
' 1'
- 20
* I value ot R chosen 10 10' f I Hz I 10'
to suit available signal
Sib

Fig. 2.30. Equalization characteristic of pre-amplifier for magnetic pick-up input.

mono (stereo
53

i...1kn
to other channel
30
Fig. 2.29. Pre-amplifier circuit for pick-up, tape and radio inputs.
dB
A BC149 and a BC148 form a directly coupled pair, the base voltage of
the BC149 being derived from the emitter of the BC148. Equalizat ion is 20

obtained by feedback from the collector of the BC148 to the emitter of

'\.
the BC149. The equalization characteristics are shown in Figs. 2.30 '\.
and 2.31. 10 I\

No value is given for R, the input resistor for the radio position,
'
because this should be chosen to s uit the ava il a ble signal. '"
I'

r--.
For an output of 60 mV, the sensitivity for the various inputs at 1

1000 Hz is : '" .... ..... 2

magnetic pick-up 4 mV
-10
ceramic pick-up 170 mV 10 10 ' fl Hz)

ta pe head with tape Fig. 2.3 1. Equalization characteristic of pre-amplifier for tape head inputs. Curve 1 is for
speed 7t in/s (19. 1 cm/s) 6.5 mV a tape speed of 3-3/4 in/sand Curve 2 is for 7-1 /2 in/s ( 9.5 and 19 cm/s).

30 31
 1 "' 2.3.8 UNIVERSAL PRE-AMPLIFI ER
The Universal Pre-amplifier about to be described was designed speci-
fically for operation with the J 5 W, 25 W and 40 W high fidelity power
,_ amplifiers which are described fully in Chapter 3. It is suitable, never-
._ ... ........... ....,.'-"" theless, for use with a wide range of power a mplifiers since it will deliver
.......... a nominal output voltage of 440 mY/ 1 kQ.
0.1
,_ The pre-amplifier employs high levels of a .c. and d.c. feedback en-
.
~ suring a very low distortion level ( < 0.15 %) and variations from the
v nominal performance characteristics given here due to component
~r..--

~
vv tolerances are negligible.
v

0.01
0 0.5 Vol VI 1.5
output
to power
Fig. 2.32. Total distortion at output of pre-amplifie r sho wn in Fig. 2.29. cmpli f 1er

Fig. 2.33. Block diagram of Uni versal Pre-amplifier.

There are five different inputs: crystal pick-up, magnetic pick-up,

radio tuner, tape recorder and magnetic microphone. In the magnetic
pick-up position the amplitude/frequency characteristic is in accordance
with the R.l.A.A. standard. In addition , the pre-amplifier provides an
output signal of 0.35 m V /I kD for tape recording.
In Table 2.9 the input sensitivity, input impedance, frequency res-
ponse (- 1 dB points) and the unweighted signal noise ratio are given
for the various types of inputs. Sensitivities and input impedances are given
at a frequency of 1000 Hz for an output signal level of 440 mV. The signal/
noise ratios are given for an output signal level of 20 dB below 440 mV,
i.e., 44 mV.
The block diagram of the universal pre-amplifier is shown in Fig. 2.33.
Four n-p-n transistors of the BCl47 family are employed. The first
two amplifier stages are direct coupled with d.c. feedback , and different
amounts of a.c. feedback are applied depending on the type of input.
The circuit diagram is given in Fig. 2.34. In the magnetic pick-up
position the feedback loop R 1 5 -C 5 -C6 provides equalization according

32 33
 >
g
.
f ~
g .~ :?-
~
. ~
Table 2.9. Performance Characteristics of Universal Pre-amplifier

~0
+
~'ii
~
il
~
g•c
:~~o ~> £ l. il
sensitivity
input
frequency
unweighted
signal/ noise
~~ l +
G~1 [ input
(mV)
impedance
(kfl)
response ratio
(dB)
+

~ crystal pick-up
magnetic pick-up
300
4
1000
47
10Hz-35kHz
see Fig. 2.35
>
>
80
90
radio tuner 150 500 IOHz-35 kHz > 80
ta pe reco rder 300 500 10Hz-45kHz > 85
magnetic microphone 3.5 22 IO Hz-65 kHz > 80

to the R.1.A .A. characteristic. All other inputs are frequency indepen-
dent and have a fl a t response. To obtain a low noise level a BCI49C has
been used as the input transistor, whilst the second pre-amplifier is a
BCl49B. The overdrive ratio of the first two stages is > 20 dB with
respect to 350 mV across the volume control R 18 • The voltage gains from
the inputs to the top of the volume control at 1000 Hz are:
crystal pick-up 1.3 dB
magnetic pick-up 38.84 dB
radio tuner 7.34 dB
tape recorder 1.3 dB
magnetic microphone 40 dB
A BCI48B is employed in an emitter-follower stage, to obtain a high
input impedance and a low output impeda nce . As a result, the operation
of the tone control circuit is independent of the position of the slider of
the volume control potentiometer.

Notes to Fig. 2.34 (page 34)

1) R 22 is omitted in R channel pre-amplifier, together with C4 , R 37 and Ci6 •
For mono only, R 22 and R 37 are each replaced by 4.7 kD. and C 15 , R 34
and R 35 omitted.
2) Pre-amplifier delivers 440 m V suitable for dri ving 40 W amplifier with
this value fo r R 3 3 • For use with 15 Wand 25 W amplifiers output may be
reduced to 350 m V by making R 33 equal to 6.8 kD. and R 32 short circuit.
3) Only one channel is shown here. For stereo, ganged potentiometers may be
used.

34 35
The tone contro l circuit is of the feedback type, the bass and treble
potentiometers being connected in the feedback loop between the col- dB t---t--t--t-+++!-+t--t--f-t-~l-l+l---+--+-r--+-1-+H+--t--t--+-t--t+~

lector and base of the outp ut transistor TR 4 • This feedback system has the
advantage of a low outp ut impedance being maintained , while the bass
10 t---t--t--t-+++!-+t-"'c--t--f-l-~l-l+t--+--¥-f-+-1-+H+--+-+--+-+-++~
and treble contro ls can be var ied independently over a wide range. When
the sliders of the bass and treble control potentiometers are at mid-
positions, the feedback is practically independent of frequency . The
equali zation characteristic for the magnetic pick-up is given in Fig. 2.35
and the frequency response and tone contro l characteri stics in Fig. 2.36. -1o t---t--+-+-+++!-++--T--t--f-t-~1-1+1---+-...P...<-+1-+H+--+-+--+-+-++~

- 20 s _J

20 _,,,,.
dB - JO1L
O --'--.L..1-...L..L.1.J.J.J10-, - L . . -L-L.L.WUJ..1.1 0 4--'--f-'-IH-'zl....1....1..u.u10 5
0-3 --'--'-'-LI....L.l..11.i...

'\
10
'\
Fig. 2.36. Frequency response characteristic of Un iversal Pre-amplifier.
Curve I : bass and treble controls in mid-position;
Curve 2: maximum treble, bass flat;
Curve 3: minimum treble, bass flat ;
........... Curve 4: maximum bass, treble flat;
' ...
...
Curve 5: minimum bass, treble flat .
When max imum bass and max imum treble are applied together, curves 2 and 4 are
- 10 ' joined by the dotted line. Similarly , when the bass and treble controls are both at minimum,
curves 3 and 5 are joined by the dotted line.
\.
'\
'\
-20
f (Hzl
in voltage amplification is about 0.6 dB. Resistors R 34 and R 35 are
10 10 ' 10'
mounted on the printed-wiring board and the leads marked 3 and 4 in
Fig. 2.35. Equalization characteristic f or magnetic pick-up. Fig. 2.34 are taken to the externally mounted mono/stereo switch and
balance control respectively. Audio o utp ut is taken from the junction
of R 34 and R 3 5 . No series capacitor is required if the pre-amplifier is
This pre-amplifier may be used in mono applications a nd for such used for driving the I 5 W, 20 W, 25 W or 40 W power amplifiers des-
purposes the audio output is taken from the collector of the outp ut cribed later, but where the 8 W ampli fier is used, or any other power
transistor TR 4 • This stage has a voltage gain of the order of 2.6 dB. amplifier which has no input capacitor, a 6.4 µF /25 V electrolyt ic capa-
lf the input circuit of the power amplifier has no series capacitor, then an cito r mu st be in serted in series, its negative co nnection to the junctio n
electrolytic capacitor of 6.4 µF /25 V shou ld be inserted between the of R 34 and R 35 .
pre-amplifier and the power amplifier. In the mono position of the mono-stereo switch, the output voltages
For stereo applications two pre-amplifiers are used , the collectors of the two amplifiers are combined, the slid er of the balance co ntrol
of the output transistors being joined by a series network comprising being disconnected from earth .
C 15 -R 34-R 35 -R 3 6 , and so on in the other channel. R 3 6 serves as the stereo As shown with the component values in Fig. 2.34 the pre-amplifier
balance control, its slider being connected to earth when stereo program- will deliver a nominal output voltage of 440 mY. This is the correct
me material is reprod uced. At the mid-point of its setting the decrease level for driving the 40 W power ampli fier described in the next Chapter.

36 37
The input signal level required for the 15 W and 25 W a mplifiers is, how- The printed-wiring board for the L-channel pre-amplifier and its
ever, 350 mY and the pre-amplifier output voltage can be reduced to component layout a re shown in Figs. 2.38 and 2.39 respectively, whilst
this level by making R 33 equal to 6.8 kQ and replacing R 32 by a shorting those for the R-channel are shown in Figs. 2.40 and 2.41. Fig. 2.42 shows
link . Other values of output voltage may be obtained by changing the the external connections which have to be made to the L-channel pre-
values of R 32 and R 33 correspondingly. The overdrive ratio of the com- a mplifier. The volume, balance and tone controls, together with the mono/
plete pre-amplifier is > 20 dB with respect to the nominal output vol- stereo switch and input selector switch a re mounted on the front panel
tage of 440 mY, so the undistorted maximum output is > 4.4 Y. of the unit enclosure. Input sockets would normally be situated on a
The supply voltage for the pre-amplifier is 30 Y. Where the output rear panel. For stereo in stallation s the L- and R-channel printed-wiring
power amplifier has a regulated supply unit the d .c. voltage for the pre- boa rds should be mounted reasonably close together in order to keep
amplifier can be obtained from there via the series resistor R 37 . Accomo- interconnection s as short as possible.
dation for this resistor has been provided on the printed-wiring board. The reader is referred to Section 3.4 and particularly Fig. 3.8 where
lf no stabilized supply is available the use of an additional active smooth- details are given of the interconnection of units to avoid earth loops.
ing filter is recommended and a circuit is shown in Fig. 2.37. This These recommendations should be foll o wed wherever possible, otherwi se
filter should be connected between the pre-amplifier and the power feedback troubles are likely to ari se.
supply unit.

R37
mono 4..7 kfi
ste reo 2.2 k.fi
+ 30 vC>-~-0--C
mono 7mA
st ereo 14 mA

+ C17
160
+ R38
CIB
l
v,

Fig. 2.37. Circuit of additional smoothing filt er for use where a stabilized supply is 110 1
a vailable. C 17 is located 011 the primed-wiring board. Wh en a stabilized supply is used,
R 3 7 is also mounted on the printed-wiring board ( see Fig. 2.39) .

The first two stages of the pre-amplifier have an additional decoupling

network comprising R 22 and C 4 • For stereo installations these compo-
nents are necessary in only one pre-amplifier, the supply voltage being
taken directly to the other channel from the point in the circuit immedia-
tely following R 22 • This is the lead marked 2 in Fig. 2.34. Similarly,
transistors TR 3 and TR 4 in the other channel are fed directly from R 37 Photograph of the Uni versal Pre-ampli-
fie r, illuminated to show th e printed
via the lead marked I in Fig. 2.34. Total d .c. con sumption for one wiri11 r: 011 the reverse side.
pre-amplifier is 7 mA.
For mono applications, where only one pre-amplifier is employed ,
resistors R 11 and R 37 are each replaced by 4.7 kQ. Output circuit com-
ponents C 15 , R 34 and R 35 are also omitted , sin ce the audio output is
taken from the collector of TR 4 , a s previou sly described .

38 39
.j:>.
0

Fig. 2.38. Printed-wiring board for left channel pre-amplifier.

7Z59517
Fig. 2.39. Component layout of left channel pre-a mplifier .

..

7Z5951ll
Fig. 2.4 1. Printed-wiri11g board for right channel pre-amplifier .

.j:>.
Fig. 2.40. Component lay out of right cha1111el pre-amplifier.
 2.4 Control Circuits

.,
0.
2.4.J CIRCU IT 9 - INPUT SWITC HING ARRANG EMENTS
c
The compatibility of hi-fi units is always something of a problem and
many difficulties in operation have arisen due to incorrect impedance
matching and unsatisfactory signal input level. The problem of input
and output impedance is much less than hitherto , thanks to standardiza-
tion between equipment manufacturers but the problem of signal level
still remains.

magnet ic
pick-up
channel
circuit

aux iliary

Rl
t uner
l Mil
hn.

R4

lM il.
lin. R7
lMil

RS
220kfi
SSb
monitor

t o R channel to R channel

Fig. 2.43. Input switching arrangements. Switch fun ctions are as follows:
S 1 magnetic pick-up S 4 tape playback
S 2 auxiliary input S 5 monitor head
S 3 tuner S 6 mono.

Figs. 2.43 offers the constructor a so lution to thi s problem , resisto rs

R 2 , R 3 and R 4 acting as varia ble input attenuators in conjunction with
resistors R 1 and R 8 • Where excess ive signal levels are present , adjustment
1- - - - g - - -- 1
..., E~ ~ +i ~
of the preset potentiometers should be carried o ut so that the same out-
0.
,
Q
e~ :i ·c .!ii
..
'+-a..~ :::J .....

42 43
put power, and hence the volume, is obtained irrespective of the type of 2.4.2 CIRCUIT 10 - STEREO BALAN CE CONTROL

input selected. This will avoid the volume control having a different The balance control circuit of Fig. 2.44 enables the voltage gain of
setting for each input. A suitable level should be first selected with the both stereo channels to be varied by 6 dB in opposite directions. The
magnetic pick-up in use, after which the appropriate presets are adjusted controlling variable 1 esistor forms part of the feedback circuit. Average
for the same volume level. gain is 23.4 dB , and Fig. 2.45 gives the total distortion for maximum
Switches S 1 to S 4 select the input required , whilst shorting to earth the a nd minimum gain. The differences are only small because of the large
unwanted inputs. An output signal for recording purposes is taken from amount of feedback employed. The noise voltage corresponds to that of
the junction of R 7 a nd R 8 to the tape recorder socket. Switch S 5 con- the 20 dB version of the basic a mplifier given in Fig. 2.8. The frequency
nects the amplifier circuit which follows to a monitor tape head , so that
the recorded programme can be instantly compared with the " original ".
This facility applies only to tape recorders with separate record and ~---~----0+ 18V

playback heads, or where an extra monitor head is fitted. Switch S6

1.8kfi
combines the L- and R-channels for mono.
output

4 .7 kfi
12ki250~F

,.. 1Z~ 94 JO

to other channel

Fig. 2.44. Stereo balance control circuit.

response (- 3 dB points) is from < 20 Hz to > 20000 Hz, whilst the

input and output impedances are 140 kQ and 85 Q respectively.

The 25 W high fidelity power amplifier

described on page JOO measures only
11 .75 cm long X 9.5 cm wide.

44 45
 2.4.3 CIRCU IT 11 - STEREO BALANCE METER

Most stereo equipment is fitted with a balance control to achieve aural

equality of the two channels. Lack of sensitivity of the ear in detecting
d tot H -+-t-t-t-+-t-t-t-++-t-t-++-t-t--++-1-+--++-1-+-l--l-W--I
small changes in volume levels can make the operation of balancing
l%1 H -++-H-++-H++-H++-H++-H++-H--l--hw--I somewhat tedious if precision is required and a visual indication that
o.3 H -++-H-++-H++-H++-H++-H+-HH--l--H---1---1 balance has been achieved will ease the task considerably.

0. 2 t-+-++-t-+-++-r-+++-r-+++-r-+-+-+-lkA':::.+-+-1---1---l--H---l--J

12.Sk Hz t.....- i
l-chonnel R-chonnet
o.1 H -+-t-H-+-t-t:. . rf".l-++_
. --±::-t-f".f:++::b-+"'f'.=!-1--+++-l--+--l loud speaker loudspeake r

l'---~-~__.______,~l
-- Fig. 2.46. S tereo balance meter circuit.

Max imum gain

A simple meter circuit which may be used for balancing the outputs
in a stereo system is shown in Fig. 2.46. Capacitors C 1 and C 1 damp the
meter movement and resistor R 1 is chosen to suit the current rating of the
centre-zero moving coil meter and the amplifier output. The value of
R 1 would be typically IO k.Q for a meter having a 1 mA movement.
d t ot H -++-t-t-+-+- t-t--++-t-t--++-H --++-1-+-l-+-H -l--l-H --I
f0 /o) H -+-J-+-t-+-J-+-t-!--l-H-!--l-H--++-H--++-1-+--+-l-H --/
o.3 H -+-t-t-t-++-t-t-++-t-t-++-t-t--++-H--++-1-+--++-H--I

0.2H -++-t-t-++-H --++-1-+--++-1-+-l--l-H-l--l-W-l--l-W--I

__
0.1H++-H++-H++-f---t,:..!..:2.65k~H,:.,z ~+-:~4;,;:0!,!jHFt-+_+~:d....f-1~kp.!H''+-f--+.....J
F'
-
OO'---'---'---"-L.......L--'--..L.JL.......L--'--..L.JL....L--'--..!-L....L--'--.L...JL....L..J..._.L...JL
~~IV-Jl---1-..J.....L...J

Minimum ga in

Fig. 2.45. Total distortion of balance control circuit.

46 47
2.4.4 CIRCUIT 12 - ACTIVE TONE CONTROL

Fig. 2.47 shows an active tone control circuit which employs a frequency-
/
dependent feedback network between the collector and the base of the
transistor. For input voltages less than 250 mV the total distortion
d tot
(
0
/o) I/
remains below 0.1 %, rising to 0.85 % for an output voltage of 2 V at
v
I
12 500 Hz. The variation of total distortion with output voltage is shown /
0.5
/
for three frequencies in Fig. 2.48.
The tone control characteristics are shown in Fig. 2.49. The range of ,,, /v /
-<Hz / /(oHz
control extends from - 22 to + 19.5 dB at 30 Hz, and from - 19 to + 19.5
12 .Skt;y
/
dB at 20 kHz. With the controls at their mid-positions the voltage gain is ...... ~ /
/,,,..
0.91. The input and output impedances at l kHz are 40 kQ and 180 Q ;:::...-- v /'

respectively. -~
---~ ~

0
0 V0 IV)

Fig. 2.48. Total distortion of active tone control.

~---<---<> + 181/
39nF 30
180kfi 3.9kfi
dB
+ 10.SV 20 1
50µF - "'i ./

.....
10 " '"'
,___~ 2
4 5 -
,,
2

5 i:""""
33kfi
.
/

//

2.2 nF 100 kfl ti n. 2.2 nF

-10
,,,,,
~

~
-20
Fig. 2.47. Active tone control circuit. 3

-30
10 10 4 f( HZ)

Fig. 2.49. Characteristics of active tone control circuit.

Curve J: max imum bass boost, maximum treble boost;
Curve 2: linear frequency response (controls flat};
Cur ve 3: maximum bass cut, max imum treble cut;
Curve 4: maximum bass boost, maximum treble cut;
Curve 5: max imum bass cut, maximum treble boost.

48 49
 0.4
2.4.5 CIR CU I T 13 - SOUND SOUR CE WID T H (D IM ENSION) CONTROL

Th e a pparent width of the "sound stage" ca n be varied in stereo syste ms

by delibera tely introducing cross-talk betwee n the channel s, part of the O.J
sign al vo ltage of one channel bein g added to the second channel. The
circuit shown in F ig. 2.50 provides continu ous co nt ro l between in-phase I/
cross-talk of 100 %, co rresponding to mono operat ion, and a nti -phase 0. 2
/
~
cross-talk of 24 %. Greater anti -phase cross-talk is not required since /
/
Q
the so und impression will fa ll apa rt fo r greater va lu es. v ~
12.SkHz v
The vo ltage ga in of the circuit is 0.5. Inpu t a nd output impedances a re 0.1
...1-1' A
1kHz & 40Hz
750 kQ a nd 47 Q respectively, and the frequency response (- 3 dB points) ......, .....
is from < 20 Hz to > 20 000 Hz. Fig. 2.5 1 shows the total d istortion
of the circuit as a function of output voltage for three frequenci es. 0
0 0.5 \1 0 IV ) 1.5

Fig . 2.51. Tora/ disrorr io11 of so1111d source widrli. co111rol.

,----~----------,.--O+lBV

L
output

IOkfi
0 .22
µF

R
output

0.1 µF
. R o-11-+- ->---H
input

68 k!l 5 k.fl. balance

~---+----------+--O+lBV

Fig. 2.50. So1111d source width ( di111e11sio11) co111rol circ11i1.

50 51
2.4.6 CIRCUIT 14 - Low-PAss/ HtGH-PAss FILTER 2.4.7 CIRCU IT 15 - NOIS E AND RUMBLE FILTER

In Fig. 2.52 the circuit of a low-pass/ high-pass filter is shown. Jt consists of The circuit of a noi se and rumble filter is shown in Fig. 2.54. Bass and
two RC networks connected in series with a buffer amplifier, the circuit treble cut are produced by a n RC network connected between two
of which was given in Fig. 2.20. The frequency characteristics are shown emitter followers , and a feedback loop from the output to the input
in Fig. 2.53. through a second RC network . A high slope of a round 13 dB/octave is
achieved. The frequency limit of the rumble filter is fixed at 45 Hz, but the
noise filter can be switched to limits of 16000, 12000 and 7000Hz.
h-C:=h-....----...--.--ooutput The frequency characteristics are shown in Fig. 2.55.
3.9 2.2 1 3.9 2.2 1 82 0
nF nF ilF nF nF nF pF
i...1kn

4 .7k11
-, -,
I I
I I 120 160 240 , - -- ....-- - - - - - - - - --.-- --.---t--o+ 18 V
I I pF pF pF
I I
I I
I I 21okn
I I
I I .- -
I I input 1
I
I o--j,_..--r-:=::i-;..I _..__-1---r-H
82kfl 100 kn 120 160

rI
I 27nF I
I pF pF
I
I
I high pass .,. " .,.
1

~~~ ~a:s___ _____ ~-----------------_-_-_-__r --- --- -- -;~9:-l~

270k0 39k11
Fig. 2.52. Low-pass/high-pass filter circuit. ! ,,.o _···· / 7l59506
I I
L---- -- ----- -----------~

dB Fig . 2.54. No ise and rumble filter circuit.

....___ '---- 1
-- ' '
''
-5
' The voltage gain is 0.95 a nd the total distortion at !000 Hz and an
output voltage of 2 V is 0.35 %, falling to less tha n 0. 1 % at I Y. The in-
-1 0 I
I
\
put and output impedances are 1.7 MD and 450 Q respectively.
I/ \

4
-15

\
-20

-25
10 10' 10 ' 10" fl Hz I 10'

Fig. 2.53. Frequency characteristics of the low-pass/high-pass filter. Th e - 3 dB points of

the c;1rves are as f ollows:
Curve 1: 40 Hz and 11 kHz Curve 3: 160 Hz and 4.5 kHz
Curve 2: 80 Hz and 9 kHz Curve 4: 270 H z and 3.2 kHz.

52 53
 2.4.8 CIRCUIT 16 - PR ESENCE CONTROL
The active presence control circuit shown in Fig. 2.56 uses frequency
"' 4
,- \
se lective negative feedback with amplitude control to achieve up to
-s 13 dB boost at 2000 Hz. The feedback network is a bridged-T fi lter. The
frequency characteristics of the presence control circuit are shown in
- 10
~
, Fig. 2.57.
Nominal gain at flat response is 0.95 and the input and output impe-
I
- 1S dances are 12 kQ and 100 Q respectively. The total distortion remains
below 0. 1 % provided that the output voltage does not exceed 250 mV.
-10 The distortion characteristics are shown for three frequencies in Fig. 2.58

-1S
10 10 ' 10' 10" fl Hz ) 10'

Fig. 2.55. Frequency characteristics of noise and rumble .filter.

Th e - 3 dB points of the curves are as follows :
Curve 1: 45 Hz (app lies to Curves 2,3 and 4) ;
Curve 2: 7 kHz;
Curve 3: l 2 kH z;
Curve 4: 16 kHz.

Fig. 2.56. Presence control circuit.

54
55
 15

dB I \ 3 Power Amplifiers
I \
I
10
/1 3.1 Performance Characteristics
I
I \
I ,...., 3. 1.1 GENERAL
h \
\ In the discussion of the performance characteristics of power amplifiers
~
which follows, the subject has been treated from a high fidelity standpoint
/.
3 ~ since this represe nts the most stringent requirements. Jt is realised
'°'
that high fidelity specifications do not apply to less expensive equipment
and accordingly the performance specification may be relaxed to meet
the less exacting requirements . Details of four economical power ampli-
-5
10 10' 10' 10 L. f (Hz ) fier circuits are given in this Chapter, in addition to five circuits which
Fig. 2.57. Frequency characteristics of presence comrol circuit .
meet the high fidelity specification .

Curve /: maximum; Curve 2: mid-position; Curve 3: minimum.

3.1.2 DISTORTION

Frequency distortion occurs if the gain is a function of frequency and

it is expressed on a relative gain/frequency basis. Using modern com-
ponents, amplifiers having a response linea r to within a fraction of a
decibel over a wide frequency ba nd are now commonplace. This has been
achieved by omitting all frequency-conscious components, wherever
possible, from the design and using directly-coupled stages in which
no coupling capacitors are employed. Electrolytic capacitors for the
input and output circuits of power amplifiers should be of high quality
and a wide range of suitable components is available for this purpose.
Another reason for avoiding the use of capacitors is that they can
I J cause phase distortion. When different frequency components suffer
v selective phase di splacements the original waveform will not be preserved .
To ensure that all frequency components of a complex waveform are
I/
0 .5

I/ v transmitted through the amplifier at the same rate, the phase angle be-
tween the output a nd the input voltages should be independent of fre-
12.skHV /
V;;;Hz 40Hz
quency. In other words, the phase angle for every frequency should be
v v
I/
v
[_....--
-v]....--' v zero . Phase di stort ion is only detectable by the ear when a phase difference
of many tens of cycles exists, or when the wavelength is comparable
- i--: -
[_....-- 1---
with the distance between the ears. This is of the order of 1,750 Hz.
0 For frequencies below this, phase distortion remains undetecta ble unless
0 Vo I Vl
the phase difference is larger, or a stereo system is used.
Fig. 2.58. Total distortion ofpresence co ntrol circuit. Because of the rising popularity of stereo systems, it is essential to

56 57
keep phase distortion in the reproduction chain to a minimum , otherwise In the concert hall a dyna mic range of a bout 70 dB could be expected
there will be a loss of the sound-location faculty, and so the use of from a la rge orchestra, but with recorded mu sic the dyn a mic range is
capacitors should be restricted . Where they must be used, they must be less than this. High quality tape recordings played on professional
high qu ality components. machines do not have a dyn a mic ran ge exceeding about 60 dB, and
Non-linear di stortion is the most objectionable form of distortion. domestic standard ta pe recorders provide considerably less. The same
It may be of two kinds: ha rmonic distortion and intermodulation dis-
tortion . Whilst 1 % total harmonic distortion might be considered ac-
ceptable, the intermodulation products a ri sing from non-linearity are
intolerable a nd only a small proportion of these have to be present for
the di stortion to become unpleasant. Jn transi stor a mplifiers having high relative
output
power outputs the main so urce of non-linear di stortion is in the power . (d Bi roted

output stage. High order odd harmonics are caused by slight disconti-
nuities in the tran sfer characteristic at the cross-over point arising from
s ignal/noise
the inherent asy mmetry of a quasi-complementary output stage. For r atio ;
approx . dy namic
cross-over distortion to be eliminated it is therefore essential that a careful range

choice of components is made. Circuit 24 is a typical example of a high I

power a udio amplifier design in which the total harmonic distortion ___ ___ _ o~~t~o~e~~ _ _ _ _ l
does not exceed O. l %. r elative input (dB)
Provided that non-linearity a nd phase distortion are low, transient
distortion in a modern transformerless power amplifier is negligible, Fig. 3. 1. Effect of noise level on dy namic range.
the main source of this di stortion normally being the loudspeaker.
Up to now the speaker has remained the last offending source of tran-
sient distortion , but recent developments have resulted in a new range
of high fidelity loud speakers becoming available. The transient-handling
capabilities of these speakers is excellent, less than I % distortion oc- applies to di sc recordings where a dynamic range of 55 to 60 dB is
curing with the I" tweeter when it is correctly mounted. possible. Both AM a nd FM radio tran smi ss ion s have a mplitude li-
miting and volume compression applied , so it is unlikely that the li stener
will experience a dynamic range greater than 60 dB unless a microphone
3. 1.3 DYNAMI C RANGE
is used.
Dynamic range is the ratio of maximum to minimum sound intensity,
expressed in decibels. In any so und reproductio n system it is a lso used
to express the ratio of maximum to minimum signal which can be hand- 3. 1.4 DAMPING f ACTOR

led by all , or part, of the system. The maximum signal level is ta ken to The damping factor is the ratio of the internal resistance of the output
be the full power output, normally meas ured at the onset of clipping, of the amplifier to the imped a nce of the load.
whilst the minimum signal level is determined by the noi se level. Noise With modern tra nsformer less amplifiers in which feedback is generous-
originates mainly in the pre-amplifier input stage and it can be clearly ly a pplied, the interna l resista nce of the a mplifier will be only a fraction
seen from Fig. 3.1 why it is so importa nt to use low- noi se components of an ohm a nd a high damping factor will be easily achieved. This en-
at tl1e input of the pre-amplifier. The dynamic range is roughly equal sures that where, for example, a single full-range speaker with a power
to the signal/ noi se ratio. ha ndling capacity of 40 W is mounted on an open baffle, adequate

58 59
 -~

Unless otherwise stated, the ratings of all a mplifiers given in thi s book
damping can be obtained. A damping factor as high as a 100 could be
are quoted in terms of the continuous r.m.s. sine-wave power which the
applied in such a case, whereas a damping factor of 3 would be adequate
amplifier will deliver for at least JO minutes into a specified load with a
for a modern sealed enclosure.
total distortion of I %.

3. 1.5 POW ER BANDWIDTH

3.2 Output Stage Arrangements
The power bandwidth is the frequency respon se at a particular level of
high output power and low di sto rtion . In most cases this respon se is 3.2.1 G ENE RAL
measured at the onset of clipping, for a distortion of I %.
A conventional push-pull output stage, with output tran sistors of the
The practice of publishing only the a mplitude/frequency response
sa me pol a rity , requires the two input signal s to be in anti-phase. These
curves to illustrate the performance of an audio amplifier was sufficiently
may be obtained from a driver tran sfo rmer. But there are many di sad-
informative in the old days. With modern solid-state high fidelity a m-
vantages of using a transformer for phase-splitting, particul a rly its
plifiers there is greater interest in power dependence on frequency and
it is now co mmon practice to publish the power/frequency characteristic, effect on the frequency response , and hence the power bandwidth.
The upper end of the audio system is affected by the stray inductance
which shows the power performance of a high fidelity amplifier over its
a nd capacitance of the transformer a nd the lower end by the primary
entire bandwidth, in addition to the a mplitude/frequency characteristic.
self-inductance. In addition , the non-lin ear behavi o ur of core material
introduces di stortion. There are also copper losses , and dissipation of
3.1.6 OUTPUT POWER RATINGS part of the a.c. driving power in the windings.
Not only do the above remarks apply to the driver transformer, but
A number of different ways of defining the power output of an amplifier
to the output transformer also. Di stortion , efficiency a nd bandwidth
are in use. The most common a re:
depe nd to a very large extent on the quality of both transformers.
l . Sine-wave Po wer: the output power attained when the amplifier Moreover, tran sformers for use in high quality audio a mplifiers are
is sine-wave driven. expensive and bulky components, and even these do not completely
2. Music Po wer : the equivalent sine-wave power attained during pas- eliminate losses and di stortion .
sages of music. Thi s may be as much as one-third higher than the sine- A further drawback is that, being phase-shifting elements, transformers
wave rating, but it depend s on the regul a tion of the power supply. limit the a mount of negative feedback which can be applied since there is
lt is measured at a supply voltage which is equal to the zero signal d .c. always a chance that the feedback might become positive, giving rise
suppl y voltage of the amplifier. Where a stabilized power supply is to insta bility.
used , the amplifier voltages do not change and the music power is the The most economical solution to eliminating the use of transformers
sam e as the sine-wave power rating. is to employ p-n-p and n-p-n transistors in complementary symmetry.
There a re many other method s of defining the output power, such as The small reduction in se nsitivity due to the a bsence of the driver trans-
"peak power" and "speech a nd mu sic power" , butt he more common former is easily compensated for by the high gain of the latest tran sistors.
methods stated above are generally preferred. The ratings may be Exceptionally, a n additional stage of amplification may be necessary .
quoted: A comprehensive range of complementary transistors is given on page 68.
(a) at a specified di sto rtion.
(b) ~t clipping, where the working point of the transistors reaches the knee 3.2.2 PRINCIPLES OF COMPLEMENTARY OUTPUT STAGES
of the characteristic, resulting in a sudden increase in distortion of the
Fig. 3.2 shows the basic circuit of a complementary output stage. For
output waveform.

61
60
the sake of simplicity the components required for d.c. adjustment have transistor is then cut off. The combined effect of these currents produces
been omitted. The transistors are assumed to operate in Class-B. an output voltage across the load resistance.
On positive half cycles of input voltage the n-p-n transistor is conduct- For single power source operation, as shown in Fig. 3.3, one end of
ing and th~ p-n-p transistor is cut off, so that current iv flows through the load is connected for the negative line, while the other end is connect-
the load resistance in the direction of arrow I. During the negative half ed through capacitor C 3 to the common emitters. The d.c. cottage Vi
cycles of the input signal , the p-n-p transistor is conducting and the at the point Q is approximately half the supply voltage. The maximum
current i,, flows through the load in the direction of arrow II. The n-p-n output power of the circuit depends on the maximum ratings of the
transistors, such as collector peak current, collector dissipation and
knee voltage, on the load impedance RL , and on the supply voltage Vs.
The driver stage is coupled directly to the output transistors ; the col-
lector resistor Rs of the driver tran sistor is connected to the unearthed
end of the load. In this way the current through resistor R s is reduced ,
because the voltage across R s is now the base voltage of the output
transistor. The d.c. conditions of both the driver stage and the output
stage are stabilized against temperature variations by applying d .c.
feedback from the emitters of the output transistors to the base of the
driver transistor by resistor R 1 , and by incorporating emitter resistors.
To minimise cross-over distortion , the output tran sistors mu st be
Fig. 3.2. Basic circuit of a complementary push-pull output stage. biased to a n emitter quiescent current of a few milliami:;eres. If both
output transistors had the same polarity, this quiescent current could be
obtained by a pplying a small negative base voltage with respect to the
emitter voltage. Since, however, the transistors have opposite polarity,
a positive and a negative base voltage with respect to the emitter voltage
are required. These base voltages are obtained by means of resistor R 3
which is inserted in the collector circuit of the driver transistor TR i ·
Some asymmetry is introduced , and to minimise this, resistor R 3 must
be given a low value.
The operation of the circuit is similar to that of Fig. 3.2. During the
positive half cycles the n-p-n transistor conducts, thus lowering the
value of Vi , and during negative half cycles the p-n-p transistor con-
ducts, raising the value of Vi . The variations of Vi are transferred to the
Cl
load via C 3 .
. ~
Vs As illustrated by Fig. 3.4, the a.c. excursions of the emitters of the
input +
output transistors are limited by several factors:

I. the peak value of the negative excursion at the onset of clipping,

neglecting the voltage drop across C 3 is
1
Fig. 3.3. Practical circuit arrangement of complementary output stage.

62 63
-
3.2.3 QUASI- COM PLEMENTARY SYMMETRY

~~~~:;., VeEJ
Complementary transistors for large power outputs are rather difficult
to make economically. Therefore in high-power amplifiers complemen-
v., tary transistors are mostly used only in the driver stage, to be followed
in the output stage by transi sto rs of the same polarity.
Fig. 3.5 shows the n-p-n output tran sistors in a push-pull configur-
ation, driven by a complementary pair. Thi s circuit configuration is
known as quasi-complementary symmetry.

RI

-v
Fig. 3.4. £miller voltages of a complementary pair of transistors.

2. the peak value of the positive excursion at the onset of clipping is:
VRL = V1 - VR7 - VBE3 - VcEKt - VR4 •
where VcEKt and VcEKz are the knee voltages of transistors TR 1
and TR 2 respectively.
It can be seen that if the voltage at the emitter junction of the output
transistors under no-signal conditions is equal to V 1 = 1 Vs , the maxi-
mum permissible excursions are no longer equal in both directions.
To ensure equal voltage excursions in both directions, point Q should
be biased to a potential V2 which neutralises the differences in potential,
so that V 2 becomes:
Fig. 3.5. Push-pull output stage using two 11-p-11 transistors driven by a complementary
V2 = t [Vs - VcEK2 + VBE3 + VcEKt + VR4]. push-pull circuit (quasi-complementary symmetry).

From Fig. 3.4 it can also be seen that no clipping will occur as a result
of the knee voltage of the n-p-n transistor TR 3 being exceeded, since:
VcEK3 < VR4 + VcEKt + VBE3· 3.3 Survey of Transistors for Power Amplifiers
In practice, resistor R 3 is a preset potentiometer with its slider connected 3.3. l TRANSISTORS FOR BATTERY-OPERATED EQUIPMENT
to one end, and adjustment of the quiescent current is a simple operation.
A thermistor may be connected across resistor R 3 to compensate for In a complementary symmetry stage the power output P 0 is determined
the temperature dependence of the quiescent current. by the pea k value of the collector current and the pea k output voltage.

65
64
-
From Fig. 3.4 the output voltage is: Germanium transistors are thus preferred for use in the output stages of
battery-operated equipment.
Vo = ! [Vs - VcEK2 - VR6 - VR7 - VBE3 - VcEKl - VR4], Referring to Table 3.1 and comparing the AC 127 / l 28 with the AC 187 /
and it follows that the smaller V 8 n becomes the greater the output vol- 188 it can be seen that the latter pair gives greater power output (up to
tage a nd consequently the power output. about 3 W) . If this level of power output is not required then obviously
Comparing silicon (V8 E = 650 mY at 5 mA) with germanium (an the AC127/ 128 complementary transistors may be used. However,
AC188 has VBE = 115 to 145 mY at 5 mA) it is clear that the lower there is a drawback in doing so. When complementary output transistors
value of base/emitter voltage of germanium transistors is preferred are used , direct coupling is employed from the driver stage. This means
and a higher efficiency results, entailing a longer battery life. For bat-
tery-operated equipment germanium complementary output transistors
are therefore preferred.
I,
But there is another reason for preferring germanium complementary !qui.scoot I / Si

transistors in battery-operated equipment. ln any complementary stage 100 -- - - ------- ---- - ----- -- !
Ge 11s
0
c/1
a small current flows under no-signal conditions to prevent cross-over I I1
distortion, and the d.c. setting is usually achieved by a small resistor 90°c /
0
2s c / I

connected between the bases of the complementary output transistors. 20 - - - - - 1I I I

/1 I I
As both germanium and silicon have a negative temperature coefficient 10
/ 1 I I
effect, the base/emitter voltage V BE decreasing with rise in temperature I I I I
I I I I
at about 2 mY/°C , the quiescent current is a function of the junction 3 ---- ~ ------ - - ------r- -
temperature of the transistors. / I / I
,/ I -/ I
In Class-B output stages the junction temperature depends on the a.c. 0
:-
_.,.
constant -l
---

excursion, and so does the dissipation. Maximum dissipation occurs at

approximately two-thirds of the sine-wave drive. Assuming that the I
- •- - - - - - I f - - - - cons tant - - - --.!

maximum junction temperature is reached, with germanium transistors

Fig. 3.6. Comparison between the quiescent currents of germanium and silicon transistors.
the value of VBE would decrease to:

VBE = (90-25) x 2 = 130mV.

while for silicon transistors V BE would decrease to: that the quiescent current setting of the driver stage must be such that
the driver can deliver the base current of the output stage when the latter
V8 E = (175 - 25) x 2 = 300 mY.
is equipped with minimum gain transistors. Hence the higher the mini-
With germanium transistors the quiescent current could rise, for example mum gain of the output transistors, the lower the quiescent current
from 3 to 20 mA, but with silicon a quiescent current of JOO mA , or setting of the driver stage will be. Consequently battery life will be pro-
even higher, could be achieved. This is shown in Fig. 3.6. longed.
It is obvious that in both cases the output stage .must remain stable. The robust AD161 / 162 transistors also feature high gain. This feature ,
With germanium transistors it is only a matter of selecting the correct- combined with a low knee voltage and a low base/emitter voltage makes
valued resistors, whereas with silicon transistors this method would them almost ideal for car radio applications. However, the breakdown
seriou'sly reduce the power output. As the latter is usually unacceptable, voltage level is such that in mains-powered applications attractive
stabilizing diodes have to be employed, instead of less expensive resistors. power output with good economy can be achieved. See Circuits 21 and 30.

66 67
 1

0\ Table 3.1. Germanium Complementary Transistors

00

' type ratings characteristics at T; = 25 °C

enve- R,,, j - c
n-p-n I p-n-p lope VcBo I VcEo
(V) (V)
I l cM
(A)
I T; max
(OC)
(oC/ W) hFEat l c
(A)
I VCEK max at l e llCBO max at Vc:B I fr typ
(V) (A) (µA) (V) (MHz)

ACl27

ACl28
TO- I 32
12

16
0.5

2.0
90
110

40
40-175

60- 175
0.3
1.0

0.6
0.5

1.0
:iOl
0
10
2.5

1.5

AC l 87 0 .8 100 5.0
TO -I 25 15 2.0 90 40 100-500 0.3 1.0 25
AC I88 0.6 200 1.5

ADl6 1

AD l 62
similar
to
T0-66
32rf.O 90
4 .5
R,h i- mb 80-320 0.5
0 .6

0.4
1.0
500

200
32

32
3.0

1. 5

Table 3.2. Silicon Dri ver Transistors

type ratings characteristics at Ti = 25 °C
envelope Ro, j - mb
VCBO VcEo feM Tj ma x (oC/ W) hFE at l'CBO max frtyp
n-p-n/ p-n-p (oC)
(V) (V) (A) l e = 0·15 A (nA) (MHz)
I I I I
BD I35/ 136 T0-126 I 45 I 45 I l ·5 I 125 i
I
IO I 40-250 I 100 I 250/7 5
BDI37/ 138 I T0- 126 60 I 60 I I ·5 I 125 I JO I 40-160 I 100 I 250/75
BDl39/1 40 I T0- 126 80 I ·5 125 IO \ 40- 160 JOO I 250/7 5

; ~11

Tab le 3.3. S ilicon Po wer Transistors

characteristics at Ti = 25 °C unless otherwise

type ratings
stated
-
Rrh j - mb
envelope (oC/ W) leBomaxat VcB
Ve1;o VeEO VCER I CM Tj max hFE at le fhfe fr
n-p-n (OC) (atTi = 200°C) (MHz)
(V) (V) (V) (A) (A) (kHz)
(mA) (Y)

BD l l5
I T0-39 I 245 180 I 245 I 0.2 200 I 12.5 > 22 I 0 .05 0.55 I 200 I - 145
typ.
I I I
T0-3 45 1.5 3 2 -
BD l 81 55
55 I 10 I 200 I I 20-70 I 45 1 15 min. I
20 typ .
I I I I I
BDl82 T0-3 70
60 I 70 I 15
I 200 I 1.5
I 20-70 I 4 5 60 15 min. I -

I I I I I 20 typ.

BD l83 T0-3 85
80 I 85 I 15 200 1.5
I 20-70 I 3 5 80 15 min. I -
I I I I I
20 typ .

°'
'Ci
-
3.3.2 SILICON DRIVER AND POWER TRANSISTORS On the magnetic pick-up position the input sensitivity is high and the
For higher power outputs n-p-n /p-n-p matched pairs are as attractive overall voltage gain from the pre-amplifier input to the output of the
as they are for lower power outputs. However, germanium transistors power amplifier may well be of the order of 50000. It is therefore essen-
have higher leakage currents than silicon and only metal can be used tial to keep the output separated and screened from the input, otherwise
as the case material. Silicon transistors are therefore preferred for high instability may occur. The magnetic field of the mains transformer may
power outputs, but at high power outputs it is rather costly to have cause hum , so the transformer should be as remote as possible from the
n-p-n transistors with a well-matched p-n-p counterpart. The quasi- input.
complcmentary configuration, however, combines the advantages of
rugged power transistors of the same polarity in the output stage with
3.4.2 EARTHING
economy of design.
The power output for a given loudspeaker impedance determines the Currents of several amps circulate in the power supply and output
supply voltage. The driver transistors listed in Table 3.2 may have rather stages. No wiring carrying these currents should be included in the input
high currents and breakdown voltages , and are therefore suitable for circuit, otherwise hum or instability due to the low but nevertheless
driving the output transistors over a wide range of loudspeaker im- significant resistance of the wires will result. The paths of currents in
pedances and power outputs. the output stage and power supply are shown in Fig. 3. 7.
It is a prerequisite of high power output transistors that they are
rugged. This ruggedness has been economically achieved at a slight
expense in the transition frequency. The resulting compromise makes
the BD181 , BD182 and BDl83 ideal output transistors.
In order to achieve a good power bandwidth care should be taken to
ensure that the impedance at the base side of the output transistors is low. +
Examples of this can be seen in the 15 W, 20 W , 25 W and 40 W ampli-
fiers, Circuits 22, 23, 24 and 25 respectively. For use in the 40 W amplifier
when an unstabilized supply is employed, the BD 183 replaces the BD 182.
The BD183 transisto r has a hig;1er value of Vcw but its characteristics
A 0
are otherwise very similar. The BD 115 high voltage transistor is not only
of interest in applications involving a high supply voltage , but is well-
Fig. 3.7. Current paths in the power supply and output stage.
suited to video as well as audio circuits. The characteristics of the silicon
power transistors are given in Table 3.3.

Between points A and B there is a voltage due to TR 1 ; between B and C

3.4 Practical Layout of Audio Circuitry
a voltage due to TR 2 ; and between C and D a voltage due to the power
supply. The input can be earthed at A, but not at B, C or D. Jn practice
3.4.1 G ENE RAL PRIN CIPLES
A , B, C and D would be combined at one common earthing point.
The circuits described in this book are capable of high quality reproduc- The earthing arrangements for stereo amplifiers are considerably
tion only if they are carefully constructed. Of special importance are the more complex than for mono , since the single power supply and common
relative positions of input , output a nd power supply, including the earth for the signal inputs make it more difficult to avoid earth loops.
earthing and screening arrangements. The arrangement recommended for the 15 W, 25 Wand 40 W amplifiers

70 71
described in this book is shown in Fig. 3.8 . The arrangement given applies One exception to those recommendations which a pplies to the 40 W
directly to the 40 W amplifier in which the output transistors are separ- amplifier, is that if solid conductors of I mm dia, or greater, (22 SWG)
ately mounted. are employed for the speaker leads and the power wiring for the output
It will be noted that the input sockets are earthed at the magnetic transistors, then these may be connected directly to the printed-wiring
pick-up input, since this is the most sensitive. The negative supply rails boa rd of the output stage, in stead of being taken to the power supply
for the pre-amplifier must be connected to the common earth at the unit as they would have to be if thin stranded flexible wire was used , i.e.
pick-up input and a heavy guage conductor used to connect this point 3 to 1 and 4 to 2 in Fig. 3.8.
to the power supply. The output transistors and speakers whose leads
carry several amperes should be connected directly to a common earth
3.4.3 STRA y FIELDS

The input loop of a pre-a mplifier is shown in Fig. 3.9. Thi s loop should
be as small as possible to prevent hum pick-up and it may also be ne-
cessary to screen the mains transformer. The magnetic field associated
with the currents in the output stage may occasionally cause trouble.

pick-up

tuner

tape

microphone

Fig. 3.9. Input loop of a pre-amplifier.

R channel

The remedy is to keep the output loops as small as possi ble, for example,
Fig. 3.8. Earthing arrangements for a stereo amplifier. Numbers in circles refer to con- by running the emitter and collector leads close together to minimise
nections in Fig. 2.34. radiation. The input selector switch may a lso need screening.

at the power supply negative terminal. Similarly, the positive lines should 3.4.4 POWER SUPPLI ES

be individually connected to the power supply positive terminal. The 1n addition to taking precautions against stray fields it is essential that
R-channel pre-amplifier positive supply may be obtained directly from the power supply, as well the conductors, has a low resistance. If the
the L-channel , the latter incorporating the smoothing components, intern al resista nce of the power supply is high, the regulation will be
but a gain the connecting leads must be kept short. Heavy guage con- poor and the output voltage may vary considerably with the signal.
necting wire should be u sed throughout. Th,i s, in turn, may give rise to insta bility.

72 73
3.5 Economical Transistor Power Amplifiers consisting of a low voltage transformer with a single rectifier diode and
reservoir capacitor common to both channels. The bias circuits for the
3.5.1 CIRCUIT 17 - ONE WATT STEREO PICK-UP AMPLIFIER input transi stors of each channel are decoupled by a common RC filter
network. The power transformer has a 6.3 V, I A secondary winding, its
Performance specification, each channel: core size depending upon the ratio of r.m .s. signal power to music power
nominal power output 900 mW into 8 Q load required. The supply voltage to the transistors must not exceed 9 V
sensitivity (1000 Hz) for P 0 = 900 mW 600 mV because, since this is a low-cost circuit, no temperature compensating
from IOOO pF source
110 to 11 500 Hz at max. volume resistors are used in the output stage. A higher supply voltage and/or a
frequency response (- 3 dB)
tone control - 12 dB at 10000 Hz lower speaker impedance will necessitate the use of an NTC resistor
nominal supply voltage 9V connected between the bases of the output transistors and also the use
current consumption at P0 = 900 mW 160 mA of emitter resistors.
Ganged toned controls, with separate or dual-concentric volume
This circuit has been selected as an example of a design that uses the controls for each channel are recommended. Jn each case these may be
absolute minimum number of components yet ensures good performance. of the linear type.
Four transi stors are employed in a direct coupled circuit with a comple-
mentary symmetry output stage. Feedback is used to achieve a high

t o other channel

1"1Jl
if'
BO µF
~ (2 5 VI 15001
µF +
(16VJ ,..
IG
470 kJl
input o-C::J-.--------,

volume
) "1Jl

Bil

Fig. 3.10. One watt stereo pick-up amplifier circuit. Nominal supply voltage to transistors
must not exceed 9 V.
Photograph of Two Watt Stereo
A mplifier described overleaf
input impedance, and a good low frequency response is obtained with a
capab tive source such as a ceramic pick-up.
Only one channel is shown in Fig. 3.10, the simple power supply

74 75
3.5.2 CIRCUIT 18 - Two WATT STEREO PICK-UP/ RADIO AMPLIFIER

Performance specification, each channel: ldB) t----!---+-+-+-+++++----t---+--+-+-+++++---+---+-i-++-+++t----!---+--+-++++H

nominal power output 2 w into 8 n load

sens itivity (I 000 Hz) for P 0 = 2 W
pick-up (I 000 pF) 320 mV
radio 70 mY
input impedance 130 kO -51----!--+--+-+-+H~'/~-+--+-+-+->-++<+-'~,r+--+-i-++-+++l----!-~''l-+-++++H

frequency respon se (- 3 dB) 85 to 25000 Hz ''

total harmonic distortion at P 0 = 2 W 2% "
-10 t----+---+--H--'ltt+t---1--1--+-t+t+tt---+--+--i'<H-tt+t---+---+-++++t~
nominal supply voltage .15 v
current consumption at P 0 = 2 W 230 mA
-1S t----+---+--H-++t+t---1---t--+-t+t+tt---t--+--i-t-+-+ttl---t---+--+-+++t~

This circuit is intended for use with stereo systems where a maximum
power output of 2 W per channel is sufficient. Sensitivity is such that
most types of ceramic pick-up and radio input level s can be accomodated.
Fig. 3.12. Tone control characteristics of two watt stereo amplifier.
Curve J: max imum bass, treble .flat;
+ +14V +ISV Curve 2: bass /fat, treble /fat;

I 80 µF
(25V)
Curve 3: bass /fat, max imum treble cut.

dance of the amplifier is 130 kQ. The ceramic pick-up cartridge is capaci-
I M.n
tively loaded so that the high impedance of the cartridge at low frequency
can be matched into a low impedance without loss of bass. Improved
noise performance also results from the lowered input source impedance
as seen by the input transistor.
Bass, treble and volume controls are located in the collector circuit of
the input transistor , which simplifies the front panel wiring. Overload
811 level is 25 dB above the input required for full amplifier output; this level
is more than adequate to enable the volume control to be used at the
output of the BC149, instead of the more usual position at the input.
The remainder of the amplifier is a conventional direct coupled am-
plifier with a complementary symmetry output stage using an AC187/
AC 188 matched pair of transistors. A negative temperature coefficient
Fig. 3. 11. Two watt stereo pick-up/radio ampl(/ier circuit. (NTC) resistor is used across the quiescent current stabilizing resistor
to eliminate cross-over distortion at extremes of temperature.
Feedback from the output stage is applied to the emitter of the BCI 77.
Tone controls are of the "cut" type, but bass boost derived from feed- This is typically 12 dB at l 000 Hz. The d.c. feedback stabilizes the mid-point
back' gives the apparent effect that both bass boost and cut are avai lable. voltage of the output stage, while further frequency selective a.c. feed-
Fig. 3.11 shows the circuit of the amplifier. The effective input impe- back is applied across the 3.6 kQ resistor, resulting in a reduction in

76 77
feedback in the JOO Hz to 150 Hz region. This gives an effective bass
boost of 6 dB at 120 Hz. Since the bass cut tone control has an ap-
proximate range of 12 dB at 130 Hz, the overall response is equivalent
to a bass control variation of ± 6 dB at 120 Hz, relative to the 1000 Hz
response.
The 3.3 nF capacitor connected between the collector and base of the
driver transistor provides high frequency feedback to reduce the h.f.
loop gain of the amplifier.
The 100 0 preset control connected between the two channels pro-
vides a ± 4 dB variation in gain and thus can be used to align the centre-
point of the balance control.
Heat-sinks are required only for the output transistors, approximately
4 sq in (25.8 cm 2 ) of 1/ 16 in (1.6 mm) thick aluminium being recom-
mended for each one. A multiple heat-sink carrying all four output tran-
sistors of a stereo version of this amplifier shou ld therefore have an area
of at least 16 sq in (103.2 cm 2 ).

l~945
10

- · - 10kHz
-1kHz
dt ot
(
0
fo)
- - 100Hz I
I
I
I
I
I
I
I

_.... ~· /
0
0
- L... -- -- -- - I/".'. I//

Fig. 3. 13. Total distortion of two watt stereo amplifier.

Fig. 3. 14. Printed-wiring board for two wall stereo amplifier.

The tone control characteristics of the amplifier are shown in Fig. 3.12,
and the total distortion at three different frequencies for power outputs
up td 2 W is shown in Fig. 3.13. For stereo applications, the printed-
wiring board shown in Fig. 3.14 may be used.

78 79
 , ....

-
7B9452
100
3.5.3 CIRCUIT 19 - FOUR WATT MONO CAR RADIO AMPLIFIER
v,
Performance specificatio n : Im VJ \

nomina l power output 4 w into 4 n load 7S

\
sensitivity (I 000 Hz) for 'I.
1-.
Po = 4W 48 mV ......
P0 = 50 mW 5 mV i-c 1
so _,_
input impedance 10 k!1 ,.._
frequency response (- 3 dB) 200 to 20000 Hz ,....
tota l harmonic distort ion a t P0 = 4 W
nomina l supply vo ltage
10 %
14 v
'I\
.....
2S

(oo...
The circuit of a direct coupled 4 W Class-B audio amplifier for use in a ~
2
car radio is shown 111 Fig. 3. 15. The amplifier uses an ADl6 1/ ADl62 0
-so 0 so Tamb f°CI 100

l
+14V

12SµF
11sv1 -~
16µF
+
F 82kfi
1n

r'1
I V,.skn I

K
- t'
ion

AC12B
TR2
Fig. 3.16. Input sensitivity offour watt car radio amplifier at various ambient temperatures.
Curve 1 for maximum power output at 10 % distortion; curve 2 for an output power of
50mW.

fv BC148 IV A0161
+:;:-0;
l16V I J ~ TRI ~ TR3

B2kfi J +400
: , µF 10ll
0.47fi
.__J T l10VJ I (w ire-
wound)
4~~µFl16VJ
~

r:
H59t..55
+" ' 8

IS~ v (
Lt son O.•m
(w ire -
wound)
P,
2.2kfi

22on
-«.(
-t'
A0162
TR4
4fi
IWI

0
7lU45!
~~
4
Fig. 3. 15. Circuit of four watt 1110110 car radio amplifier. A II the transistors are mounted L... 1' d: 10°/o
on one heat-sink which has a thermal resistance ofRrh h- a .;;; 5.5 °C/ W. ...... '1..
.
complementary output pair, an AC 128 d river transistor, and a BC148 or l .1 I
2
, L-~ ,.... d:5°/o '\
BCI 08 transistor as a first stage amp lifier. v ..... I'\
I/ , I"' I'-
The amplifier is a conventional four-transistor circuit with one ex- [/ r...
ception: the decoupling capacitor in the input stage is returned to the 0
-so 0 so Tamb 1°C) 100
emitter of the BC l48 , instead of to chass is. This neutralizes the effect of
r ipple a t the supply line and increases the input impedance.
Fig. 3.17. Power output at two distortion levels as a function of the ambient temperature.
The amplifier has been designed for operation in ambient temperatures
from ' 20 to 70 °C, but its performance characteristics over a wide range
of temperat ures are given in Figs. 3.16 to 3.1 8.

80 81
.jl :

-
 3.5.4 CIRCUIT 20 - FIVE/ TEN WATT Low-COST AMPLIFIER

Performance specification:
nominal power output 5 w into 8 n load
into 4 n lo ad
20
10 w
sensitivity (IOOO Hz) for P0 = 10 W
pick-up ( IOOO pF) 155 mV
15 radio 48 mV
input imped a nce 150 kn
84°C 75°C 55"c
I 65°c /I frequency response (- 3 dB) 40 to 30000 Hz
I I '31 • tota l ha rmonic distortion at P 0 = 10 W 5%
10 I I nominal supply vol tage 24 v
j I I
I~ I I
/ I I
v I I
/ v /// This amplifier is a low-cost transformerless amplifier which will deliver
~ ,, / // l 0 w into an 4 n load , or 5 w into an 8 n load at a total harmonic
- ~ "/
distortion of less than 5 % at 1000 Hz. It is intended for radio tuner and
ceramic pick-up inputs. The circuit diagram is shown in Fig. 3.19.
10
-1 P0 (W) 10 The pre-amplifier is a BCl49 transistor operating in the common-
emitter mode. Its input impedance is made fairly low to reduce noise
and make the stage more independent of component tolerances, parti-
20 cu larly in the pick-up position where the transistor base is voltage driven
using a capacitive divider to match a 1000 pF pick-up cart rid ge into a
150 kQ input impedance.
15 A BCI48 n-p-n silicon transistor is employed in the pre-driver stage,
-59°C -51 °C -42•c -32°C
I -21°C its bias point determining the mid-point voltage of the output stage.
/ +3°C 1 - The driver, TR 3 , is an AC128 high gain p-n-p germanium transistor
I 1 •1s 0 c
I I I operating in the common -emitter mode.
I II II
I II The output stage is directly coupled to the driver transistor and
I /, consists of the germanium transistor pair AD161 / AD162 operated in
/ J //
// C lass-B complementary symmetry. The quiescent current of the output
I/ /,R'l'
transistors is determined by R 20 and R 22 . Resistor R 20 is a 15 Q NTC
...../
used for thermally stabi li zing the quiescent current. For best results,
it should be mounted on the outp ut transistor heat-sink. The qui escent
P0 (W) 10 c urrent shou ld be set at 15 mA by means of R 22 .
Diode BY 126 is used across the output to prevent the possibility of
Fig. 3.18. Distortion as a function of output power at different ambient temperatures. transistor breakdown due to negative voltage peaks produced by the
inductance of the speaker on transients.
Treble cut is achieved by a n RC network shunting the ou tput of the
pre-amplifier stage. Resistor R 8 controls the time-constant of the net-
work, thus determining the degree of high frequency attenuat ion. At

82 83
 7 Z\9l..56
10

I
d tot
fO/o)
/ 10 0Hz
1

)I j
I
+ Cll.
BOµF ............: ./'
10kt/z

- --
i2 5 V)

RJ R5 . /:::::.-
/
J

1··
rad io 1Mn 56 kn
µF ~ --/ / /
1kH z

----
£..VI
C5
/ _... /

.,~
SOk!l. volume
0
_,.. 7

C4 0 10 P0 IWJ 15
10 nF ,..

Fig. 3. 20 . Total distortion of fi ve/ten watt amplifier.

r
R2
1Bkn
R7 RB R13
a2on lOOkn 1sn
an 1sw ·
i.n11ow

7259457
20

Fig. 3. 19. Circuit of fiv e/ten watt low-cost amplifier.

ldB I

10
F
1,
"'
I
I 2 '
~ 3 -......_ ....
/
maximum treble cut, the network introduces 20 dB attenuation at I
8000 Hz /
I
"' '\.
I
Bass boost is provided by a bridged-T feedback network. When the - 10
I '\
~
time-constants of the two branches are equal, the attenuation of the whole
~
network is independent of frequency. When the time-constant of the
'
shunt branch increases, by reducing R 1 7 , the attenuation of the lower -20
10 10' 10' fl Hz J 10'
frequencies increases. Since this network is in the feedback path, the
lower frequencies are boosted. With the control in the maximum bass Fig. 3.2 1. Frequency response of th e fiv e/ ten watt amplifier.
boost position, a boost of about 10 dB at 100 Hz is achieved. Cur ve 1: max imum bass boost, treble flat ;
' The total distortion as a function of output power is shown in Fig. 3.20 Curve 2: bass flat, treble flat;
and the frequency characteristic is shown in Fig. 3.21. Curve 3: bass flat, maximum treble cut.

84 85
3.6 High Fidelity Power Amplifiers

3.6.I CIRCUIT 21 - 8 WATT H1-F1 AMPLIFIER USING ADl61 /ADl62 >

0
N ~
~-~Bi-----IS~
I
O UTPUT TRANSISTORS
Performance specification:
nominal power o utput 8 w into 4 n load
sens iti vity ( I 000 Hz) for
P0 = 8 W 85 mV
P 0 = 50 mW 6.7 mY
input impedance 330 k!.1
frequency response (- 3 dB) 14 to 36000 Hz
total harmonic distortion at P0 = 8 W 1%
unweighted signa l/noise ratio (ref. P 0 = 8 W) 76 dB
weighted signa l/ noise ratio (ref. P 0 = 8 W) 86 dB
internal resistance at o utput socket 0.25 n
damping factor with 4 n speaker 16
nomi na l suppl y voltage 20 v
c urren t co nsumption a t P 0 = 8 W 0.85 A

The block diagram of a n eight watt hi-fi mono/stereo a mplifier is shown

in Fig. 3.22. Seven transistors a re employed in the circuit which has a
complementary symmetry output stage. A high input impedance is

"'
M
0:

complementary
symmetry
output stage

input •n
aw
85mV
330kn

m
0:

Fig. 3.22. Block diagram of eight wall hi-ft amplifier.

achieved by means of feedback and the amplifier meets the European

Standard for high fidelity sound reproduction.
Jn the circuit diagram shown in Fig. 3.23, alternative networks may
be ' connected between the emitter of TR 1 and the collector of TR 2 ,
depending upon whether the a mplifier is required for mono or stereo use.

86 87
 For stereo applications, the balance control network, shown separately,
is connected between the emitters of the first pre-amplifiers of each
channel.
The frequency characteristics of the amplifier are shown in Fig. 3.24.
With the bass and treble controls flat , the response is linear from I00 Hz
to 6000 Hz, and is within J dB of the response at 1000 Hz from 32 Hz
to 18 000 Hz. It is only 2 dB down at 20 Hz and J .2 dB down at 20000 Hz.
f---+-4---+-!-+-!~ - - 40Hz
The total distortion at 40 Hz, 1000 Hz, and 12 500 Hz is shown in dtot f---+-4---+-!-+-!~ - - 1kHz
Fig. 3.25. For I % total harmonic distortion the output power at 40 Hz ( o/o) •- 12 .Sk Hz t--+-+-+-++<f----1---4--'---'-'--~

is 6.1 W, at 1000 Hz it is 8. 1 Wand at 12 500 Hz it is 7.8 W. The heat-

sink for the output stage shou ld be at least 11 5 cm 2 for 2 mm thick
aluminium.

20
/'
ldB I
"
10 "' ,"\. v
1 I/
I' /
I'-
2 -/ Fig. 3.25. Total distortion of the eight wait hi-fi amplifier.
.- ......_ ~

v !',.
I/
/
V3
-10 ~
/
/ 5
v

-20
10 10 ' 10 ' f(Hzl 10'

Fig. 3.24. Frequency response of !he eight wait hi-fi amplifier.

Curve I: maximum bass boos/;

Cur ve 2: bass /fell, treble /fell;
Curve 3: max imum bass cut;
Curve 4: maximum treble boost;
Cur ve 5: maximum treble cut.

88 89

l
3.6.2 CIRCUIT 22 - 15 WATT HI-FI AMPLIFIER USING BDl8l OUTPUT RS O.BA
,------,--if--C::::J---r--------r--- - r--t--£:3---0 + 3BV
TRANSISTORS +
C2
SS kfl. fuse
T 6.4.~F
Perfor ance specification: ~ i 2SV )

15 w into 8 n load
Rl Rt.
nomin a l power output ISO kfl. 3.Jkfl.
sensitivity ( I 000 Hz) for P 0 = 15 W 350 mV
input impedance 150 kn
frequency response (- 0.5 dB) 14 to 100000 Hz v.
+ C10
total harmonic distortion at P 0 = 15 W 0.1 % 2SOO "F
intermodulation distortion at P 0 = 15 W 0.5 % i2S VI

(measured with f 1 = 250 Hz a nd

f 2 = 8 kHz where Vfl : V12 = 4 :1)
unweighted signal/noi se rat io (ref. P 0 = 50 mW) 76dB
Bfl.
internal resistance at output soc ket o.o5 n
damping factor with 8 n speake r 160
voltage feedback factor 350
nominal supply voltage 38 v
current consumption at P o = 15 W 0.625 A
1.Bkl1

This amplifier fully complies with the European Standard for high Fig. 3.27. Circuit of 15 watt hi-ft amplifier.
fidelity sound reproduction. It will deliver 15 W r.m.s. output power
into an 8 Q load, Silicon transistors have been employed throughout
and the amplifier uses BDl81 power transistors in aquasi-complemen-
is employed in the pre-amplifier stage. Use of this high current gain
tary symmetry single-ended Class-B push-pull configuration. The block
transistor enables large amounts of feedback to be employed by means
diagram given in Fig. 3.26 shows the general arrangement.. .
of resistors R 3 , R 4 and R 15 . The transistor has an operating current
When used with the recommended Universal Pre-Amplifier descnbed
of 0.5 mA. This stage also functions as a mid-point voltage stabili zer
in Sub-section 2.3.8. the combination is highly suitable for all 15 W high
and for good stabilization R 4 must not have too high a value. On the
fidelity equipments, particularly for 2 x 15 W stereo. .
other hand, for a high a.c. feedback factor from the speaker R 4 must be
The circuit diagram is given in Fig. 3.27. A BC! 58B p-n-p transistor
as high as possible with respect to R 15 , since R 4 is in parallel with R, 5 •
Jn this circuit a value of 3.3 kQ for R 4 has been chosen. Due to the large
feedback factor, the input impedance of the amplifier is equal to the
value of R 1' that is 150 kQ.
complementary push-pull The Class-A pre-driver stage uses an n-p-n BCl47B which operates
symmetry output
driver stage with a d.c. collector current of 4 mA. This is because the complemen-
tary driver transistors have a peak base current of 2.5 mA and there is a
loss of I.I mA via resistor R 10 • The maximum dissipation of the pre-
input sn driver transistor is 89 mW. To diminish the high frequency behaviour
3SOmV 1SW
1SOk0
of this transistor an additional capacitance of 27 pF is connected bet-
ween the collector and base of TR 2 •
The complementary driver transistors TR 3 and TR 4 are BO 135 and
Fig. 3.26. Block diagram of 15 watt hi-ft amplifier. BDI 36 respectively. These act as a phase inverter stage and have to

90 91
 input from
pr•-ampl.

60181 60181

0.8A
2500µF

l+J 1-1
power power
supply supply

Fig. 3.28. Primed- wiring board for 15 watt hi-ft amplifier.

Fig. 3.29. Component layout of printed-wiring board f or I 5 watt hi-fi amplifier.

92 93
deliver the drive current for the output transistors and supply the cur- To compensate the quiescent currents of the complementary drivers
rents through the resistors R 16 a nd R 1 9 connected between the base and and output tra nsistors against temperature and supply voltage variations,
emitter of each of the output transistors. The resistance value of R 1 6 and the base/emitter voltages of three transistors mu st be compensated .
R 1 9 cannot be too high or the relatively low frequency behaviour of This can be done by three silicon diodes or by a transistor circuit which
the BD 181 output tran sistors will be affected (see Sub-section 3.3.2). a cts as a kind of zener diode. Due to its superior characteristics the
In this amplifier R 16 and R 19 have a value of 56 Q. transistor circuit has been incorporated in this a mplifier. The circuit
At peak output, the maximum current which TR 3 has to deliver is
118 mA ; TR 4 requires a somewhat lower current. ln series with the
bases of the driver transistors 470 Q resistors R 1 2 and R 13 are included
!dB I 1---+--+-r+t+tH--+--+--t-+++++t- +-+-+++-H+t--+--+-t-++++H
for the short-circuit condition. The maximum drive current for the driver
and output transistors is then limited when the output terminals are
short-circuited. R 12 and R 13 are also important to limit the operation
-1H--+--+-r+t+tH--+--+--+-++++++-+-+-+++-H+t--+--+-+-++++H
of the driver transistors in the overdrive condition when a complex load
impedance is connected to the output terminals. Typical quiescent cur-
-1 >+---+---+---t-+-t+++l--+--+-+-++++++--+-+-+-++-H+t--+--+-+-++++tt
rent for the driver transistors is about I 0 mA, and in the worst case their
dissipation becomes 310 mW.
- J f---+---+---t-+-t+++l--+--+-+-++++++--+-+-+-++-H+t--+--+-+-++++tt
The output stage uses two matched silicon power transistors BDJ81.
These n-p-n transistors ha ve a maximum power dissipation of 78 W at
- 4f---+---+--->-+++++1--+--+-+-++++++--+-+-+-++->+++--+--+-+-++++H
25 °C and a peak collector current of JO A. The quiescent current of the
output transistors is 40 mA, which can be adjusted by potentiometer Rs.
In the worst case the dissipa tion of each transistor is 6.5 W. The peak
output current 10 for P0 = 15 W into an 8 Q load R u is
Fig. 3.30. Frequency response of 15 waif hi-Ji amplifier. 0 dB = 6 dB below 15 watts.

1/ 2 Po
10 = V -- = 1.95 A.
RL comprises a plastic n-p-n transistor BCl48 , TR 7 , two resistors R 7 and
R 9 , and a pre-set potentiometer Rs with which the quiescent current can
The peak output voltage for 15 Wis 15.5 V. Resistors R 17 and R 1 s be adjusted. The nominal value of the collector/emitter voltage of TR 7
provide thermal stability. The volt age losses in the upper part of the is approximately 1.8 V.
amplifier are due to the voltage drop across R 1 7 , Vn E of TR 5 and VcEK Fig. 3.28 shows the printed wiring board for the 15 W amplifier and
of TR 3 and together these amount to 3.5 Y. Similarly, the voltage losses the component layout is shown in Fig. 3.29. lnterconnection of this
in the lower part are determined by the voltage drop across R 1 3 , VnE amplifier with the power supply unit and the Universal Pre-amplifier
of TR 4 , V cEK of TR 2 and the voltage drop across R i s· Again the total is has been dealt with in Section 3.4.
3.5 Y. To guarantee 15 W output power with minimum devices the sup- The frequency characteristic shown in Fig. 3.30 is linear from 20 Hz
ply voltage a t maximum power output must be equal to the sum of the to 70000 Hz and drops to - 0.5 dB with respect to the 1000 Hz response
total voltage losses and twice the peak output voltage excursion. This at 100 000 Hz.
becomes Total harmonic distortion as a function of output power at the three
frequencies 40 Hz, 1000 Hz and 12 500 Hz is shown in Fig. 3.31. Inter-
3.5, V + 3.5 V + 2 U5.5 V) = 38 V.
modulation distortion, measured with frequencies 250 Hz and 8000 Hz
The mid-point voltage VA is 19 Y. in the proportion 4: I , is 0.5 % at maximum output power.

94 95
 3.6.3 CIRCUIT 23 - 20 WATT HI-Fl AMPLIFIER USING BDl81 OUTP UT
1----+--<--+-+-+-+++< - - 40Hz
1kHz TRANSISTORS
d tot 1----+--<--+-+-+-++<~ - -- 12 .SkHz f-+++++l--+--+-+-+-H~
I 0 10 l 1---+--1--+-l-l-++l+--~~-+-++++++--nr+--+--+--+-t-+++1
Performance specification:
nominal power output 20 w into 5 n load
se nsi tivity (1000 Hz) for
Po = 20 W 2 10 mV
P0 = 50 mW II mV
input impedance 100 kO
frequency respo nse (- 1 dB) < 20 to > 20000 Hz
total harmo nic distortion at P 0 = 20 W 0. 1 %
un weighted signa l/ noise ratio (ref. P 0 = 50 mW) 66dB
weighted signa l/ noi se ratio (ref. P 0 = 50 mW) 81 dB
internal resistance a t ou tput socket 0.060
damping factor with 5 n spea ker 83
10 P, IWI nominal s uppl y vo ltage 40 v
Fig. 3.31. Total harmonic distortion measured with an 8 n load and a source resistance current cons umpti on a t P 0 = 20 W 1. 2 A
of I kn.
The block diagram of a 20 Watt hi-fi amplifier is shown In Fig. 3.33.
20
driver output
P,
IWI
18 pre - ampl if ier pre-drive r
input
210mV
[ r-11s n
IOOkO ~ 20 W
16

14
,,
Fig. 3.33. Block diagram of 20 watt hi-fi amplifier.
12

Nine transistors a nd two diodes are employed in the circuit which has a
10 quasi-complementary sy mmetry output stage. Two of the transistors,
together with the diodes, provide protection agai nst short circuiting
B of the output. The amplifier fully complies with the European Standard
10 10 ' 10 ' f IHzl 10'
for high fidelity sou nd reproduction.
Fig. 3.32. Power bandwidth of 15 watt hi-fi amplifier. Jn the circuit diagram shown in Fig. 3.34 the short circuit protection
network compri ses the transistors TR 4 , TR 5 a nd di odes D 1 a nd D 2 • .
The power bandwidth characteristic shown in Fig. 3.32 is for a total The operation of this circuit, which also applies to the 25 watt and 40 watt
di stortion of 1 %, measured with an 8 Q load and using a source impedan- amplifiers, is described fully on page I 08.
ce of 1 kQ. The frequency response and power bandwidth characteristics are shown
Protection against the short-circuiting of the speaker lead s is by means in Fig. 3.35. The frequency response is virtually linear from 20 Hz to
of a fast acting fuse of 0.8 A rating. No higher rating should be used. 20000 H z.
Eacli output power transistor requires a heat-sink with an effective area Total distortion at 40 Hz, 1000 Hz and 12 500 Hz is shown in Fig. 3.36.
of at least 12 cm 2 with a thickness of 2 mm . From 300 mW to 20 W o utput the distortion does not exceed 0.1 %-

96 97

l
 ldBl,___+--+-+-t-t-t++t--+--+-+-t-t-t++t--+--+--+-+-+-t+++--+--+--+-+-+-++H

-11---++-+-+-H-t++t--+--+--+-H-t++t--+--+--+-+-+-t+++--+--+--+-+-+-++H

-2f---l-+-+-++++++--+--l-++-t-++1-t--+--++-lr+1Htif----l-t-i-+H-1-i1
R7
270!l
-31--+--+-+-H-t++t--+--+--+-H-t++t--+--+--+-+-+-t+++--+--+--+-+-+-++H

-41--+--+-+-t-t-t-+t+--+--+-+-t-t-t-+t+--+--++-t+tti+--+--++-t+t-tt1

Fig. 3.35. Frequency response and power bandwidth characteristics of the 20 watt hifi-
amplifier.
Curve 1: amplitude/frequency , 0 dB = 6 dB belo w 31.2 W;
Curve 2: power/frequ ency, 0 dB = 3 1. 2 W .

S!l
1--t--+-+--<-+-t-t+<- - 40 Hz
1k Hz
dtot 1 - -t--+-+--<H-t-t+<- • - 12 .Sk Hz H-1++---+-H-+-++l++---+--iHH-+-++li-tj
(O/o) J--+-+--H-+-i++\--~~~-+-1++\--+--+-+-H-++++--+-+a+-H-++H

Fig. 3.34. Circuit diagram of 20 watt hi-ft amplifier. The output transistors are mounted
a 90 x 90 x 2 mm aluminium heat-sink.
0 11

~OL_-
, -L--'-.L..l..J...J...L.L10L_-,-L--'-.L..l..J...J...L.LL--L-..J.....L.JU...L.U.1~
0--'--P~
,-IW.L.J)u...~10 2

Fig. 3.36. Total dist ortion of the 20 watt hi-ft amplifier.

98 99
3.6.4 CIRCUIT 24 - 25 WATT H1-F1 AMPLIFIER USING BO 182 OUTPUT
TRANSISTORS RS H

Performance specification: + C2
T 2.S µF
nominal power output 25 w into 8 n load Jz!Slo.VJ

sensitivity ( I 000 Hz) for P 0 = 25 W 350 mY

input impedance 150 kO Rl
150 kfl
frequency response (- 1 dB) < 20 to 90000 Hz)
tota l harmonic distortion at P0 = 25 W 0. 1 %
intermodulation distortion at P 0 = 25 W 0.6 %
(measured with f 1 = 250 Hz and f 2 = 8 kH z
where Vn: Vf2 = 4 : I)
unweighted signal/ noise ratio (ref. P0 = 50 mW) 76dB
internal resistance at output socket o.o5 n
damping factor with 8 n speak er 160
voltage feedback factor 370 Bfl
nominal supply voltage 48 v
current consumption at P0 = 25 W 0.83 A

This amplifier complies fully with the European Standard for high RlS

fide lity sound reproduction and will deliver 25 W r.m.s. continuous 1.8 kfl

sine-wave power into an 8 Q load. Silicon transistors have again been

used throughout and the amplifier employs BD 182 power transistors in a Fig. 3.38. Circuit diagram of the 25 wait hi-fi amplifier.
quasi-complementary single-ended Class-B push-pull configuration.
driver output
0.5 mA . This stage also functions as the mid-point voltage stabilizer.
Due to the high feedback factor, the input impedance of the amplifier is
input
350mV
an equal to the value of R 1 , that is, 150 kQ. In this respect, and also the
25W
150kfi operation of the remainder of the circuit, this amplifier is very similar
to the 15 W amplifier previously described in Sub-section 3.6.2.
The Class-A pre-driver stage uses an p-n-p BD137 transistor TR 2 but a
BC147 may be employed if a stabi lized supply voltage is available.
Fig. 3.37. Block diagram of the 25 watt hi-fi amplifier.
The complementary driver transistors TR 3 and TR 4 are BDl 37 and
BD138 respectively. At peak output, the maximum current which TR 3
The block diagram given in Fig. 3.37 shows the general arrangement. has to deliver is 133 mA ; TR 4 passes a somewhat lower current. Resistors
When used in conjunction with the recommended Universal Pre- R, 2 and R 13 are again included to limit short-circuit and overdrive
amplifier described in Sub-section 2.3.8 the combination is very suitable conditions. Typical quiescent current for the driver transistors is about
for all 25 W high fidelity equipments, particularly 2 x 25 W stereo. 10 mA , and in the worst case their dissipation is 440 mW, so no heat-
The circuit diagram is given in Fig. 3.38. A BC 158B p-n-p transistor sinks are required for the driver transistors.
is employed in the pre-amplifier stage. Application of this high gain The output stage uses the robust BD 182 power transistors in a matched
transistor enables large d.c. and a.c. feedback to be used by means of pair. The quiescent current of the output pair is 40 mA, which can be
resistors R 3 , R 4 and R 1 5 • The transistor has an operating current of adjusted by potentiometer R 8 • With no protection circuit the dissipation

100 101
in the worse case for each output transistor is J0.3 W. The peak output is given by
current / 0 for P0 = 25 W into an 8 Q load is 2.5 A .
VcE = IR1 + (I + IB)R2 ,
The peak output voltage for 25 W is 20 V. Voltage losses in the upper
part of the output circuit are 3.5 V, and in the lower part are 4.5 V. To and
guarantee 25 W output power the supply voltage at maximum power VBE = IR1.
output must be equal to the sum of the total voltage losses and twice
the peak output voltage excursion. This becomes: The ratio VcE! VBE is given by:
3.5 V + 4.5 V + 2 (20 V) = 48 V.
The mid-point voltage VA is therefore 24 V. - - = - ------
VBE IR1
Since IB « I, even with a BC148 with the lowest gain, the ratio VcE! VBE
becomes:

-- = ----

This expression shows that a variation of VBE produces a variation in

stab ilizat ion
VcE depending on the ratio RcE!RBE> so
vol tage
( VcE )
Re£
RI

l
'-----L-~'.__. to bas e TR 4
VcE = - - ·LI VnE·
RnE

The temperature dependency of VcE is thus given by:

to class-A driver

Fig. 3.39. Quiescent current stabilization network of the 25 watt hi-fi amplifier.

This temperature dependency of Vc£ must be made equal to the V BE

To compensate the quiescent currents of the drivers and output temperature dependencies of TR 3 , TR 4 and TR 5 which have to be com -
transistors against temperature and supply voltage variations, the base/ pensated . V BE has a simi lar value (about 2 mV j°C) in each case, and so
emitter voltages of three transistors must be compensated. The circuit by arranging that the ratio RcE/RBE = 3, quiescent current stabilization
employed comprises a plastic n-p-n transistor BC148, TR7 , two resistors can be easily achieved. It will be seen from the circuit diagram in Fig. 3.38
R 7 and R 9 , and a pre-set potentiometer Rs with which the quiescent that R 9 = 2.2 kQ, Rs = 1 kQ and R 7 = I kQ. Rs may be adjusted to
current can be adjusted. The nominal value of the collector/emitter vol- make RBE = 1.4 kQ when , since Re£ = R 7 + Rs + R 9 = 4.2 kQ,
tage of TR 7 is approximately 1.8 V. their ratio is 3.
Operation of the quiescent current stabilization network which is Supply voltage variations will change the d.c. setting of the Class-A
common to the 15 W, 20 Wand 40 W amplifiers also, can be explained pre-driver stage, causing a small variation in the emitter/ base voltage
by reference to Fig. 3.39. In this circuit the stabilization voltage V cE of transistor TR 7 , but the quiescent current for the output transistors wi ll

102 103
 hardly be affected. The stabili zing transistor must not be mounted on the
24
Ic heat-sink of the output tran sistors otherwise over-compensation will
IAI 20
occur.
Short-circuiting of the output terminal s produces heavy currents
12
80182
upper stage
in the output a nd complementary driver transi stors. The immediate
effect of short-circuiting does not, in itself, present a problem . Although
the feedback fall s off and a very small input signal will cause a large
overdrive, the output transistors have an enormous margin before their
0.2 0.4 0.6 a.a 1 ratings are exceeded. Fig. 3.40 shows the short-circuited conditions of the
t {ms)
BD I 82's, from which the following values can be obtained:
24
Vee upper BDl 82 lowe r BDl 82
IVI 20 TR 5 TR 6
peak dissipation Peak co ll ecto r c u rrent 6A 15 A
16 =108 W
Pea k collector/emitte r vol tage 18 v 13 v
average dissipation
12 · BOW Duty cyc le 74 % 26 %
duty cycle =:.74°/o Peak power dissipat ion 108 w 195 w
Average power dissipation 80 w 51 w
4
Now the upper BD 182 should have a total thermal resistance of:
0.2 0.4 0.6 0.8 1
t(msl
Tj m ax - Tamb 200 - 50
R,,, 10 1 = - -·- - - - = l.9°CfW.
24 Pav 80
Ic
IAI 20
This cannot be realised, because the transistor already has a thermal
16 resistance from junction to case of 1.5 °C/W, so some form of short-
12
8018 2
low er stage
circuit limiting is essential for temperature reasons.
Two short-circuit protection networks are shown in Fig. 3.41. The
letters used to indicate connections correspond with those at appropriate
4
points on the circuit of Fig. 3.38. Consider Fig. 3.4l(a) where Protection
0 oL-~o-"-.2--'----'-0.4--'-'o.~
6 -'-'-0.....
.e _._-L-
1 Circuit I is shown. This circuit uses two transistors BC148 and BCJ 58
t(ms)
whjch are norm ally in the off condition. When a sufficiently high a.c.
Vee
24 current is flowing through R 17 and R 18 (Fig. 3.38), the base/emitter
IVI 20 vo ltage is becoming more positive for the BC148 and more negative
peak dissipation for the BC! 58. At a certain voltage level, depending upon the a.c. cur-
16 •195W
average dissipat ion
ren t through R 17 and R 18 and the base voltage-dividers R 21 (J)/ R 22 (J)
12 •S1W and R 23 (f)/R 24 (J) both transistors switch on. Due to the low on resistance
duty cycle::26°/o
of the transistors, the a.c. drive current for the complementary driver
4 tran sistors is limited to a certain level, controlled by the preset po-
tentiometers R 2 1(I) and R 24 (f). For additional safety, two fast-acting
1.6 A fu ses are inserted . The short-circuit and overdrive conditions of
Fig. 3.40. S hort- circuit couditious of the 25 wall amplifier without protectiou. the co mplementary dri ve r and output transistors is shown in Fig. 3.42,

104 105
from which the dissipation of the output transistors can be calcu lated.
0.8

Average supply current 1.42 A P11pn driver = 1.63 w le

!Al
8 0137 { n.p. nl
dr iver stage
le
!Al
80182
output stage
Regulated supply voltage 48 v p pnp drive r = 2.6 w 0.6 6
Supply power 68.16 w P11pn outp u t = 21 w
P npn output = 42 w 0.4 4
67.23 w
0. 2 f--- - - - - - ----,

A
0 0'---'--o~.L-2-'--o~.4-'--~o.~
6 -'-'-o~.8-'--1'-­ 0
0 0.2 0.4 0.6 0.8 1
tfmsl t(msl
TR81ll B P =0.2 x12x0.68=1 .63W P•2.2 x14 x0.68 • 21W
BC148
4.7 k!l.
30 30
R22 Ill
Vee v,.
IVI IVI
c 20 20

R23 1ll
10 10
4.7 k!l.
TR91ll
BCISB
R24
111 0 0'---'---'0.-2......_o~.4-'-~
o.~6~--,o~.8,.......__._1_ 00 0.2 0.4 0.6 0.8 1
t(msl tlmsl
E

(a) protection circuit 1

1ZH415

0.8
Ic B0138lp.n.pl Ic 80182
driver stage IAI output stage
IAI
6
A H 0.6

4
--- - ------·--- ---.
TR81III 0. 4
BCl4B
- - - ------ -- -- 1'----~
R211IIJ 0. 2
8.2k!l.
0 0'---'---'0.-2 ......_o~.4-'--~
o.~6_..._o~.8-'--1'­
0.2 0.4 0.6 0.8 1 tlmsl
tlmsl
P •4.4x30x 0.32 •42 W
R241III P ' 0. 27 x 30 x 0.32' 2.6W
8.2k!l.
30 - --------- - ~---~
TR9 IIII
~ v,.
30 ---- ---- -- ·---~
v,.
(VI
IVI
20
20
G
10
10
{b) protection circuit fl
9\
i>. 0 0'--'---o~.-
2 --'---,o~.4-'--~o.-6-'-'--,o~.8-'-~,'­
0 0'--'---o~.-2--'--o~
. 4-'-~o.-6_.._...o~
. 8-'-~,'-
tlmsl
tlmsl
Fig. 3.41 . Short-circuit protection networks for 25 W amplifier.
Fig. 3.42. Short-circuit conditions with protection circuit I.

106 107
Total dissipation is roughly 68 Wand a very large heat-sink is therefore Protection Circuit II
required with this method of protection. Due to the high dissipation
(a) Set R 22 (//) a nd R 23 (/l ) to mid-position.
under short-circuit conditions, R 17 and R 18 must be IOW types. Without
(b) Connect a 2 n, 20 W resistor across the output.
short-circuit protection 2 W types may be used for these resistors, and a
(c) Connect an oscilloscope across R 1 7 , a nd preferably, if it is a double-
2 mm thick bright aluminium heat-sink measuring 4.5 x 4.5 cm em-
beam instrument, across R 1 8 as well.
ployed for the output transistors.
(d) Apply a 1000 Hz signal to the input, of sufficient amplitude to
Short-circuit Protection Network II shown in Fig. 3.41(b) is similar
measure 2 V peak across R 1 7 /R 1 8 .
to that used on the 20 W amplifier described earlier. The circuit uses
(e) Adjust R 22(lf) for clipping at 1.9V across R, 7 as shown in Fig. 3.43(a).
plastic transistors BC148/BC158, requiring at the same time two BAl45
(f) Adjust R 23 (11) to give protection as indicated by the voltage measured
diodes, but no fuses. Under normal operating conditions the transistors
across R 18 , shown in Fig. 3.43(b).
are again in the off condition. When the output current increases due to
short-circuiting, the base potentials of the complementary driver tran-
sistors change very rapidly with reference to the d.c. mid-point voltage
VA- At a certain level the protection transistors switch on and the a.c.
drive currents for the driver and output transistors flow through the
protection transistors instead. The level at which the protection network
starts to operate is adjusted by pre-set potentiometers R 22 (/ I) and R 23 (II).
When the output terminals are not short-circuited, but when a com-
plex load condition exists and strong overdrive is present, the output
current becomes higher than the maximum sine-wave excursion and the
protection circuit conducts. The output and driver tran sistors are then (a) ( b)
reversed-biased due to the presence of the protection transistors and the
energy in the complex load. If no additional protection was provided
the output and driver transistors could be damaged due to the reverse
voltage breakdown and to overcome this two BA 145 diodes are connect-
ed in reverse across the output transistors.
Although Circuit II requires two diodes more than Circuit I, it has the
advantage that the same size of heat-sinks as are used in the unprotected
amplifier may be employed , that is 4.5 x 4.5 cm.
The short-circuit protection networks are adjusted as follows:

Protection Circuit I Fig. 3.43. Adjustment of protection circuit fl . fn the oscillogram at th e left both wave-
forms are superimposed.
(a) Connect a 2 Q, 50 W resistor across the output.
(b) Apply a 1000 Hz signal to the input, of sufficient amplitude to give
a 2 V peak signal across R 1 8 , ( =:: 4 A x 0.47 Q), measured with
Fig. 3.44 illustrates the printed-wiring board for the 25 W amplifier.
an oscilloscope.
(c) Adjust R 24 (/) to cause clipping to commence at 1.9 Y. The component layout is shown in Fig. 3.45. Interconnections between
(d) Repeat (b) and (c) adjusting R 2 1(/) and meas uring across R, 7 • thi s a mplifier, the power s upply unit and the Universal Pre-amplifier

109
108
 input from
pre-ampl.

60182 60182

l +J 1-1
power power
supply su ppl y

Fig. 3.44. Printed-wiring board for 25 wat t hi-ft amplifier.

Fig. 3.45. Component layout of primed-wiring board for 25 watt hi-ft amplifier employing
protection circuit II.

110 111
have been dealt with in Section 3.4. Short-circuit Protection Network l1
components are included on the printed-wiring board a nd can be clearly 1159<.78
1
seen in the photograph of the completed amplifier on page 44. I I
- - 40H z I
The frequency characteristics shown in Fig. 3.46 is within 1 dB of the - - 1kH z I
- - - 12 .SkH Z I
I
0.75
I

0.5
I
I
11

II
II
0.25

-- ..
II

10 P0 (W ) 10 '

(a) without short-circuit protection,

or with Protection Circuit If

f (Hz l 10'

Fig. 3.46. Frequency response of 25 watt /ii-fi amplifier. 0 dB = 6 dB belo w 25 watts.

d tot __ ~~=:
( O/o) 1---+-+-+-<_._..,'-+< - •- 12 . SkHz f-tttt---IH-l-+-H+f+---+1-jlt-J-+I~

response at I 000 Hz from below 20 Hz to 90000 Hz. Total harmonic

distortion as a function of output power at the three frequencies 40 Hz, o.sr----t---t+tffi-tt--t--t-+++l+lt--+--+-++H+l+----lr--S+--l-l-l-+1-1-1
1000 Hz and 12 500 Hz is shown in Fig. 3.47(a) for an amplifier without I
short-circuit protection or with Protection Circuit II, and in Fig. 3.47(b) ,
0.25 r--t----t--tt-tti+r--t--r-+~+++--+--+-!--til-++l+--i-':+-++--+-+1-+-++l
for when Protection Circuit I is used. Up to 25 W output the total har- II,
monic distortion is approximately 0.1 % with intermodu lation distortion --
at 0.6 % for ampli fiers employing either no protection network at all , or
using Protection Circuit 11. If Protection Circuit l is used , intermodulation ~o~_,;--..__'-'--'-L1-ill10~
-.-L...J'-LLLI.il.L-..l......l....Ll.J...il.l1Lo_..(_~Ll~W~)_lljW10 2

distortion rises to 1.2 %. Intermodulation distortion is measured with

the frequencies 250 Hz and 8000 Hz in the proportion of 4: 1 ( b) wirh Protection Circuit I
The power bandwidth characteristic shown in Fig. 3.48 is for a tota l
distortion of 1 %, using an 8 Q load and a source impedance of 1 kQ. Fig. 3.47. Tora/ harmonic distortion of 25 watt amplifier.

112 113
 Co nstruction of a 20 + 20 W Hj-fi Stereo Amplifier
The 20 + 20 W hi-fi stereo amplifier pictured here utilizes two Universal
Pre-amplifiers (Circuit 8), two 25 W Power Amplifiers (Circuit 24) and
an Overdrive Indicator (Circuit 26). BD 182 si licon power transistors are
employed in a quasi-complementary output circuit.

The unit is divided into three sections. On the left of the unit the inputs
are taken from the sockets at the rear, via a source se lector switch, to
the left and right channel pre-amplifiers. In the centre are the two 25 W
power amplifiers toge ther with the overdrive indicator circuit. At the right
are the power supply components.

24

22

Fig. 3.48. Power bandwidth characteristic of 25 watt amplifier. Full line - without
short-circuit protection, dotted line - with protection circuit I.

In the picture of the front panel shown here, the source selector switch
is at the far left and the mono/stereo switch is above the vo lume control.
Two 24 Y, 0.02 A lamps, centra ll y mounted , are used in the overdrive
indicator circuit. The on/off switch and the pilot lamp are on the far
ri ght.

114 115
The left-ha nd side view of the stereo am plifier give n here clea rl y shows
the positions of the two pre-amplifiers a nd also the screen which encloses
the selector switch. Wiring run s are kept as short as practicable without
the ri sk o f bum pick-up .

This picture shows the right-h an d si de of the a mplifier. T he ma in s trans- The pl a n view shown here clearly illustrates the simple and com pact
former at the rear has a seco nd ary voltage of 37 V. The smoothing layo ut. The sheet-m etal heat-sinks of the outp ut transistors are screwed
cap acitors of 2500 µF and 1600 µF can be seen, whil st the two smaller to the ma in chassis members and a mple heat dissipation .is obtained.
ele trolytics are the o utpu t ca pacitors. Fo ur BY 126 bridge-connected T he overdri ve circuit co mpo nents are mounted o n the sma ll printed
diodes provide the suppl y vo ltage. wiring board , centrally located, behind the front panel.

11 6 117
 3.6.5 CIRCUIT 25 - 40 WATT H1-F1 AMPLIFIER USING 80182 OUTPUT
TRANSISTORS

Performance specification:
nominal power output 40 w into 8 n load
sensitivity (1 000 Hz for P 0 = 40 W) 440 mV
input impedance 150 kn
frequency response (- 0.5 ct B) 12 to 95000 Hz
total harmo nic distort ion at P 0 = 40 W 0.2 %
intermodulation distortion at P 0 = 40 W 0.8 %
(measured with f 1 = 250 Hz and f 2 =
8 kHz, where Vn :V12 = 4 : 1)
unweighted signal/ noi se ratio (ref. P 0 = 50 mW) 78 dB
internal resi stance at output socket o.o5 n
damping factor with 8 n speake r 160
voltage feedback factor 280
nominal supply voltage 60 v
current consumption at P0 = 40 W I. I A

The 40 W amplifier described here is very similar in design to the 25 W

amplifier discussed in the last Sub-section. Complying in every respect
with the European Standard for high fidelity sound reproduction, it

driver output

pre-amplifier pre-driver

input,... ~ an
ts~ k~~
0 40W

Fig. 3.49. Block diagram of the 40 watt hi-Ji amplifier.

will deliver 40 w r.m.s. continuous sine-wave power into an 8 n load.

Si licon transistors have been used throughout the circuit and the output
stage comprises two rugged BD 182 power transistors in the usual
quasi-complementary single-ended C lass-B push-pull configuration.
Close-up of the left-channel power amplifier. Protection Circuit IT has The complementary driver transistors, however, are now the BD 139/
beeh used and the components can be easily indentified. The 0.47 Q BDl40 types because of the higher supply voltage (60 V). An n-p-n
emitter resistors are fitted between the BO 182 output transistors . BO 139 is used for the Class-A pre-driver stage, whil st a p-n-p BC 157

118 119
 R5 H
which TR 3 has to deliver is 209 mA; TR 4 passes a somewhat lower
+ C2 current. TR 3 requires a heat-sink but it is recommended that the driver

r 25µF Rll F R12 A

164VJ
lk!l tra nsistors are each mounted on separate .I mm thick bright aluminium
470!1
heat-sinks of 2 x 2 cm 2 •
The peak output current for P 0 = 40 W into a n 8 Q load is 3.2 A.
The peak output voltage for 40 W is 25.5 V. Upper circuit losses are
abo ut 4.0 V, and lower circuit losses total 5.0 V. To guarantee 40 W
output power with minimum de vices the supply voltage must equal
the sum of the total voltage losses and twice the peak output voltage
excursion. This becomes:
4.0 V + 5.0 V + 2(25.5 V) = 60 Y.
sn The mid-point voltage VA is therefore 30 V.
To compensate the quiescent currents of the complementary driver
and output transistors against ambient temperature a nd supply voltage
variations, the circuit previously described for the 25 W amplifier has
1.Bkfl
been used. TR 7 is an n-p-n transistor BCJ 48. Each output transistor re-
quires a 2 mm thick bright aluminium heat-sink measuring at least
Fig. 3.50. Circuit diagram of 40 the watt hi-Ji amplifier. 8 x 8cm 2 •
Short-circuit protection networks are required for this amplifier,
as with the 25 W amplifier, for the same reasons. Jn Figs. 3.5 1 and 3.52
two protection networks are given. That shown in Fig. 3.51 consists of
functions as the pre-amplifier and mid-point voltage stabilizer. The block two parts; a regulated power supply unit and the other an additional
diagram given in Fig. 3.49 shows the general arrangement. transistor between the base of the driver transistor TR 4 and the mid-
The circuit diagram is given in Fig. 3.50. In view of the similarities point of the amplifier. The circuit of Fig. 3.52 is similar to that of the
between this amplifier and the 25 W amplifier no detailed description of 20 W amplifier and also Protection Circuit 11 of the 25 W amplifier.
its operation is necessary , but a few of the circuit parameters may be of Where an unregulated power supply is used BD 183 output transistors
particular interest. The operating current of the pre-amplifier stage in the should be used instead of the BD 182.
final amplifier is 0.4 mA and feedback is ap p lied around the circuit as Firstly, consider Fig. 3.5l(a). The regulated supply unit delivers a con-
before, a voltage feedback factor of 280 being used. The input impedance stant d.c. voltage up to a certain level of output current. When the current
is again 150 H2. consumption exceeds this level , for example - due to a short-circuit
The Class-A BD 139 pre-driver stage operates with a collector current in the speaker leads, the d.c. output voltage of the supply unit drops to
of 7.7 mA and no heat-sink is required. a lmost zero. As soon as the short-circuit is removed the stabilized d.c.
Complementary driver transistors TR 3 and TR 4 aga in act as a phase voltage commences to return to its previously set value, falling again
inverter. Typical quiescent current is about JO rnA . •In the worst case, to zero when the current drawn becomes too high. This repetitive beha-
represented by a 10 % higher supp ly voltage, minimum gain transistors viour is a function of the time-constant of the supply unit and the magni-
in the output stage, a 20 % lower external lo ad imped a nce and driven tude of the current consumption .
by a signal hav ing an amplitude of 2/n times the peak sine-wave input, With this form of protection the current through the upper part of the
the dissipation of the driver tran sistors is 840 mW. Maximum current output circuit (TR 5 ) and the driver transi stor TR 3 is limited. But not so

120 121
 for the lower stage of the amplifier due to the output coupling capacitor
C 1 0 acting as a "s upply unit ''. Additional protection is therefore neces-
sary for the lower part of the amplifier. This is obtained by connecting
76Vdc at no signa l _ R32
22k n
a p-n-p transi stor BC 157 between the base of the BO 140 (TR 4 ) and the
6SV!k at 2x40W
mid-point of the amplifier. The circuit is given in Fig. 3.51(b).
Under norm al conditions the transistor is non-conducting. When the
output current increases due to a short-circuit, the base potential of
110V Cl1 the p-n-p driver transistor TR 4 changes rapidly with respect to the d.c.
+ mono :
1600 µF
stereo :
stereo : 1500 µF
8x8Y126 164V)
*) mono SOOD µF (i..0 V) A H
ster eo 8000 µF (4.0 V)

TR9
BCl47

lfi R14
IOkfi

(a) regulated power supply unit

R11
IOkfi
TR 8
BCl57
H

Fig. 3.52. Protection Circuit II.

R11
10 kfi

mid-point voltage VA- At a certain level the protection transistor TR 8

(F ig. 3.51(b)) conducts and the a.c. drive current flows through TR 8
instead.
When the output is not short-circuited , but when complex load and
(b) part of Protection Circuit I overdrive conditions apply, the output current also becomes higher than
the maximum sine-wave excursion. The upper output stage is limited
by the supply unit, but the protection transi stor conducts and the output
and complementary driver tran sistors become reverse-biased . To avoid
damage to the upper transistors, a BA 148 diode is reverse connected
Fig. 3.51. Short-circuit protection network I. across them.
The supply unit of Fig. 3.51 (a) can also be used for stereo applications.
In such cases the current limiting value must be set twice as high. The
bridge rectifier uses 4 x BY126 rectifier diodes for mono ; on stereo
8 x BY126 diodes are required.

122 123
 24 A BO 182 t ransistor (TR 10 ) is used as a current regul a to r. Thi s acts as
Ic
IA! 20 th e current so urce with the load in the collector circui t. Th e c ur rent
so urce is co nt rolled by a 80140 p-n-p tra nsisto r TR 11 , whi ch d eter-
16
mines th e ma ximum a va il a ble base dri ve current for the BD 182. Th e
12 60182
upper stage suppl y current limitin g va lu e is set by th e potenti o meter R 27 • The d .c.
o utp ut voltage is co ntro ll ed by a 12 Y ze ner di ode BZY 88-C l2 a nd a
BC 147 tra nsisto r, TR 1 2 • The vo ltage di vider netwo r k R 33 -R 34 -R 35 is
used to o bta in exactl y 60 Y d .c. s uppl y voltage, by adju stme nt o f R 33 .
0. 2 0. 4 0.6 0.8
tfms)
1 F ig. 3.52 shows Protectio n C ircuit JI. It is identi cal with th at used
in the 20 W a mplifi er a nd a lso as a n a lte rn a ti ve in th e 25 W a mplifier.
24 ~~~~~~~~~
VcE W hil st Fi g. 3.53 shows the sho rt-circuit co nditi o ns in th e 40 W a mplifi er
IVI
20 o utput stage witho ut pro tecti o n, reference to Fi g. 3.54 shows wha t ha p-
16 peak dissi pation pens under sho rt-circuit conditi o ns when Protecti o n C irc uit II is used .
= 168W
average diss ipation
12
=124W
du ty cy c le:: 74°1 0

0
0~---'---'
0.-2 -'--o~.4--'~oL
. 6--'-.L.J
o.-
8 -'---'-1-
t( msl

24
Ic
IA! 20 time

16
-1
12 6 0182
lower stage

-2
4

0 0~...._~
o .~2-'--o~.4--'~oL
. 6--'-.J....JQ~
8 _._--11_
t(msl

24
VCE
IV! 20

peak dissi pa t ion

16
=3 85W
average dissipat ion
12
o!O OW
duty cycle : 26°/ 0
I

0 o~...._7
0 . ~2_.__0~4--'~0L
. 6---'-.1.....1
0.- 8 -'---11- circui t no t wo r k in g circui t wo r k i ng
t(ms)

Fig. 3.53. S hort-circuit couditio11s of the 40 W amplifier without protectiou. Fig. 3.54 ( a). S hort-circuit conditions fo r Protection Circuit II with 2 D load.

124 125
··'(.
-
The output currents flowing through R 1 7 and R 1 8 (Fig. 3.50), are
shown in Fig. 3.54, before and after the protection circuit is functioning.
On the left of Fig. 3.54(a) the voltage drop across R 1 7 and R 18 is shown ,
short-circuit
without the protection circuit working, when a load impedance of 2 n
cond it ion
Rt-::Ofi
is co nnected . Peak current under these conditions is 4.1 A. At an in-
creased current, protection commences and the voltage across R 1 7 and
t ime
R 1 8 falls as shown on the right of Fig. 3.54( a). The level at which pro-
tection starts working is adjusted by potentiometers R 22 and R 23 in
-1
Fig. 3.42. The 2 r2 load condition is shown here because this value mu st
be used to adjust the protection circuit.
-2
VR1e Fig. 3.54(b) shows what happens under actual conditions when an
(VI
8 n load impedance is connected. The left hand side of Fig. 3.54( b)
shows the voltage drop across R 17 and R 18 for 40 W output and at clip-
ping. The right hand side of Fig. 3.54(b) shows the effect of a short-circuit
for the same a.c. input signal as for clipping. The average currents through
the output transistors are now much reduced .
The short-circuit protection networks are adjusted as follows:
Protection Circuit I
(a) Disconnect the amplifier from the regulated supply unit.
(b) Connect a d.c. voltmeter across capacitors C 13 -C 14 (100 V range).
(c) Adjust R 30 for maximum resistance value.
(d) Adjust R 27 to the TR 11 end of its track.
at 40 W output (e) Switch on mains supply.
(f) 1f a d.c. voltage appears across C 13 -C 14 , connect d.c. voltmeter
across C 12 .
(g) If there is no voltage across C 12 , turn R 3 0 slowly until there is about
60 V. Do NOT return R 30 , after this step.
(h) Adjust R 30 for exactly 60 V across C 12 •
short-circuit conditions
(i) Switch off the power supply and connect the amplifier. Connect a
2 r2 resi stor (40 W) across the output, or two 2 r2 resistors for a stereo
installation.
(j) Connect an oscilloscope across R 1 8 •
(k) Inject a IOOO Hz signal at the amplifier input of sufficient amplitude
at clipping to produce 2.5 V pea k across R 18 (approximately 4.1 A peak current).
The result is shown in Fig. 3.55(a).
(I) Increase the input signal voltage further. The oscilloscope should
now show a trace similar to Fig. 3.55(b) indicating that protection
Fig. 3.54 { b) Co11di1io11s in Pro1 ec1iv11 Circuit II. is taking place. Both ste reo channels should indicate the same way.

126 127
 (a)

id ea l case

(b) tim~

(a)
(a) (b)

oseillograms
obtained in
practice

(b)

Fig. 3.56. Adjustment of Shorr-circuit Protection Network II.

Fig. 3.55. Adj11stme111 of short-circuit Protectio11 N et work I.

Protection Circuit 11
(a) Set potentiometers R 22 and R 23 to their mid-positions.
(b) Connect a 2 Q resistor (40 W) across the amplifier output.
(c) Connect an oscilloscope across resistor R 1 7 • If thi s is a double-beam
(m) Reduce input to obtain the trace of Fig. 3.55(a). instrument, its other input can be connected across R 18 •
(n) Adjust with R 27 the maximum current level just before the point at (d ) Inject a IOOO Hz input signal of sufficient a mplitude to produce a
which the protection of the regulated power supply starts to work , i.e. 2.4 V peak signal across R 1 7 .
before the trace of Fig. 3.55( a) falls away to .become Fig. 3.55( b). (e) Adjust R 23 for clipping of the waveform a t 2.4 V as shown in Fig.
(o) Check the supply unit again by shunting C 12 with a 2 Q resisto r, ' 3.56( a).
the d.c. voltmeter sti ll connected across C 1 2 • If the supply voltage (f) AdjustR 22 fora trace similar to that shown in Fig. 3.56(b) of the volt-
1
does not re-appear after removing the 2 Q resistor, a very slight re- age across R 18 • This indica tes that the protection circuit has just
started working.
adjustment of R 3 0 is necessary.

129
128
r
+1

(dBi
The frequency characteristic shown in Fig. 3.57 is within 0.5 dB of
,, the 1000 Hz response from below 15 H z to 95000 Hz. Total harmo ni c
/

distortion as a function of output power a t the three frequencies 40 Hz,

-1
1000 Hz a nd 12500 Hz is shown in Fig. 3.58. These c urves are va lid
for amplifiers with or without protection network s incorpo rated , and
- 2
the harmonic distortion does not exceed 0.2 %at 40 W. Intermodu lation
di stortion is 0.8 %at maximum power output , measured with frequencie s
- 3
of 250 H z and 8000 Hz in the ratio of 4: I.

-4

-5 45
10 10 ' 10' f !Hz)
P,
IWI
40
Fig. 3.57. Frequency response characteristic of 40 wa tt hi-Ji amplifier. 0 dB = 6 dB
below 40 W.
35 ,

30

25
?n9490

I I I I
- - 40Hz
20
--1kH z
- - - 12.SkHz I
I
0.7 5 15
10 10 2 10 3 f(Hzl
I
I
I
I
Fig. 3.59. Power bandwidth characteristic of40 wa tt amplifier.
0.5
i
I
I
I
0.25
I T he power bandwidth characteristic is given in Fig. 3.59, for a tota l
I
- -.-,.__ • .-t di stortion of I %, using a n 8 Q load and a source impedance of I kQ .
Fig. 3.60 illu strates the printed-wiring board for the 40 W amplifier.
10 P, IWJ The co mponent layout is shown in Fig. 3.61. Interconnections between
thi s a mplifier, the power supply unit and the Universal Pre-amplifier are
Fig. 3.58. Tota l harmonic distortion of 40 wa it hi-Ji amplifier. most important and the recommendations of Section 3.4 must be closely
fo ll owed otherwi se instabi li ty is most likely to occ ur. The components
for Protection Circuit ll can be clearly identified in the photograph of
the completed amplifier.

130
13 1
r
3.6.6 CIRCUIT 26 - OVERDRIVE INDICATOR FOR 25 W AND 40 W
AMPLIFI ERS
Fig. 3.62 shows the circuit of a simple overdrive indicator which may
be used with the 25 W and 40 W high fidelity power amplifiers. It is
intended for use in stereo applications and causes a lamp to light at the
onset of clipping, indicating that audible distortion is about to occur.
The inputs to the circuit are taken from the output terminals of each

3300 330fi

R channel
L channe l input
Fig. 3.60. Printed- wiring board for 40 watt amplifier. input

r----===----------::--:----;c::a:=:+--

·+;; ~ ·"-· ·
Input from

~ W' or~11 . ·~
Fig. 3.62. Circuit of overdrive indicator for 25 Wand 40 W amplifiers.

~
'li
680

{'l
pf 1~kn~~~ QA~ ~
l ~ eci 57 Y21 F 80139
~BC148 ~ ~V ..am& br:ilJ ~ channel amplifier. The audio signals are rectified by the BAX13 diodes
~ ~
] - 1-1
3JOpF
,.._.-1--~ power a nd a d.c. voltage is developed across each potentiometer. The BCI47
~BA'48 ~~ 0f~1!~lc-l ~~
supply

~
transistors are normally held cut off by the BZY88-C5V6 zener diode,

l+I
power
~
00139 ..j
f~£1:.~' ffl :!: ~ ®~~ ~ '...'ec1s1
..j s6n f.. ··._u· : ; ! !~i
b~
41on r
'-J
:·: .. ,..
L!j
I .:t
b~
eo140
but when the voltage on the base of the BCJ47's exceeds 6 V the transis-
tors are switched on and the lamps light.
For setting-up, an oscilloscope is connected across each pair of input
supply 4 .7kn 4.7kn terminals and with a music input signal the potentiometers are adjusted
for the lamps to light at the onset of clipping. The values of R 1 and R 2 ,
a nd also the supply voltage, shown in brackets, apply when the circuit is
used on a 40 +
40 W stereo equipment. No other components are af-
fected .
lSOOµF

Fig. 3.6 1. Con1ponent layout u/ printed-wiring board for 40 watt amplifier employing
Protection Circuit I. Dotted components are added if Protection Circuit fl is used.

132 133
 4. Integrated Circuit Amplifiers
4.1 Application of Integrated Circuits to Audio Amplifiers
Integrated circuits in the audio field offer a number of special advan-
tages. Not only where small size is the main consideration but also on
grounds of cost, the use of an integrated circuit is a particularly at-
tractive alternative to discrete components.
The main advantage of employing integrated circuits lies in the use of
less di screte components which require individual storage, handling
a nd mounting. Also the overall circuit reliability will be higher, since
there are less soldered connections to be made in the final product.
The reduced dimensions of an integrated circuit make it of special
a ppeal for microphone applications. With capacitor microphones, for
exa mple , the pre-a mplifier may be mounted in the body of the microphone
close to the sensitive element. Another clear example is that of a
hearing aid amplifier which, by employing an integrated circuit, can be
completely inserted in the ear.
To illustrate the a pplication of integra ted circuits in the audio field ,
four different types have been selected and practical circuits are given
showing examples of their use. The TAA300 has a 1 W Class-B output
stage and needs only 10 m V drive for full output ; the T AA 320 is specially
designed for high impedance applications such as crystal pick-ups ; the
T AA435 is a pre- amplifier and driver stage intended primarily for car
radios and the T AA3 l 0 is a low-noise pre-amplifier for tape recorders.
The T AA300 has been described in greater detail than the other integra-
ted circuits for the benefit of interested readers.

The 40 W high fidelity po wer amplifier.

Components for Protection Circuit II ha ve
been incorporated in this model.

135
134
4.2 Practical Circuits RB

4. 2. 1 CIRCUIT 27 - ON E WATT TRANSFORM ERLESS CLASS-B AUDIO

AMPLIFIER FOR R ECO RD PLAYERS USING THE TAA300 INTEG RATED
CIRCUIT
Performance specification, each channel:
nominal power o utput I W into 8 0 load
sensitivi ty ( I 000 H z) for P 0 = I W
pick-up (I 000 pF) 1.5 v
at TAA300 input 8.5 mV (typ.)
input impedance of TAA300 > 10 kO (typ. 15 kO)
frequency response (- 3 dB) 120 to 26000 Hz
total harmonic distortion at P 0 = I W 10 %
at onset of clipping P 0 ""' 0.8 W ""' I %
unweighted signal/noise ratio (ref. P 0 = I W) 73 dB
voltage feedback factor IO
nomin a l supply voltage 9V Fig. 4.2. Circuir of the TA A JOO .
cu rrent consumption at P 0 = I W 180 mA (typ .)

In many portable audio applications the required output power does not To meet these requirements the TAA300 has been specially developed.
exceed I W. The sensitivity and the input impedance in this case should It contains a silicon chip of only 2 mm 2 , on which 9 planar n-p-n tran-
sisto rs, 2 p-n-p transistors, 5 diodes, 14 resistors and I capacitor are
integrated. The TAA300 only requires a few external components as
show n in the basic ci rcuit of Fig. 4.1 which will be used for purposes of
discussion.
The circuit diagram of the TAA300 is shown in Fig. 4.2. The TAA300
co nsists of a n input stage (TR 1 a nd TR 2 ) , a driver stage (TR 3 to TR 5 ) and
a n output stage (TR 6 to TR 1 1)·
The input stage is a differential a mplifier. Since resistor Rs in the com-
mon emitters of the differential stage is very large with respect to the
differential resistance of the transi stors, it can be regarded as a constant
current so urce.
The voltage at the base of TR 1 is obtained by means of a low resistance
voltage divider across the supply. Resistor R 3 between the voltage divider
Fig. 4. 1. Basic circuit arrangement of one warr amplifier. a nd the base of TR 1 increases the input impedance to about 15 kQ. Two
diodes a re connected in series with R 1 so that the voltage at the base 01
be such that the maximum output power is obtained when the input is TR, does not vary directly with the supply voltage. This allows the circuit
connected to a ceramic pick-up element or a detector in a portable radio. to be used at supply voltages of 4.5 to 10 V. The point below the two
Other requirements a re that the circuit shou ld sti ll operate at half the diodes in the bias network of the first transistor is connected to pin 6,
nominal supply voltage and that the total current drain is low . Further- thus enabling the supply line to be decoupled by means of an external
more the complete a udio circuit should be compact, easy to mount, have capacitor. The layout of resistors R 1 , R 2 and R 8 is such that the spread of
a low noise a nd a reaso nably low di stortion factor. the base voltage of TR 1 remains below 5 %.

136 137
 The driver stage consists of 2 n-p-n transistors TR 4 a nd TR 5 in cas-
cade. T he d.c. co upling between the differential input stage and the driver v.
stage is by means of a p-n-p transistor (TR 3 ). This transistor has the func- IV I

t ion of the level shifter, but it also presents a sym metrica l load to the dif-
ferentia l input stage. Conseq uent ly the noise factor of the input stage is max
I/,,. nom
no worse than that of a single transistor, whereas the advantages of a / min -~-

differential input stage are mainta ined. ..... ..........

4
/I//
The voltage gain of the driver stage is rather high and therefore this .,,.
stage is used to cut off the freque ncies above the audio spectru m by ~,::::;
"'
~

means of a small integrated capacitor C 1 inserted between the base of .,.

;:%"'
v
TR4 and its collector. The voltage gain of the cascade driver stage wi ll "'I"'
~

be about 200 when the quiescent current of the stage is 0.47 mA. This
gain figure, in combination with a co llector/ base capacitance C 1 of IO pF,
4 10 Vs IV) 12
makes the capcitance between the base of TR 4 a nd earth about 200 x IO
= 2000 pF. Fig. 4.3. Variat ion of mid-point voltage VA with supply voltage V 8 .
Without feedback the upper freq uen cy is lim ited to approx im ate ly
3 kHz (- 3 dB). The total freque ncy response can be expanded to about
30 kHz (- 3 dB) by using large overall feedback. To reduce the spread replaced by transistor TR 8 and resistors R 12 and R 13 • Since the integra-
of the input impedance of this stage, and thus the spread in cut-off ted components may spread , it shou ld be possible to adjustthe quiescent
frequency, a resistor of 30 H2 is shunted across the input of the driver current of the outp ut transistors, for which purpose the base and emitter
stage. D4 and Ds a re connected in series with the resistor to decrease of TR 8 have external connections.
the current of the p-n-p level shift transistor (TR 3 ) and to stabi li ze the For a supply voltage of 9 V, the total quiescent curren t consumption
coll ector current of TR 4 and TR 5 again st temperature vari a tions. must be preset at 8 mA. The typical current consumption of all stages
In the TAA300, the convent ional complementary pair principle is (except for the output stage) is then about 3.5 mA as shown:
used. T hi s gives a very simple and stable direct-coupled single-end ed - bias netwo rk of the input stage 500 µ A
push-pull output stage. - tota l curre nt of differential in put stage 380 µA
To reduce the coll ector d.c. cu rrent of the driver transistor the output - drive r stage 470 µ A
transistors must have large cur rent gain s, therefore two transistors in - typ ica l co ll ector current of stab ilization
transistor TR 8 2 100 µA
cascade are used. The output stage is made complementary by placing
- co ll ector curren t of TR 7 and TR 9 100 µA
p-n-p transistor TR 6 in front of the lower cascade. Symmetrical drive
3550 µA
of the output transistors is ensured by making the c urrent gai n of the
p-n-p transistor unity . The typical quiescent current of the output stage is therefore 8- 3.5
The output voltage is limited only by the knee voltage of o utput 4.5 mA which is necessary to minimize the cross-over distortion .
transistors TR 10 and TR 11 and , for symmetrical cl ipping of the output sig- Fig. 4.4 shows the spread of the total quiescent current consumption
nal , a mid-point vo ltage of half the supply vo ltage is required. Fig. 4.3 of the T AA300 versus supply voltage when preset at 8 mA for a supply
shows the mid-point vo ltage as a function of the supp ly voltage. vo ltage of 9 V. The dotted li ne shows the typica l quiescent c urrent con -
The quiescent current of the outp ut stage cou ld be stabilized against sumption without the quiescent c urrent of the output stage.
the 1nfl uence of battery a nd temperature variations of the V 11E ' s of The maximum dissipation for sine-wave drive of a Class-B amplifier
TR9 , TR10 and TR6 by using three d iodes. In the TAA300 these diodes are is theoretically obtained when the peak collector current of the o utput

138 139
 20

Io to t
Im AI

..... - max
/ I ..._
0
v
.....

I/
I/
v
/

,,,. v
nom

l
I

m. i~i-
~-
- ,_

I/ v
.....
-....
5 ,
I,/ I/ ..... ,_ - ,_ -
I J"' , .L... "- j.
!!:~ ,_,_·-
0
4 Va IVI

Fig. 4.4. Spread in total quiescent current / 0 tot with supply voltage V8 •

transistors is 2 I cM/n. For a 9 V circuit with a load impedance of 8 Q the

theoretical maximum peak current is:
VB
l cM = - - = 560 mA.
2 RL

At a current of 2 l cM/n the power from the supply is:

VB x 2/nlcM 9 x 360 x J0- 3
1020 mW.
n n
The output power fed into the load impedance is:
2

( v2
2 I cM/n)
· RL = 520mW.

Consequently the dissipation in the output tra nsistors amounts to

( 1020- 520) mW = 500 mW. Together with the current of about 10 mA
Th e TAA300 integrated circuit enlarged 80 times. It combines 1 l transistors, 5 diodes, 14
resistors and I capacitor on a single chip of 2 mm 2 •
for a ll pre-stages, the total dissipation wi ll be increased by approximately
90 mW. Thus the total dissipation becomes approx im ately 600 mW
at a supply voltage of 9 V.
For sine-wave drive the current drain is 180 mA at P 0 = I W, so at a
supply voltage of 9 V the total dissipation of the T AA300 will be (9 X
0.1 8)- 1 = 0.62 W. Fig. 4.5 shows the dissipation as a function of output
power.

140 141
The maximum dissipation for music and speech is always lower than The thermal resistance of the TAA300 without heat-sink is 225 °C/W.
for sine-wave drive. Therefore, when this amplifier is used as a normal Due to the good thermal coupling between the output transistors and
a udi o amplifier it is permissible to assume a maximum dissipation of the stabilizing components, use can be made of the high junction temper-
500 mW at a battery voltage of 9 V. ature permitted in sil icon transistors. With a maximum dissipation of
500 mW and the maximum crystal temperat ure of 150 °C, the 1 W ampli-
BOO
fi er can be used without heat-sink up to an ambient temperature of
/
~-i-..
.., 150- (0.5 x 225) = 37.5 °C .
(mW I
/
T a king into acco unt a maximum voltage of 10 V and continuo ns
6 00
/ _..... sin e-wave drive, the maximum dissipation is 750 mW . For this condition
Va ~ 10V
_,,.,-- -- 1,,,.,--
/ ,,/
9V/

400
v _,,.,, 10'
v
/ V" V;
/
""
/ Im VI p~ ·~ 1W
/ /
,,/ 10' O. SW
200

/V
'/
0
10 10' P0 !mW) 10'

Fig. 4.5. Total dissipation of th e TAA300 f or different output po wers. v,,

1000

1
(m WI

..,.., I" ' \ 10 10'

""'"""' "-.
\
~ Fig. 4. 7. Input voltage required as a function of th e value of f eedba ck resistor f or power
500
~,.._

!'-.... r-!.....
~ ..,..,.., ""'I" ' ~ ~
\
1\4
outputs of0.5 Wand 1 W.

--....,..,..,
" '~\ \
r-..........
r-..........
..........
!'-....
I~ the thermal resistance can be improved by means of a cooling clip.
Fig. 4.6 gives the derating factor of the TAA300.
!'-.... i-....--...., ~\ To reduce the distortion and to improve the frequency response and
I'-..:: ~ the input impedance, voltage feedback is applied. This is obtained by
so 100 150 means of integrated resistor R 14 , and the external components Rf and
Fig. 4.6. Max imum permiss ible dissipation over a range of ambient temperatures for C3 shown in the circuit diagram of Fig. 4.1. With a feedback resistor of
various cooling systems: Rf = 47 Q the negative feedback is about 20 dB. This, together with the
Curve 1: without cooling; high open loop gain, gives a sensitivity of about 8.5 mY for an r.m .s.
<:;urve 2: with cooling clip No. 56265;
Cur ve 3: with cooling clip No. 56265 and a heat-sink of 20 cm 2 ;
output voltage of 2.8 V. With a load impedance of 8 Q the output
Curve 4: with an "infinite" heat-sink. power will then be 1 W.

142 143
 Id Bl l-l-+-+--+-+-i-l-j--J--!---1--1-l-l--J-+--l--L..l- '----l- _ 1-l-+-+--+-+-i-l-j
d tot
(O/o) J----+-J--t-t-+t+f+---t- +--+-+-+-+l+f---+--+-+-+-++++-1

-10 r--t-+1/:tt--t-t-r--t-+-t--t-t-lr-+-+-t--t-t-lf-+-+-t-+-t-1f-+-+-+-+-~

'
-2o f---1-t-+-t--t-t-1-t-+-t--t-t-1-t-+-t--t-t-1-+-+-+-+-l-l-+-+-+-~

- 3 0,.
.L.J--'--'--'--'-L..l-L-+--,L-.L..L..l--1.-1......L.L..L..l--1.-1......L1-L
Vs~l-
Vl.J_L..J_J__J
10

F g . 4.8. R elati ve voltage gain f or variation in supply voltage. 0 dB corresponds to 0 .5 W. Fig. 4.10. Total harmonic distortion as a fun ction of output power.
Cur ve 1: with f eedba ck; Cur ve 2: without.

1254341 .1
7B l.o l46.2

P,
IWI
ldB I
~

......
v
' - / r- ~ r-...
"' .......

v
/
II
f\
"
rl. 1
\ "'"""-"" , 1 t'-...
11 .........
-5
\
~
1.

I
I 2

~
"" ...
[',."
- 10
I
"" "" I'-
f\ "

10 '
10
"\ , 100
-1 5
10 10' 10' 104 flHz l

Fig. 4. 9. Frequency response characteristic with and without the 47 n f eedback resistor Fig. 4. 11. Output po wer as a fun ction of loudspeaker impedance. Curve 1: at dtot = l O'X,.
Rf in Fig. 4. 1. 0 dB corresponds to 0.5 W . Cur ve 1: with f eedback; Cur ve 2: without. Cur ve 2: j ust below clipping.

144 145
 Fig. 4.7 gives the typical sensitivity for P 0 = I W, and P 0 = 0.5 Was a The preferred output power can be obtained ,by choosing the appropri-
function o~ the feedback resis~a nce R f. Due to the rather high feedback, ate spea ker load. However, a speaker load below 8 .Q is not permissible
the spread 1s very small. The high feedback also m akes the gain independ- because the peak collector current of the output tran sistor would be too
ent of the supply voltage as shown in Fig. 4.8. high. Fig. 4.11 gives the output power as a function of loudspeaker
Fig. 4.9 shows the frequency response without feedback and also impedance. Curve l is measured at d10 , = 10 %and Curve lI is measured
just before clipping.
Fig. 4.12 shows the practical circuit for use with a ceramic pick-up.
The se nsitivity for a source capacity of I 000 pf is 1.5 V for I W output
a nd the half-power frequency response is 80 Hz to 26000 Hz. The 0.05 µF
capacitor must be directly connected across pins I a nd 2 and a ceramic
type should be used . Similarly, the 470 pF capacitor should be directly
co nnected across pins 7 and 10. Equalization according to the R.l.A.A.
characteristic is provided.
When using a practical low-cost power supply with a n -filter (5 .Q
decoupling resistor) and an overall impeda nce of 16 .Q, the rms output
is 600 mW for 10 %distortion ( < I % up to 500 mW). Under these con-
ditions no heat-sink would be necessary for an ambient temperature up
to 50' C.
A printed-wiring board for stereo models is shown in Fig. 4.13. Sep-
Fig. 4.12. Circuit diagram of one wa tt amplifier for use with a ceramic pick-up. arate volume controls for each channel are intended to be used with this
particular layout.

w.ith a 47 .Q feedback resistor Rf. The cut-off frequency (- 3 dB) at the

~1gh frequency side is determined mainly by the integrated JO pF capac-
itor between the collector and the base of the driver transistor TR
together with the applied feedback. The - 3 dB point for the lower fr;~
quencies is determined by the external capacitors. In Fig. 4.1 O total
harmonic distortion is given as a function of the output power, measured
in the circuit of Fig. 4. l. The distortion level at an output power of 0.5 W
is less than I %. Cross-over distortion is very small and , due to the stabili-
zation of the quiescent current, it remains small even when the supply
voltage shifts from 5 to JOY.
Typical noise input voltage is about 2 µ V, measured over a bandwidth
from 30 Hz to 15 kHz with a source impedance of 2 k.Q. Related to an
input signal of 8.5 mY, this means that the signal-to-noise ratio is 73 dB.
Wi~h the input short-circuited, the r.m.s. noise output voltage is 280 µV,
wh1eh represents a noise power 80 dB below the maximum output
power of I W.

146 147
 . d-wiringlifter.
Fi . 4. 13. Printe board/or stereo
g . 01.r one wall amp
verswn

de/ of the
Stereo mom lifter. The .
one wa tt a 'P d circulls
00 integrate
T AA3 I
can be clear Y seen in
the photograph.

148
 4.2 .2 CIRCUIT 28 - Two WATT P1 cK-UP AMPLIFIER USING THE TAA320 enhancement type, a n n p-n silicon transistor a nd a resistor. Its circuit is
AND THE HIGH VOLTAGE CLASS-A TRANSISTOR BDll5 given in Fig. 4. I 5.
Performance specification Rl
, ;_ _r-t----00
nomin a l power outpu t 2 w into 4 0 load lkll
se nsi tivity ( I 000 Hz) for P 0 = 2 W 140 mV
(fo r minimum devices)
frequency respon se (- 3 dB) 40 to 12000 Hz
tota l harmoni c distortion at P 0 = 2 W 3.5 %
vo ltage feedback facto r 6.3 (typ.)
nominal supply voltage IOO V
c urrent consumption 60 mA (tota l) Fig. 4.15. Circuit of the TAA320.

By using a TAA320 integrated circuit as pre-a mplifier in combination

with a BD 115 high-vo ltage silicon power transistor, it is possible to
The biasing resistor R adjusts the operating current l s of the MOS
transistor. Since its transconductance is proportional to Vls , resistor R
must be as small as possi ble to obtain maximum gain; on the other hand
a low value reduces amplification of the silicon transistor. A value of
a bout J kQ gives optimum overall transconductance of the whole inte-
grated circuit.
The TAA320 has advantages in that it combines a high transconduc-
tance (g 1 e) with a high input impedance and a low noise level.
R2 Rl G
The BD I 15 high voltage silicon transistor is intended for Class-A
1
MO
log
output stages in audio amplifiers delivering an output of 2 W at a supply
volume
TR1
voltage of 100 V. The VcER of a lower-limit transistor BD115 is 245 V.
l!Q!l? The TAA320 requires an operating current of JO mA to ensu1e a
C4
4.7nf 1.2nF RS R7 s ufficiently high transconductance and low output conductance. Since
200~~
3900 3.30
•: :--i Cl R9
560
a high supply voltage is available it is possible to m a ke the. TAA3~0
4.7nF (6.4VI
CZ t--<- - < - - - - + - - - - < -- -..._- _ _ _,._ __.__ __.__ __, self-biasing. Despite the spread in VGs, the operating current will remain
fairly constant.
Fig. 4 .14. Circuit diagram of t wo watt pick-up amplifier. The maximum collector-to-emitter swing of the output stage is given by :

design a simple 2 W ma ins-fed Class-A amplifier which has an input Vee = V8 - VcEK - Ic(R E+ Rrn).
impedance suitable for a crystal pick-up and a very high signal-to-noise in which:
ratio. If, by accident, the pick-up is dropped on the record a high peak VB supply voltage ( 100 V)
voltage is generated. However, the T AA320 has been designed to with- VCEK knee vo ltage of the BD 115 at max imum peak current (6.5 V)
le d.c. operating c urrent of th e BD 115 (a pprox. 50 mA)
stand up to JOO V peak input without damage. In record players the mains
RE minimum required e mitter resistance for thermal stabi lity (fro m measur-
tra nsformer can even be dispe nsed with if a tapping on the motor is used. ements it follows: RE = 56 0)
The circuit diagram is shown in Fig. 4.14. d.c. resistance of the output transform er primary (p ractica l value approx.
The T AA320 is monolithic integrated circuit comprising three compo- 1400).
nents mounted in a TO- I 8 envelope: a MOS transistor of the p-channel Hence, ~ce will by a pproxim a tely 85 V.

150 151
The a.c. collector load is given by:

(vce)2 10

RL = - - .
2 P0
d to t
1%1
For P 0 = 2 W and RL = 1.8 kQ , the collector pea k current Uc) is
47 mA. To avoid di stortion owing to the current setting, a d.c. operating
current of 50 mA is required. With h FE = 20 for a lower limit tran sistor
5
BDI 15, I Bmax = 2.5 mA . The d .c. base voltage to the common line
(VR 5 ) is 3.5 Vat an operating current of the BDI 15 of 50 mA and at
VBE = 0.7 v. I
I
To calculate the hea t-si nk required for the BD 115, allowance must be /
made for an absolute maximum di ssipation (P 101) of about 6 W and a ....
maximum ambient temperature of 50 °C. As the maximum permissi-
0
ble junction temperature is 200 °C, the total thermal resi stance must be: P0 (W) 10

Ti - Tamb 200 - 50 Fig. 4. 16. Total distortion of the two watt amplifier as a fi111ction of output power at
R1h J-a = = 25 °C /W. 1 000 Hz.
Po 6
The thermal resistance R 1h J- mb of the BD 115 is 12.5 °C/ W, so the thermal
resistance from mounting base to ambient must be 25- 12.5 = 12.5 °C/ W.
This can be obtained by two methods:
(a) with the transistor mounted directly on a horizontally positioned
bl ackened aluminimum heat-sink of 30 cm 2 (R,h = 12.5 °CjW);
(dB I
(b) with the transi stor mounted via a mica washer on a blackened alum-
inium heat-sink of 50 cm 2 (R, 11 = 9 °C/W). The mica washer adds
0
a thermal resista nce of about 3.5 °C/W. ''
In the circuit diagram of the 2 W amplifier in Fig. 4.14, a tone control '
(R 1 , R 2 a nd C 4 ) has been included. The upper limit of the bandwidth
(- 3 dB) can be varied between I and 12 kHz by means of R 2 • The supply
2
is obta ined from the mid ta p of a 220 V a.c. turnable motor and the pick-
up element is connected via two 4.7 nF capacitors becau se otherwise
the common supply line might become connected to the live terminal
of the m ains.
The sensitivity for minimum devices is 140 mV for full output. 4
10' 10' fl Hz I
Jn Fig. 4.16 the di stortion is plotted as a function of output power,
measured at the primary of the output transformer. The spread in distor- Fig. 4.17. Frequency response characteristic of the two watt amplifier. 0 dB = 2 W.
tion for minimum and m aximum devices is negligible for output powers
below 1.5 W.

152 153
 The frequency response of the amplifier is shown in Fig. 4. 17, the - 3 dB 4.2.3 CIRCUIT 29 - FOUR WATT PICK-UP AMPLIFIER USING THE TAA320
point at lower frequencies being determined by the output transformer. AND TWO BO 1J5 OUTPUT TRANSISTORS
Fig. 4. 18 shows the power ba ndwidth for d, 0 , = 5 %.
Performa nce specification:
nominal power output 4 w into 800 n load
sensitivity (I 000 H z) for P 0 4W 100 mV
(for minimum devices)
frequency response (- 3 dB) 50 to 12000 H z
to tal ha rmonic distortion at P 0 = 4 W 5.5 %
voltage fe edback factor 4 (typ.)
nominal suppl y volta ge 200 v
current consumption 72 mA (tot a l)
!d Bi t-----t--t-tt+t+tt--H-t--l-H+H-- -l---Hl-+l+t+4--+-l--H.J-.l-l-lj
Fig. 4. 19 gives the circuit for a 4 W amplifier, using the integrated

r ----.----......-- ----Q +200V

I
I

- 2r-----i--t-Tttttllt"--1-1-t-ttit-t+--+-+-!-+t-t++f---+--+-+++++~

1 Rl G
Mil
log
volume

Fig. 4.18. Power bandwidth of the t wo watt amplifier at 5 % distortion. O dB = 2.2 W. C4 soon
1.2 nF
'4..7 nf C)
\H
4.7nF
C2

Fig. 4. 19. Circuit diagram of four watt pick-up amplifier.

circuit TAA320, in which two BDl 15 transistors are connected in a

single-ended Class-A push-pull configuration. Since the output transis-
tors are effectively in series for d.c., a supply voltage of about 200 V is
required. The drive signal for the upper transistor is obtained from the
voltage drop across the collector resistor of the lower one.
The a.c. collector load is 800 Q , so that either an 800 Q loudspeaker,
or an output transformer must be used.
With the exception of the output stage, this circuit is similar to the two

154 155
watt amplifier described under Circuit 28. Distortion as a function of out- 4.2.4 CIRCUIT 30 - fOUR WA TT TRANSFORM ERLESS CAR RADIO AM-
put power is given in Fig. 4.20. PLIFI ER USING THE T AA435 AND AD 161 /J 62 OUTPUT TRANSISTORS
Performance specification :
nominal power output 4 w into 5 n load
10 '·' sen sitivity (I 000 Hz)
R 4 = 220 Q
I R4 = I k!1

for P 0 = 4 W I5 mY 65 mV

'
for P 0 = 50 mW
input impedan ce
1.7 mY I 7.3 mV
220 kQ
I
frequency res ponse (- 1.5 dB) < 30 to 20000 Hz
total harmonic distortion at P 0 = 4 W < 1.0 % I < 0.6 %
(from 40 Hz to 12.5 kHz)
unweighted sign a l/ noise ratio 50 dB 59 dB
I
(ref. P 0 = 50 mW)
I weighted signa l/noi se ratio 58 dB 68 dB
' (ref. P0 = 50 mW)
/
' internal resi sta nce at output socket 0.21 n 0. 11 n
_,., da mping factor with 5 n speaker 23.8 45.4
nominal suppl y voltage 14 v

P0 {W)
F ig. 4.21 gives the circuit of a four watt amplifier which combines a
10
T AA435 with AD 161 / 162 transistors. This com bi nation provides an
excellent transformerless amplifier which follows the car radio trend
Fig. 4.20. Total distorti on of th e four watt amplifier at 1 kHz.

~l~+-------'--1
~
250 µF
ll SVI R2
IBO kJl
RI C2
input~ TAA435
SJkJl O.l µF Cl
200 pF
10
Rl
39 kJl

R4

Fig . 4.2 1. Circuit diagram of four watt car radio amplifier.

156 157
to smaller size. There are two versions of the ampli fier in which different
values of negative feedback resistors are used.
1---+--+-+-+-++Tti - - 40 Hz
- - 1kHz
1---+--+-+-+-++Tti

RS
dtot
10/0) 1-1-===jt=::t=::t=t±±±~~=:.1~2~. Sk~H~z-HH=++~==t=t=mJ+mi
l-_ _ji---l---l--t-+-t+++----+-+--HH-++H---t"-t-1ttt--t'"rtt1
R1
21k0

100--.-------r----r-~
. ..
6 7Z55l27. 2

5 10
Fig. 4.22. Circuit of the TAA435 .
Fig . 4.24 . Total disturtion of the four watt car radio amplifier, where R 4 = I kO..

+1
Id Bl

r- 1

llllllllill~~0Hlzilllllllllll
-1 .v 4
1kHz
dtot >-------+----+- 12.SkHz
-2 (
0
/ol

~ 2
-3

-4

-5
10 10' 10' fl H
z 105
I
--
[J'

Fig. 4.23. Frequency response characterisric of the four watt car radio amplifier. 0 dB =
- - ·- 1--
2 V. Curve I: with I k O.feedback resistor; Curve 2: with 220 O.feedback resistor.

An integrated circuit T AA435 is employed serving as pre-drive and Fig. 4.25. Total distortion of the four watt car radio amplifier, where R4 = 220 0..
driver stage for the complementary symmetry output stage which
uses AD 16 l / 162 transistors. Fig. 4.22 shows the ci1 cui t of the T AA435.

158 159
 With a value of 220 Q for the feedback resistor R 4 , an input sensitivity 4.2. 5 CIRCU IT 31 - RECORDING AND PLAYBA CK AMPLIFIER USING THE
of l 5 mY for 4 W output will be obtained. This value is adequate for all TAA310 INTEGRATED CIRCUIT
normal applications except a magnetic pick-up. A considerable im- Fig. 4.27 shows a block diagra m of a complete tape recorder incorporat-
ing the TAA310 which has been designed for operation as a recording

'
-1i----t--fti,t+ttfft--t-++++++++----+-+-H+ti4--~:........i-1--w+~

Fig. 4.27. Block diagram of tape recorder using the TA A3 IO .

R = record, P = playback. Poi/Ifs A and B

refer to input and output respectively of
the circuit shown in Fig. 4 .28.

Fig. 4.26. Power bandwidth characteristic of the four watt car radio amplifier. a nd playback pre-amplifier with a tape speed of 4.75 cm/s (1-7/8 in/s)
OdB = 4.6 W. and an operating voltage of 7.0 Y.
Curve 1: with 1 kD.feedback resistor; The circuit of the recording and playback amplifier is given in Fig. 4.28
Curve 2: with 220 D.feedback resistor. and that of the TAA3 I 0 in Fig. 4.29. The first transistor of the input
stage is a low noise transistor with a maximum wideband noise of 4 dB
provement in p_er~ormance is, however, obtained by increasing the negative measured in a bandwidth from 30 Hz to 15 kHz. The current setting
feedback and 1t 1s recommended that unless a sensitivity of l 5 my is (100 µA) of this transistor is chosen as a compromise between gain,
essential, R 4 should be increased to 1 kQ . input impedance and noise performance.
The frequency response characteristic is given in Fig. 4.23 for the two During recording the volume is controlled between the first and second
values of feedback resistor. Figs. 4.24 and 4.25 show the total harmonic transistor by R 2 , the slider of which is earthed , and a control range of
distortion at three frequencies , 40 Hz, 1 kHz and 12.5 kHz, for R 4 = approximately 70 dB is achieved.
l kQ and R4 = 220 Q, respectively. The power bandwidth characteris- Transistors TR 3 and TR 4 form a differential amplifier. This type of
tics are given in Fig. 4.26. circuit was chosen because it is a simple method of obtaining effective
negative feedback for both a.c. and d.c . Only TR 3 is involved in am-
plification, whilst TR 4 serves for the negative feedback. The operating
points of TR 3 and TR 4 are set so that approximately the same currents
(0.35 mA) flow in both transi stors.
The operating point of the output transistor TR 5 must be located in the
centre of the load lines so that the highest possible undistorted power
output can be obtained. A strong d .c. negative feedback applied to

160 161
 TR 4 of the differential stage causes the operating point to adapt itself 11
automatically to the supply voltage.
If the recording head of a tape-recorder is fed with a current of con-
stant amplitude, the voltage delivered by the sound head on playback is
C10 2SµF
>-----.-~-.---:+41f----os

10
C1
en V111 11 l--+--+--+-1-+-.._....+---+--+--+-+-t-++-1+---+---+--+--+-t-+++1
A o-jlh+r--- + - - - ' - l TAA310
0.61.µF 56 Vh!1k H1 l1-- +--+--+-1-+-.,_....4---+--+--+-+-t-++-l+-- -+----<---+--+-t-+++1
nF
ldBI
l
R1 4.7mH
47k0
R10
150k0 R15 I/
82n ,
I I
I I -10 "---h<-1--+-+++Hl+----+--+-+--+-+++-l+---+----+-+-t-1H-t+i
~---------------------~
,
Fig. 4.28. Circuit diagram of the recording and playback amplifier . Switches S 2 and S 1
,
,,
are in position 1 for recording and position 2 for playback.

-2~0L,--L--'L....J.-l-.L.J....U1~
0'_ __._..___'--'-.........~,o-4--+-f-l~Hz-)~~~10'

Fig. 4.30. No-load voltage of playback head as afi1nctio11 offrequ ency.

10

strongly frequency dependent. The dependency of the playback voltage

on the frequency is shown in Fig. 4.30 for a tape speed of 4.75 emf s
(1-7/8 in/s).
For low frequencies, the voltage rises almost linearly with the frequency
because for these frequencies the tape flux is constant and the voltage
induced in the head is proportional to the change in flux dcJJ/dt . Above
2 kHz there is a decrease in the induced voltage with rising frequency due
to the ~ elf-demagnetisation of the tape, the gap width of the head and the
iron losses.
The 3 dB points of the overall amplitude response of a home tape
recorder at a tape speed of 4. 75 cm/s ( 1- 7/8 in/s) should be at 80 Hz
Fig. 4.29. Circuit of the T AAJ 10.
and 6300 Hz. The overall amplitude response should lie within the toler-
ance zone shown in Fig. 4.31 .
In order to meet these requirements , the frequency respon:;e shown
in Fig. 4.30 must be corrected. Switchable equalizing networks in the

162
163
recording/playback ampli fier ensure the required increase m the treble
( 13 dB) during recording and in bass ( 17 dB) during playback (dB values
20
with respect to !000 Hz). The nature and range of correction are stand ard-
ized so that the tapes p layed on different recorders are interchangeable. ldB I
I\
Fig. 4.32 shows the nega tive feedback equalizing networks employed , \
10
whi lst Fig. 4.33 shows the frequency responses obtained . The load resis- IJ \
II \
tance of the amplifier will differ for each source and , s111ce no single II \
___ ...... v \
1

v
I/ \
1

-10
I
'.

-2 0
Fig. 4.3 1. Tolerance zone of overall amplitude response. 10 10' 10' fl Hz) 10'

( a) recording amplifier

network can be used , it is necessary to calculate the va lues of the input

load resistances required. To ensure that a flat characteri stic of the loaded
signal source is obtained and high frequency attenuation is avo id ed ,
the - 3 dB point of the characteristic shou ld be at a much higher frequency
20
and if a value of 30 k H z is taken the overall amplitude characteristic at
6.3 kHz will then be scarce ly affected. (dB I I/

10
J

''
I
J
' ~
I'
---i C9 CB ~

R9
R11 1'-
......
TR4 TR4
.....
Cll - 10
" I"\
I'

R10 R12

-2 0 4
10 10' 10' fl Hzl

( b) playback amplifier

( a) treble boost on recording ( b) bass boost 011 play back

Fig. 4.33. Frequency response characteristics.

Fig. 4.32. Negative f eedba ck equalization 11et works.

164 l 65
 As an example, two of the required load resistances will be calcul- thus made suitable for playback , radio , dynamic microphone, and dynam- i I

ated. For the radio input, it shou ld be assumed that the cable between ic pick-up. I 1

radio and tape recorder has a length of 2 m and a capacitance of For the crystal pick-up and crystal microphone inputs the required
40 pF/ m. The load resistance that should not be exceeded is load resistances are determined by the permissible attenuation of the
input voltage at lower frequencies. Because the input resistance of the
R1 = l/ (2nf_3doC) = (2 n X 3 X 10 4 X 8 X J0- 11 ) - 1 = 66 kQ amplifier will be chosen between 8 and 50 kQ (playback and radio
For the playback head input, the series inductance of the head (L = recording), the required input resistance for crystal pick-up and crystal
40 mH) will determine the high frequency roll-off. Here the required microphone ( < I MQ) will have to be obtained by means of an extra
load resistance will be resistor which wi ll then have to be switched in series with the input.
The output requirements of the amplifier are for playback 0.6 V (line
R1 = 2 nf_3du L. = 2n X 3 X 10 4 x 4 x 10- 2 = 7.5 kQ, output) and for recording 0.75 V across 5 n. The playback head (same
at least. Table 4.1 gives the source resistance of variou s signal sources as recording head) is fully modulated with 150 µA at I kHz (the bias
together with their e.m.f. and required load resistances. current is I mA at 35 kHz). To get full magnetization at high frequencies
Volume contro l cannot be incorporated in the input network of the without overmodulation at low frequencies a current drive of the head is
amplifier because it would entail an unacceptable noise and hum level, required. With the amplifier described this current drive can be realized
by connecting a resistance of 5 kQ in series with the head. Full magnetiz-
ation thus requires an r.m.s. voltage of 0.75 V across this series combin-
Table 4.1. Values of L oad R esis1a11ce for Difj'eren/ Signal Sources
ation.
so urce required load From Table 4.1 it can be seen that the input is lowest for recording
sign a l source e.m.f.
impedance resistance with a dynamic microphone (0.5 mV). Because the input resistance of the
amplifier is high in relation to the source impedance of a dynamic micro-
radio (from detector) 0.5 MO IV < 50 kO
crysta l pick-up 3 nF
phone, it may be assumed that the full e.m.f. of the microphone is
1.6 v > 0.5 MO
dynamic pick-up 80 kO I to 2 V 10 kO
available at the input. For recording, the required voltage gain will thus
crystal microphone 1.5 nF 5 mV > I MO be Gv = 0.75/(0.5 x 10- 3 ) = 1500 (64 dB). To this must be added
dyn a mic microphone
playback head
200 o
L = 40 mH
0.5 mv • 10 kO 13 dB for treble correction, which makes the total G" = 64+ 13 = 77 dB.
typ. 0.5 mV > 8 kO During playback the required voltage gain is Gv = 0.6/(0.375 x 10- 3 ) =
R = 100 0 min. 0.375 mV
1600 (64.5 dB) for a minimum head, to which gafn an extra 17 dB must
• At a so und pressure of I µ bar (normal speech at a di stance of 0.5 m from the be added for bass correction. Thus the total gain during playback has
microphone). to be at least 64.5 + 17 = 81.5 dB. Jn addition , a further gain reserve
is desirable to minimize the effects of spreads between the individual
especially when the available input signal power is low (dynamic micro- integrated circuits on gain by means of a ppropriate negative feedback .
phone). As a result some of the input sources with a higher power level The minimum gain of the amplifier described here is 83 dB .
such as radio, will tend to overload the input. Without precaution~
a~d because the chosen system of volume control also influences the neg-
at1v~ a.c. feedback at the emitter of TR 1 , the input resistance might
attain values higher than 50 kQ (depending on the setting of the volume
control). The requirement to keep the input resistance of the amplifier
belo w 50 kQ will be met by shunting a 47 kQ resistance across the input.
The lowest input resistance will then be 8.3 kQ . The input impedance is

166 167
 5. Loudspeaker Systems

5.1 Principles of Loudspeaker Selection

5.1.1 SURVEY OF LOUDSPEAK ERS

The selection of a loudspeaker for a reproducing system is determined by

a number of factors , including the frequency range required , the permis-
sible distortion, the power requirements and the method of mounting
the spea ker.
For applications where a restricted frequency range is adequate,
such as with inexpensive radios, record-players and tape recorders,
the Standard range of loudspeakers offers a wide choice. The resonance
frequencies of these speakers varies from 75 Hz for an 8-inch unit, to
360 Hz for a 2.5-inch unit. High frequency roll-off is extended to 10 kHz
for a few types. These a re listed in Table 5.1 .
Where a wider frequency band is required the High Quality range of
speakers given in Table 5.2 provides an excellent soluti on. These speakers
are all double-cone full-range types and , with exception of the 5-inch
and 7-inch units , they have high sensitivity due to the powerful Ticonal
magnets used. This range of spea kers is intended for use in acoustic
enclosures of good quality sound installations.
Finally, there is the High Fidelity range of speakers given in Table 5.3.
For these, the frequency band is divided by means of a cross-over
filter network. The specially designed bass units, or woof ers, provide
an excellent low frequency respon se when mounted in the recommended
enclos ures. Each woofer has a very low resonance frequency, that of the
12-inch woofer being only 19 Hz. High frequency units, or tweeters,
are also available and a smooth frequency characteristic to 20 kHz will
be obtained with the I-inch tweeter. It is therefore possible to obtain an
overa ll frequency characteristic, match.i ng that of the high fidelity am-
plifiers, with this range of speakers.
The most irritating kind of distortion is that occurring at high fre-
A selection of the high quality and high fidelity loudspeakers which are used to build quencies on transients. The I-inch tweeter, which operates on the same
the recommended systems. principle as all moving coil speakers, uses a specially contoured thermo-
plastic diaphragm resembling part of the surface of a sphere instead of a
cone. This results in negligible interference between the waves leaving
the surfaces of the " dome" and a smooth polar response is obtained as
shown in Fig 5.1. Because the moving parts of the tweeter are extremely

168 169
Table 5. 1. Standard Range of Loudspeakers Tabie 5. 1. (continued)

powe r I reso - I
power I reso- I
overa ll overa ll tota l ha ndling impe- nance tot a l flux tyµe
tota l handli ng impe- na nce total fl ux type d iameter dept h capacity dance frequen - fl ux density nu m be r
diameter depth ca pacity dance freque n- fl ux density number
(mm) (mm) (mm) (W) (Q) cy ( Hz) (µWb) (mT)
(mm) (W) (Q) cy ( Hz) (µWb) (mT) I
I
7" round
2t" ro un d
64 20 0.5 4 360
I 63 740 AQ2070/Z4
166 47 3W 4
8
115 189 700 AD709 1/X4
AD709 1/X8
8 AD2070/Z8 800 AD7091 /X800
15 AD2070/Z l 5 4 95 175 980 AD709 1/ M4
25
I A D2070/Z25 8 I AD709 1/ M8
800 AD709 1/ M800
3" round 58 4W 4 105 AD7080/ M4
81 28 1.0 4 250 63 740 AD3070/ Y4 8 AD7080/ M8
8 A03070/ Y8 6W 4 11 5 AD7080/X4
15 A D 3070/ Yl5 8 A D7080/X8
25 AD3070/ Y25
81 0 28 1.0 150 250 63 740 8" round
AD3370/ Y l 50
206 68 6W 4 95 177 980 AD8080/X4
4" ro und 8 AD8080/X8
105 29 l.O 4 75 AD8080/ M4
4 200 63 740 A D4070/ Y4
8 8 AD8080/ M8
A D4070/ Y8
15 A D4070/ Y l 5 3" x 5"
25 A D4070/ Y25 ova l
105 39 3.0 4 165 177 1000 A D4080/X4 76 x 13 1 42 2 4 200 118 1000 AD3590/X4
8 A D 4080/X8 8 A D 3590/X8
15 A D4080/X l 5 15 AD3590/X l 5
25 A D4080/X25 50 A D 3590/X50
105 37 2.0 8 180 118 IOOO AD4090/X8 400 AD3590/X400
2.0 15 175 AD4090/X l 5
0.6 400 190 A 0 4090/X400 3" x 8" I
ova l I I I
5" round 83 x 206 I 51 I 2 I 4 120 177 1000 AD3880/X4
129 48 3W 4 155 180 1000 AD5080/Z4 8 A D 3880/X8
8 A D5080/Z8 15 AD3880/X 15
15 A D 5080/Z l 5 54 2 4 120 I A D 3890/ X4
25 AD5080/Z25 8 AD3890/X8
4W 4 130 AD5080/ M4 800 125 I AD3890/X800
8 AD5080/ M8 4" x 6"
15 AD5080/ M l 5
oval
25 AD5080/ M25
6W 102 x 154 48 3 4 155 180 1000 AD4680/Z4
4 140 AD5080/X4
8 8 AD4680/Z8
I A D5080/X8
15 15 AD4680/Z 15
AD5080/X l 5
25 25 A D4680/Z25
AD5080/X25

170 171
 Table 5. I . (co ntinued)
'<!' 00
'<!' 00 r- r- r- 0.. 0..
I "'::E V)

overa ll
diameter
tota l
depth
power
ha nd ling
capacity
impe-
dance
I reso -
na nce
frequen-
to ta l
flu x
flu x
den sity
type
number
&~
....

....» E
:::>
::E ::E
00
'° '°
00
0
'°
0
r-
::E
0
"'
-
0
---------
::E ::E ::E
00V"'IV)V)
\C VI V"'I VI VI

-----
N N N
:c :c
N N

(mm) (mm) (w) (fl) cy ( Hz) (17 Wb) (mT)

c: "'"'
00 0 0 00000
I <!'. <!'. <!'. <!'. <!'. <!'. <!'. <!'. <!'.

4" X 6" I I
ova l I
102 x 154 48 4 4 I 125 AD4680/ M4 »
8 AD4680/ M8 >< .~ !=' 0 0 g g 8 0000
lecu~
E
00
"' ~ ~ ::: 8
15 AD4680/ M 15
:i
cc:
'O
0\ 0\ M 00 00
--
6
25
4 140
AD4680/ M25
AD4680/X4 I - ~

0
"E
:::>
8 AD4680/X8 >
15 AD4680/ X l 5 ,.......
25 AD4680/X25
"(; >< .0
.... :::> ;:::
v
0\ '°
N
M
00
0
00
0
00
'°
N
0
00 V
0 0
-
-~
"'
52 3 400 124 151 850 AD4690/ M400 .sq:::~
N
'<!' "' 0\
0\
--
'i::t' 0\ M N

"'
N
800 135 AD4690/ M800 '-
4 4 135 AD4690/ M4 -- 0
cu
....
cu » :::>
u u
8 AD4690/ M8 c: c:,....... "'0
cu N
6 4 140 "'0
c: :::> :c "' "' 0 0 0 0000
'° uc:
I AD4690/X4
"' cu
~ <!::
C' ~
00
"' '° "' "' V) V) V)
cu
5" x 7" 'O
cu
ova l -- t;
cu

.::"'
133 x 183 cu
58 3 4 11 5 117 980 AD5780/X4 u
c:
8 AD 5780/X8 "',.......
'O c: v 00 r- V'lr""'-t'VOO ,.......
15 AD5780/Xl5 cu~
0. "' "' r-
"'
25 AD57 80/X25 .§
4 4 100 AD5780/ M4 - - -- -- --
8 AD57 80/ M8
15 AD57 80/ M 15 ... Oil
o ·-
»
c: ....
·u .--
,....... ,.......
"'0 '°
"E
:::>
25 AD57 80/ M25 "'0 :0
c "'0. .._.
;::: '° - 0
-
0 0000
- N N ""2" 0
>
6" x 9"
0. ..c:
"'
"'
u
"'cu
oval -- ~
161 x 234 68 6 4 85 174 980 AD69 80/X4 0\

-s e
'-
8 AD6980/X8 c; 0 MOOO
0
4 72 AD 6980/ M4 o fr
.... 'O ~
E "''° r-
-'°
M
----
N\Ct- t- cu
:;
8 AD6980/ M8 "'0
1-
uc:
cu
.... 'O

=E E EE
cu ,.......
r- -'° t;
cu

- '° '° "'M-

o:> 0\ cu
-
I
N 0
cu '° N N N "' I
~ ~
I~ I
'-' .:: II
b 'N ,.......

"" - - I

172 173
 .~
-;;; ligh t in weight, virt ua ll y no time is lost in getting the diaphragm mov ing
v 00 v 00 ;::: ;::: v 00 v 00 v 00 ;:::
....0) v 00 v 00 c:
;::: in response to a signal. The step-function response of the tweeter is there-
.,... .,... ~~
~~ .,... .,... ~~
Oil
f- f- f- f- ·;;;
l) .D
o, E 00 00 00 .,...
.,... .,...
.,... :o~
.,... .,... fore excellent, and it is able to hand le high frequency transients without
r- r- 0)
.::- :::l
c: '° '° 0
00 N N
0 0'°.,... 0'°.,... 0
'°r- 0'°r- 0'°
00 00
'° 0 0 N N -5
0
any noticeable distortion. When the I-inch tweeter is correctly mounted
ClCl 00 ClCl Cl Cl Cl Cl Cl Cl Cl Cl 'O
<< << << << << << << "c: and the recommended cross-over network is used , distortion is less than
.... I % a nd is therefore inaudible .
---· -- -- ---- ---- -- ~
ti: Overloading is a lso a common cause of d istortion, often due to bass-
.::- ....
0)
boosting the signal whe n the power hand li ng capacity of the speaker has
x -~ p 0
0
0 0 0 0
0
0
0
0
r-
>
9
~~s N °''° M

°' °''° °' °' 0

"'"'
been reached but, prov ided that t he frequency and power requirements
2
u
are carefu ll y studied before a choice of speaker is made, distortion shou ld
- - - - - - - - 'O
0)
be a minimum. Development work aimed at reducing inherent distort ion
'O
c:
0)
is constant ly in progress and modern , well-desig ned speaker units em-
0
0 E ployed in a properly engineered system can now be relied upon to con-
°'
M N E
0 tribute on ly a min ute amount of distortion to the reproduced sound.
u
0)
....
__ __ O)

-5
~
'- 5. 1.2 POWER REQU I REMENTS
.,...
0 "'00 0
0
8
00
00
N N N :; The maximum loudness level depends on the background noise in the listen-
0.
c:
ing room and the dynam ic range of the reproduced programme.
0)

- - -- - - - - - - - - - - - - - - - -5 Assuming a room noise level of 30 phons and a dynamic range of 60 dB,

<lJ
u
the maximum loudness level required wi ll be 90 dB above threshold.
c:~
Because of the large number of variables affecting room acoustics
-g" c:
'-" it is on ly possible to approximate the power requirements. Realistic
0.
E results can be obtained using the fo ll owing simplified method of deter-
mination. The acoustic power at thresho ld at I 000 Hz can be taken to be

v
Puir es1w1d = 4 x 10- 14 - watts,
T

whe re V is the room vo lume in cubic metres and T is the reverberat ion
time in seconds. For small rooms, up to say 100 m3, T has a value of
approximately 0.5 seconds.
Assuming that the listening room measures 5 m long, 4 m wide and
3 m high, its volu me wi ll be 60 m 3 • For the maximum loudness level of
90 dB above threshold referred to above, the acoustic power level be-
....
- 0) comes:
~~E
<lJ E E
I ~.~'-'
'O
= 4.8 mW.
0.5

174 175
 Fig. 5. 1 (a). Polar response at 1000 Hz.

60~~~~~~.,.-~----.~..,-,...,...rr-~-=.

dB f--j-+++++-~~+--+~-+-+-H-1-+-~--l

Fig. 5.1 ( b) . Polar response

at 3000 Hz.

Fig. 5.1 ( e). Frequency response curve

of the AD0/60/ T.

500 1000 2000 5000 10000 20000

f(Hz)

Fig. 5. 1. Performance characteristics of the I-inch tweeter, A D0/60/ T.

Fig. 5. 1 ( c). Polar response at 10000 Hz.

177
176
 Taking the efficiency of the speaker system as I %, the electrical power
input required will be:

100 x 4.8 mW = 0.48 W.

The following discussion is based on the desire to obtain the greatest

realism in the reproduced sound and applies to only high fidelity instal-
lations. 1t is realised that there are very many applications where the
specifications may be relaxed , but it is important to remember that the
power handling capacity of the speaker must always be adequate, and
preferably with a small margin , to accept the peak output power of the
amplifier without distortion.
It can be seen from the Fletcher-Munson contours in Fig. 2.1 on
page 4 that the 90 phon curve is substantially flat over the bass region
and so the relatively low electrical power of 0.48 W in the above example
will be adequate to reproduce the signal at a level of 90 dB above the
I000 Hz threshold over the whole band.
However, at 40 Hz the threshold is nearly 60 dB above the threshold
at IOOO Hz and at low volume levels over the bass region the sensitivity
of the reproducing system should be increased , otherwise the signal
may be inaudible. The physiological volume control (contour) has this
property and is a good solution , but it is more usual to employ bass
boost using the normal tone controls to overcome this bass deficiency.
A bass boost of around 20 dB maximum is normally provided but,
unless a contour control is employed , the increase in output can occur
at all signal levels , not only those of low volume. In order that distortion
is prevented under bass boost conditions, the power handling capabilities
of the system should be increased. In the example quoted , the power
requirement becomes 0.48 x 10 2 = 48 W .
The ear is more tolerant of high volume levels where stereo reproduc-
tion is concerned. If the calculated power requirement of 48 W is applied
to each channel , the power handling capacity of each speaker system and
the power output of each of the reproducing amplifiers under bass boost
conditions should therefore be at least 48 W and the acoustic power
requirements will be correctly satisfied for each channel individually.
The one-inch "dome" tweeter gives superb reprod11ction of all frequencies from This does not mean that the full 96 W will ever be required at one time,
~000 Hz to over 20 000 Hz . it has a dome-shaped diaphragm which gives since the total signal is divided between the two channels , but only that
a nearly ideal polar diagram and, beca11se of the low 1nass of its mo ving if all the signal is required on one channel there will be adequate output
parts, transient distortion is negligible. to meet the acoustic power requirements.

178 179
 Such is the ideal case , but in practice the power requirements can 20

normally be reduced. By how much depends on the positioning of dB

the speakers. When a speaker is mounted close to a wall the effective
radiating area is theoretically doubled at low frequencies, owing to the ~

reflection of the wall. A room corner has three mutually perpendicular ,,. ~

surfaces and thus the effective radi ating area for a corner speaker is / I"-
doubled three times. A gain of 8 would , however, on ly be possible if
the surfaces had a reflectance of I00 % and , allowing for the absorption
-20
''
of the walls, together with that of fitted carpet, adjacent curtains, etc.,
which reduce the effect, a reflectance of 50 % would be more likely. - 40
~

.In practice, a gain of from 2 to 4 times (3 dB to 6 dB) can be expected.

'\
The power requirements of the system can thus be reduced by this amount '\
and , in the example quoted , the 2 x 48 W system could be reduced to - 60
4
10 10' 10' 10 f (Hz )
2 x 25 W. The system will be operating at a very much lower level that
this most of the time. Individual cases vary, however, and the speaker Fig. 5.2. Relati ve acoustic power levels arising in speech and music ( DIN 45573) .
positioning and room absorption should be carefully considered before
the calculated power requirements are modified.
The mono case is slightly different. An intensity level of 90 phons
when reproduced by a mono system sounds very much harder and more
irritating than when stereo is employed , or when it is produced by a live
5. J .3 METHODS OF MOUNTING
orchestra, since the directional discrimination of the ears is of no im-
portance as the sound is emitted from one small source. This leads to a The method of mounting a loudspeaker has a profound effect upon its
preference for reduced levels of loudness and smaller frequency range performance. The frequency response and power handling capacity are
with mono reproduction . Something less than the calcu lated power level particularly affected where sealed enclosures are employed.
of 48 W will be required and in practice a 40 W installation wou ld be The result of using a baffle board is to remove the effects of acoustic
considered very satisfactory. Again , the system will be operating at a short-circuiting and thus improve the bass response. The larger the
very much !owe! level most of the time. baffle board the greater will be the improvement.
Speakers are tested by applying a signal, of the same spectrum as shown Small open-back enclosures, or enclosures with a slotted or perforated
in Fig. 5.2, to the speech coil. The signal is obtained from a " white " panel at the rear, are an improvement on the open baffle but it is only
noise generator, via a special filter, and corresponds to a frequency dis- when the enclosure becomes nearly air-tight that the performance is
tribution approximately that of the relative acoustic power levels arising greatly modified. Vented enclosures such as the bass-reflex enable the
in speech and music (DIN45573). The power handling capacities given volume of air to be used to control the acoustic response.
in the Tables are, therefore , for an input signal in conformity with the For good low frequency reproduction with a vented enclosure a large
frequency distribution of Fig. 5.2. The power handling capacity of the cabinet is necessary but, because of the trend toward miniaturization ,
tweeters, however, also depends on the cross-over frequency and the the original bass-reflex enclosure has fallen out of favour on account
higher this is the greater will be the power handling capacity. For the of its size. A modern loudspeaker, specially designed for use in a sealed
I-inch tweeter AD0160/T , the power handling capacity is 40W (input enclosure, can produce a comparable frequency response to that ob-
to the filter) for a cross-over frequency of 4000 Hz, or 20 W for a cross- tained with the older bass-reflex in a fraction of the volume . The design
over frequency of 1500 Hz. of sealed enclosures is discussed briefly below.

180 181
 I
1:
!
5.2 Sealed Enclosures for Hi-Fi Systems the resonance frequency to ri se. The new re sonance frequency now
becomes: :
5.2.1 PRINCIPLES OF DESIGN

Under " free-space " conditions, normally achieved on ly in an anechoic

}.;ys = ~-
2n
VS, Md+ Sb ' (5-5)

room, the resonance frequency of a loudspeaker is given by:

where f ,ys is the resonance frequency of the syste mas a whole, comprising
I 1/ SS the speaker and the enclosure, and Sb is the s tiffness of the enclosure.
fo =-
2n Md
v- , (5-1) Combining equations (5-1) and (5-5) the prop ortional increase in fre-
q uency becomes:
where S, is the stiffness of the speaker cone and suspension and M 11 is
the dynamic mass. f~ys _ S, + Sb _ Sb )
The dynamic mass , Md = M c +
M" , where Mc is the mass of the mov-
--
j~
- v - 1
-
s,
;
- v I + -
s,,
. 1
; (
(5-6)

ing parts and M" is the mass of the air moved on both sides of the cone.
From eq . (5- 1), the stiffness of the speaker may be calculated:
Table 5.4. Values of Mass and Stijfi1ess for Hig h Fidelity W oof ers
(5-2)
woofer nomin al effective
The method of determining both the stiffness and the dynamic mass is type dia. piston dia . Mc M" ') Md SS fo
to take two measurements. Firstly, the resonance frequency j~ is found (in) (mm) (g) (g) (g) (N / m) (Hz)

by applying a contro lled signal to the speaker in an anechoic room. AD5060 5 90 5 0.3 5.3 530 50
AD7065 7 120 13 0.7 I 3.7 430 28
A known mass m is then applied to the cone and the new lower reson-
AD8065 8 150 24 1.32 25.3 790 28
ance freq uency,};,, , is determined. From eq. (5-2), ADI055 10 190 28 2.0 30 480 20
ADl256 12 240 45 5.4 50.4 730 19
(5-3) 1) The air load may be calculated from the equation M a = 2 x 0. I35 (!A 2 /r, where

12 = density of the air, A = area of equivalent " piston " a nd r = cone radiu s.
Since the value of the stiffness S, was the same during both measure-
ments, eqs (5-2) and (5-3) may be combined , from which

4 n 2/,,,2 (Md + m) = 4 n 2j~ 2 Md

The stiffness of an enclosure is given by:
(! C2 A 2
and Sb = - - - , (5-7)
,/ ' 2
v
Md = ___!!Y_"_'- . (5-4) where
j~ 2 - f,,,2 e density of the air,
The value of Md obtained from eq . (5-4) may be substituted in eq . (5-2) c = velocity of sound ,
a nd hence the stiffness s. calcu lated . Typical values of mass and stiffness A = area of the equivalent " piston " ,
are given in Table 5.4. V = enclosure volume.
When the speaker is mounted in a sealed enclosure, at low frequencies The stiffness thus depends on the size of baffle hole and volume of the
the internal volume of the enclosure will act as a stiffness which must be enclosure. The stiffnesses of enclosures for va rious woofers are given
added to the stiffness of the loudspeaker suspension system . This ca uses in Table 5.5.

182 183
 f

Table 5. 5. Stiffneses of Various Sizes of Enclosures

enclosure woofer st iffness

vo lume dia. Sb
(I) (in) (N/ m)

3 5 1900
7 7 2600
10 7 1820
15 7 1210
15 8 2900
20 8 2200
25 8 1750
35 10 2100
40 10 1830
50 12 5600
80 12 3500

If the woofer baffle hole is less than one-third the area of the baffle,
the rise in the resonance frequency is approximately 7 % less than that
obtained from eq . (5-6) and the resonance freque ncy is then given by :

f~:_s = 110.87(
I + Sb) . (5-8)
Jo Ss
This is shown graphically in Fig. 5.3 .

f sys L---1--l--1---l--l--l-1-1-1-- -l--+-H--H-t+t---j---T-t-"H-TtTI

~ L---1--l--l---l--l--l-1-1-1---l--+-t-+-H-+tt----t-;t--t--t-ti---tiT

u L---1--l--l---l--l--l-l-1-1---l--+-+-+-H-+t+--~,'-t----t---t-ti---ti1i
I
I

A high fidelity speak er 0 11 test in the anechoic room . Th e speaker is held

in a clamp on the test fix ture and the microphone positioned 50 cm away.
1
Th e poly urethane ' wedges' which cover the walls and ceiling absorb all ~o io2 10
3
~• 10
4

reflections and the room simulates 'free space' conditions. Wedges are Fig. 5.3. Proportional rise in the resonance fr equency of a loudspeaker when fit ted in a
also fi tted on the floor belo w the metal grid. sealed enclosure.

184 185
5.2.2 FREQUENCY RESPONSE AND ENCLOSURE VOLUME

The frequency response of a loudspeaker is determined by recording

through a microphone, 50 cm away along the axis of the speaker, the 90
I
response to a constant voltage signal of 50 mW ( 12- inch full range types
25 mW) as the frequency is swept slowly to 20000 Hz. The test takes
t .,. \
I Hz I f 0 ::2 8Hz

\
~

place in a n anechoic room and the result is p lotted with a pen recorder.
0 dB on the response curve corresponds to 52 dB above the threshold 0

of hearing (2 x 10 - 4 µ bar). Fig. 5.4 shows a typica l response curve for

a full range high quality speaker. \
I~
---
0 0
I
dB
1"1. if'"..iV\ .............
-
81/i " high quality
40

30
!/ _,.....--- \
v
20
v
l,..---"
~ 3 00 20 30 Vlll 40
10

0 Fig. 5.5 ( b) . 7-inch woofer, AD7065/ W.

o 50 100 200 300 500Hz 1 2 3 4 5 10kHz 20

Fig . 5.4. Frequency response cur ve of a typical fu ll-range loudspeaker, obtained without
baffle in an anechoic room .

110
90
'
f sys
(Hz I f,, . I\
\
f 0 : 50Hz
- I Hz I

\
f o=28Hz
-
90

\ J\. 70
\
r-- ·
~
~ f'...... "' ~
70
............
r--r-- ,..__ so ......._
r--- r----.
500
10 15 Vil) 10
30 0 10 20 30 Vl ll 40

Fig. 5.5 ( a) . 5-inch woofer, A D5060/ W.

Fig. 5.5 (c). 8-inch woofer, AD8065/ W.

Fig. 5.5 . Variation in sy srem resonance fr equency wirh enclosure volume for lypical
samples of rh e highfideliry woofers.

186 187
 When a loud speaker is fitted into a sealed enclosure, its frequency
response is considerably modified. For example, bass roll-off below
1H950l
0 resonance frequency which normally takes place at 18 dB/octave then

f .,
(Hz I
. becomes 12 dB/octave. The lower the resonance frequency of the speaker,
and hence the combination of speaker and enclosure, the better will be
~
the bass response. Fig. 5.5 shows the variation in reson a nce frequency
I'-.
0
""' with enclosure volume for typical samples of the high fid elity woofers.
The high fidelity standard DlN45500 defines the requirements for

""' ~
~"--....._
frequency response by reference to a standard curve, which is shown in
Fig. 5.6. When a frequency response curve has been determined for a
4c
r--_
- t--
loudspeaker system, the curve of Fig. 5.6 is overlaid on it, the middle
of the standard curve adjusted to the average of the loudspeaker system
response. Provided th a t the system response lies within the upper and
lower limits defined by the top and bottom lines respectively of the over-
20 lay , the frequency response conforms to the high fidelity standard.
0 20 60 Vlll BO
Fig. 5.7 shows the frequency response curve of a loudspeaker system
Fig. 5.5 ( d) . 10-inch woofer, AD/055/ W. employing one 12-inch woofer, four 5-inch mid-range speakers and four
I-inch tweeters. It can be seen from Fig. 5.8 that the loudspeaker system
conforms to DIN45500 when the standard curve is superimposed on the
response curve.
For details of suitable enclosures the reader is referred to the Applica-
0 7Z59 504
tion Book " Building Hi-fi Speaker Systems " . Although this book was

.
f .,
!Hz I
\ written primarily for the hi-fi enthusiast, constructional details of 11
different enclosures are provided , together with the frequency charac-

\ .._______
f o= 19Hz
teristics of 24 speaker systems.
0
'

40
"""' ~
~ I'-....
~
40 BO 120 VI ll 160

Fig. 5.5 ( e) . l 2-inch woofer, AD1256/ W.

188 189
 SO.----~.----.---.-.,-..-,.-no~""T"--,""T"'T'T"TTI-r--~;-...,-,- r"TTTTT~'"""T'---,

dB 1--~1--+-1-+-++1-tt-~-+--+-+++++1+-~+--t-+-t+t+tt~--+---i

301--~1--+-1-+-++1-tt-~-+-+-+++++1+-~+--+-+-+-+++++~-+---i

201--~1--+-1-+-++1-tt-~-+-+-+++++1+-~+-+-+-+-+++++~--+---i

101--~t--+-1-+-++1-tt-~-+--+-+++++1+-~+--t-+-+-+++++~--+---i

01--~t--+--+~~~--+---+--+-++~+-~+--+-+-+-++~~--+~

10 20 so 100 200 soo 1000 2000 sooo 10 000 20000

flHz l
Fig. 5.6. Frequency response requirements to meet the high fide lity standard DIN 45500.

SO .----~.----.----r~no~-r--,-,-'T'TTTI-r--~.---...,-T"T"TTTTT~-'T""'

dB i--~i--+--+++1~1-~-+--+-+++++1+-~+-+-+-+-++tt+~--+---i

401--~t--+--+.....,f-H-~--+---+-+-++++i+-~+--+-+-+-+++++-~--+----1

201--~t--.;<-,-+++11-tt-~-+--+-+++++1+-~+-+-+-++++++~--+---i

01--~t--+--+-.C...W~~-+---+-+-++~+-~+--+-~-++~~--+~

10 20 so 100 200 so o 1000 2000 sooo 10000 20000

f(Hz)
Fig. 5.7. Frequency response characteristic of sealed enclosure system employing a 12-inch
woofer, f our 5- inch mid-ra11ge speakers and four 1-inc.'i tweeters.

so .----~.----.----,..-,.-no~-,--,-,.,...,..,..,..,,-~,-...,-T"T,,.TTT-""'-i'~

ds i--~t--+-1-+-++1-tt-~-+--+-+++++1+-~+-+-+-++++++~--+---i

201~~1--.;<-,1-+-++1-tt-~-+--+-+++++1+-~+--t-+-+-+++++~--+---i

In the laboratory outside the a11echoic room th e recorder plots the freq11e11cy response of
1 01--~+-+--+.....,f-H-~--+---+--+-++++i+-~+--+-+-+-+++++~--+---i the speaker under test . The signal f ed to the speaker is swept from 20 Hz to 20000 Hz
whilst the graph paper is f ed through the recorder i11 sy nchronism. The amplitude of the
01--~1--+--+-.C...WCU..~-+---+-+-++~+-~+--+-+-+-++~~--+~
response in the microphone controls th e pen movement. When the polar response is
10 20 so 100 200 soo 1000 2000 sooo 10000 20000 required, the speak er is rotated in front of the microp hone and a polar plotter is used,
f(Hzl
the test frequency remaining consta11t.
Fig. 5.8. Frequency response characteristic of Fig. 5.7 with th e requirements of DIN 45500
superimposed. Th e loudspeaker system clearly meets the highfidelity sta11dard.

190 191
Appendix
for low-power tran sistors using a mounting clip and heat-sink:
Heat-sink Design and Calculations
Up to a certain point the junction temperature of a transistor rises ~
J R1hj -c c Rthc -h h R1hh ·a a
linearly as a function of the power di ssipated. The junction temperature
T 1 is given by: ·
(A-1) for high-power transi stors:

where Ta,,,b is the ambient temperature, R,h i-• is the thermal resistance
~
J R th 1-m b m R th mb · h R t h h- a a
between the junction and its surrounding air, and Pro, is the total power
dissipated.
The maximum junction temperature is usually given by the transistor
manufacturer, the ambient temperature for which the equipment is Fig. A .1. Equivalent thermal circuits for transistors.
intended is known by the designer, and the power dissipation can be
calcu lated for the worst case of operating cond ition s.
The worst case dissipation for output transistors in a high fidelity Class-B
configuration is given by:
In the case of small , low-power transistors no electrical connections a re
1.21 v 2
made to the case and the problems of electrical insulation do not arise.
Pr or = n2(0.8 RL + RE)' (A-2) Hence a small cooling clip is normally employed which may be directly
connected to the case of the transistor. If the heat transfer of the cooling
where V is the total d.c. voltage across the transistor and emitter re-
clip is insufficient, the clip may be screwed to a heat-sink. For efficient
sistor (in complementary and quasi-complementary output circu its this
heat transfer between the clip and the heat-sink it is very important
is taken to be the mid-point voltage) , RL is the external load impedance
that the correct force is applied to the screw in order than proper con-
and RE is the emitter resistance . Equation (A-2) corresponds to a set of
tact is maintained. The torque on the screw is therefore normally spe-
conditions in which the supply voltage is IO % higher than nominal ,
cified by the manufacturer of the cooling accessories. Heat transfer
an unfavourable sine-wave excursion, and a load impedance 20 % below
takes place by conduction and radiation , the different thermal resistances
the nominal value.
shown in Fig. A . I take both forms of heat loss into account.
In order that the junction temperature shall not be exceeded it is
High-power transistors in complementary and quasi-complementary
necessary to calculate the thermal resistance between the junction and
output stages have to be electrically insulated, since the collector is
its surrounding air. Equation (A- I) can therefore be re-arranged as
normally connected to the case. Hence it is usual to place a mica washer
follows:
between the transistor mounting base and the heat-sink . In addition ,
insulating bushes are used for the mounting screws. Heat transfer
R,h i-• (A-3)
P,o, first takes place from the mounting base, through the mica washer,
to the heat-sink as shown by R,h mb - h• and then from the heat-sink to
The total thermal resistance between the junction and the ambient
the ambient air.
air can be given an equivalent circuit as shown in Fig. A.I.

193
192
Since a wide range of accessories for heat dissipation to suit all the shows how the envelope influences the thermal resistance, because with
transistors listed in thi s book is readily avai lable, with precise data on flat heat-sinks the total cooling area becomes larger.
their therm al resistances, the remaining problem is tha t of calcu lating Fig. A .3 gives the heat-sink curves, which apply to aluminium heat-
the sizeoftheheat-sink, if any, req uired. Since the values of R,hj -m b and sinks on ly. An example shou ld make the use of them clear. Consider the
R,h mb - h a re known , the value of the unknown quantity Ru, 1t - a is given BD 18 1 output transistors of the 15 W amplifier described in Section 3.6.
by:

R,,, h -a = Ru, j- a - (R,11 j - mb + Ru, mb - h). (A-4)

The therm al resistance R, 11 h -a depends on the power dissipation, the

~ length of extruded hea t -si nk !cm) --~
surface conditions, the mounting position and, in the case of flat heat- 1 10 100
sinks , the thickness of the heat-si n k and the type of envelope of the I I I I II I I I I
I 11 II type of heat-sink ! pow er dissipation I I
transi stor. The d imensions and type of hea t-sink may be determined ~thickness of flat aluminium
heat -sin k (mm) -0: ~ --- 2ow
10w

~ ~ --
by graph ical means as shown in Fig. A.2. This graph is bui lt up of four
~ ~- -- ->--
_,_
-- ,.._ - - - .... ~ iw
- ,w
sw

length of extrusion (cm) .,,

.,
se ction m section II
... ........ 1

~-
~ 56290 'l.01:1 I
-- ---- extruded . . . . . .- ~ .... """-':2 ~ ~/ I
heat-sinks
r--- ........ ,.....,_ 3
~~ l
"'
I
I I
I
j ~ ~
I I ~o I
I ...__ I

j
c I
I
type of
heat- sink
power I
dissipation I rooo I
1

I type of or ientation and type of envelope ~ or'"ientat ion ?nd

1
t envelope surface finish su r face finish I I
'C
1 I

-f
1
ii= I I fi/
~
1; c I I ~/
~
.,
~
~
]
I
I / I ~
? bright . her. - /,f,
...
0

E s ect ion nz:: section I a:

~
0
ii=
...
0 100
I
I / bright . vert. -
blackened her.
blackened vert .
~ 1
0
.,
~

I In'
I I ~
Fig. A .2. Showing how th e heat-sink curves of Fig. A.3. are used. If the type of heat- I /, I :,?/
sink is an extrusion, th e line should be continued vertically up wards from point A in I /, !IC -- ro3 )'""/
Section Ill to give the length of the extrusion. I/; f9'. ..... {T01
T039
T066
,i
~J
T0126
)}
sections. Section I shows the dependence of the thermal resistance on the 1 111 1 1

orientat ion and surface finish of the heat-sink. S.ection 11 shows the
influence of the power dissipation under conditions of free convection Fig. A.3. H eat-sink design cur ves for audio power transistors. The dotted line shows an
example in which the required R,h h - a is 20.75 °C/ W, worst case dissipation is 6.5 W,
on the thermal resistance. Section III shows how the thermal resistance
and a 2 mm thick, flat, horizontal heat-sink of bright aluminium is used. For a transistor
varie as a function of the area a nd thickness of the heat-sink (for flat heat- with a T0 -3 envelope, it is shown that a heat-sink with a surface area (o ne side) of
sinks), or the le ngth (in the case of extruded heat-sinks). Section IV 12 cm 2 is required.

194 195
From equation (A-2) , the worst case dissipation is given by:
1.21 x 19 2
P1 0 1 = = 6.5 W,
n 2 (6.4 + 0.47)
since the mid-point voltage is 19 V, the external load impedance is 8 Q
and the emitter resistor is 0.47 n. INDEX
Assuming that the amplifier is designed for a maximum temperature
of 50 °C, then for a maximum junction temperature of 200 °C for a Acoustic power . . . . . . . . . 175
Adjustment of protection networks . 108, 127
BDl81 (from Table 3.3), the value of R 11, j - ao from equation (A-3),
Amplifier
becomes: 15
-, basic voltage .
200 - 50 150 - , buffer. . . . . 22, 52
R,,, J-• = - - - - = -- = 23 ° C/ W. - , low distortion 21
6.5 6.5 - , microphone . 24
- , mixer. . 26
From Table 3.3, the R 11,j - mb for the BDI 81is1.5 °C/W. The recommended 55
- , presence . .
mounting accessory Cat. No. 56201e, which includes a lead and a
mica washer, has a R, 1, mb - h of 0.75 °CjW. From equation (A-4), the Balance control 45
value of R,h h-a becomes: Balance meter . 47
Basic voltage amplifier 15
181
R,h h-a = 23 - (I .5 + 0.75) = 20.75 °CjW.
Bass-reflex enclosures.
Battery
equipment . . . 66
Entering Fig. A.3 at this value of R,,, h-a in Section I and moving horizon- internal resistance 10
tally until the curve for a bright horizontal heat-sink is reached, a vertical Buffer amplifier 22, 52
line is then drawn to intersect the 6.5 W power curve in Section II. By
Capacitors
interpolation between the 5 W and I 0 W curves a point is obtained from - , output . . . . . . . . . 36
which a horizontal line is drawn to intersect the 2 mm thick flat heat-sink - , use of . . . . . . . . . 57
curve in Section 111. From this point a vertical line is drawn downwards Complementary output stages 61
Cooling clip. 193
to intersect the T0-3 envelope curve in Section lV and moving horizon-
Cross-over
tally to the left will give a heat-sink area of 12 cm 2 • The length/ width distortion . 58, 63
ratio of the chosen dimensions should not exceed 1.25. filters . . . 175
Where the area of a flat heat-sink would be excessive, an extruded heat-
sink may be used. When the horizontal line enters Section I II and inter- Damping factor 59
Differential input stage 137
sects the appropriate heat-sink curve, the line should then be continued 189
DIN Standards for high fidelity .
vertically upwards to give the length of the extrusion. Disc equalization 27
An enlarged version of the heat-sink design curves for laboratory use Distortion
is supplied with this Application Book. -, cross-over. . . 58, 63
- , harmonic . . . . 9, 58
-, intermodulation . 9, 58
- , phase . . . . 57
-, power amplifier 57
Dynamic range . 57

196 197
 Earth in g . . . . 71 Microphone
Effective loud ness 5 ampl ifi er . . . . . . . . 24
Enclosures - , sensitivity for . . . . . . 4
- , bass-reflex 181 Mid-point voltage calculation . 94, 102, 121
-, sea led . . 182 Mixer with two in puts 26
- , resonance frequency of 183 Monitor
- , stiffness of . . . . . 185 output . . . . . . 33
Equalization characteristics tape inpu t 44
- , disc recordings 27 Mono power requirem en ts 180
- , R. l.A.A . . 28 Mu sic power rating 60
- , tape heads .. . 31, 164 Musica l instrum ents, frequency range . 8

Filters Noise
- , cross-over 175 contours 12
-, low-pass/ high-pass . 52 filter 53
- , noise . . 53 leve l . . 58
-, rumble . . 53 - , low . . . 9
- , scratch . . . 7, 53 -, therma l 9
-, smooth ing 38 Output capacitors 36
Frequency range of musical in struments 8 Output power ratings . 60
Frequency response of loudspeakers 186 Overdrive indica tor 133
Over loading 175
Gain
-, playback 167 Physiologica l volume contro l 6
-, recording . 167 Playback ga in 167
- , transistor . 10 Power
-, acoustic . 175
Ha rmonic distortion . 9, 58 bandwidth 60
Hea t-sink design . . 192 handling capacity 179
High fidel ity DIN Standards . 189 Power supply
High output vo ltage amp lifier 21 internal resistance 73
unit . . . . . . 121
Input resistance calcu lations . 166 Pre-amplifier
Intermodulation distortion . 9, 58 - , magnetic pick-up 27
- , Universa l . . 33
Loudness, effect ive . 5 Presence contro l
Loudspeaker amp lifier . . 55
acoust ic power. 175 principle . . 6
distortion . . . 169 Protection networks, adjustment 108, 127
enc losures 181
frequency response . 186 Quasi-complementary symmetry 65
overloading . . . . 175 Quiescent current stabi lization 102
positioning . . . . 180
power ha ndling capacity 179 Rated output power . . 60
resonance frequency . 182 Recording character istics 3
Low distortion amp li fier 21 Recording gain 167
Low nb ise . . . . . . 9 R .l.A .A. characteristic 3
Low-pass/ high-pass filter 52 Rumbl e fi lter . . . . 53

198 199
Scratch filter . 7, 53 Technology relating ta the products desc ribed in th!s book is shared by the following companies.
Sealed enclosures 182
Sensitivity for various inputs . 2 Germany New Zealand
Argentina
Short-circuit conditions . . . 105, 121 FAPESA l.y.C. VALVO G .m .b.H. EDAC Lid.
Melincue 2594 Valvo Haus 70-72 Kingsford Smith Street
Signal input level adjustment 43 Burchardstrasse 19 Tel. 873 159
Tel. 50-9941 / R155
Signal-to-noise ratio . 9 BUENOS AIRES Tel. (0411) 33 91 31 WELLIN GTON
Sine-wave rating . . . 2 HAMBURG 1
60 Australia Norway
Small-signal transistors II Philips Industries Ltd. Greece Electronica A / S
Smoothing filter M iniwalt Electronics Division Philips S.A. Hellenique Middelthunsgate 27
38 20. Herbert St. Elcoma Division Tel. 46 39 70
Stereo Tel. 43-2171 52. Av. Syngrou OSLO 3
pre-amplifiers ARTARMON , N.S.W. Tel. 915.311
36 ATHENS
power requirements Peru
179 Austria
CADESA
Stabilized supply voltage WI VEG Hong Kon g
. 60, 121 Prinz Eugenstrasse 32 Philips Hong Kong Ltd. Av. Abanca y 1176
Stray fields . . . . . Offs. 606-607
73 Tel. 65 16 21 Compo nents Deps.
Tel.77317
1041 WIEN St. George's Building 21 st Fi.
Tel. K-42 82 05 LIMA
Tape head equalization 31, 164 Belgium HONG KON G
M.B.L.E. Portugal
Tape playback voltage 163 Philips Portuguesa S.A.R.L.
80, rue des Deux Gares India
Thermal noise . . 9 Tel. 23 00 00 INBELEC Div. of Rua Joaquim Antonio de
1070 BRUSSELS Philips India Ltd. Aguiar 66
Thermal resistance 192 Band Box Building Tel. 68 37 21 /9
Tone control 48 Brasil 254-D, Dr. Annie Besant Road LISBOA
IBRAPE S.A. Tel. 45 3 3 86, 45 64 20, 45 29 86
Transformers 61 Worli , BOMBAY 18 (WB)
Rua Manoel Ramos Paiva 506 South Africa
Transients Tel. 93-5141 EDAC (Pty) Ltd.
SAO PAULO lndoncsia
-, amplifier . 8 P.T. Phi!ips-Ralin Electronics
South Park Lane
New Doornfontein
-, loudspea ker . 169 Canada Elcoma Division Tel. 24/670 1-2
Philips Electron Devices Djalan Gadjah Mada 18 JOHANNESBURG
11 6 Vanderhoof Ave. Tel. 44 163
Transistor Tel. 425-5161 DJAKARTA
TORONTO 17, Ontario Spain
biasing . 10 Ireland COPRESA S.A.
dissipat ion 192 Chile Philips Electrical (Ireland) Pty. Balmes 22
Philips Chilena S.A. Newstead, Clonskeagh Tel. 2 32 03 00
gain . . . 10 Av. Santa Maria 0760 Tel. 69 33 55 BARCELONA
noise . . . 12 Tel. 394001 DUBLIN 6
working point 10, 64 SANTIAGO Sweden
Italy ELCOMA A.B.
Tweeters . . . 169 Colombia Philips S.p.A. Lidingovagen 50
SADAPE S.A. Sezione Elcoma Tel. 08 /67 97 80
Calle 19. No. 5-51 Piazza IV Novembre 3 10250 STOCKHOLM 27
Volume control, physiological 6 Tel. 422-175 Tel. 69.94
BOGOTA D.E. MILANO
Switzerland
Width control . . . . 50 Denmark Japan Philips A.G.
Miniwatt A /S l.D.C.C . Ltd. Edenstrasse 20
Woofers 169 K okusai Building , 7th floor Tel. 051 /44 22 11
Emdrupvej II 5A
Worst case dissipation 120, 192 Tel. (01) 691622 Marunouchi CH-8027 ZUERICH
DK-2400 K0BENHAVN NV Tel. (213) 6751.7
TOKYO Taiwan
Finland Philips Taiwan Ltd.
Oy Philips A.B. Mexico Plastic Building, 10th Fi.
Elcoma Division Electr6nica S.A. de C .V. No. I. section 2, Nanking East Rd.
Kaivokatu 8 Varsovia No. 36 Tel. 55 97 42
Tel. IO 915 Tel. 5-33-11-80 TAIPEI
HELSINKI JO M EX ICO 6, D.F.

France Netherlands Turkey

R.T.C. Philips Nede rland N.V. Ti.irk Philips Ticaret A.S.
La Radiotechnique-Compelec Afd. Elonco EMET Department
Avenue Ledru Rollin 130 Boschdijk, VB Gilmilssuyu Cad.78-80
Tel. 797-99-30 Tel. (040) 433 33 33 Tel. 45 .32.50
PARIS 11 EINDHOVEN Beyoglil. ISTANBUL

200 201
Un ited Kingdom U nited S ta tes U rugu ay
Mulla rd Ltd . Amperex E lectro nic Corp. Lu zilec tron S.A .
Mullard Ho use Rondea u 1567, piso 5
Torrington Place E lect ro n Tubes Div. Tel. 9 43 21
Tel. 0 1-580 6633 Tel. 516 W E 1-6200 MONTEVIDEO
LONDON W. C . HICKS VILL E N .Y.
Venezuela
Sem . and Mi crocirc uits Div. C .A. Philips Venezo lana
Tel. 401-762-9000 Elcoma Departmen t
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Tel. 72.01.51
Electronic Components Div. CA RA CAS
Tel. 5 16-2 34-7000
HAUPPA GE N .Y.

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