Comprehensive Comparison of a SiC MOSFET and Si IGBT Based
Inverter
Maximilian Nitzsche1 , Christoph Cheshire2 , Manuel Fischer1 , Johannes Ruthardt1 ,
Jörg Roth-Stielow1
1
University of Stuttgart, Institute for Power Electronics and Electrical Drives (ILEA), Germany
2
Hochschule Esslingen, University of Applied Sciences, Germany
Corresponding author: Maximilian Nitzsche, nitzsche@ilea.uni-stuttgart.de
Abstract
The investment which is necessary to replace Si IGBTs with SiC MOSFETs in medium to high power
DC-AC inverters needs to be balanced carefully against the advantages SiC offers. This paper compares
a 20 kW Si IGBT inverter with a 20 kW SiC MOSFET inverter. The power semiconductor components are
operated identically in a modular half bridge module to ensure comparability. Thereby the measurement
of the switching losses is explicitly not the focus but the overall efficiency while taking volume, current
ripple, switching frequency and inductance into account. The limits of reasonable operating range shall be
evaluated and an overview on the benefits of SiC on system level will be given.
1 Introduction The insulated gate bipolar transistors (IGBT)
traditionally represent the high voltage (<2 kV)
In the scope of electro mobility developers strive and high current (<200 A) region while MOSFET
for higher efficiency whilst achieving high power technology in discrete packaging usually is
density and reducing overall weight of power considered for applications below 1 kV and 40 A.
electronic systems. This also applies to the ongoing As SiC enables higher blocking voltages whilst
electrification of passenger aircraft – either to satisfy reducing conduction losses (especially in partial
the rising power demand of ‘More Electric Aircraft’ load conditions) the use of SiC MOSFET technology
(MEA) [1], [2] or to even realize full electrification offers an alternative to conventional Si IGBTs. This
including that of the propulsion system (All Electric is especially the case when considering operating
Aircraft, AEA). Important constraints for the power voltages of around 800 V as discussed for electric
electronic systems are size, weight, efficiency, cost vehicles as well as electric aircraft [6]. Figure 1
and electromagnetic compatibility (EMC) [3]. In illustrates the anticipated area of application for
this context wide bandgap power semiconductors SiC MOSFETs in comparison to traditional silicon
(WBG) offer an alternative to conventional silicon based power transistors. The high power and high
power semiconductors [4], [5]. Academia and switching frequency region that is made accessible
industry are already focusing their research and by the SiC MOSFET is best suited for use in electric
development on silicon carbide (SiC) for high power mobility applications and therefore the motivation
and high voltage (>600 V) applications, with gallium for this paper.
nitride power semiconductors being the focus of
investigations into medium power and medium to Thus, this paper presents a comprehensive
low voltage applications [6]. As many reliability comparison of a Si IGBT and a SiC MOSFET half
aspects are still under investigation the application bridge module in regards to switching behavior as
of WBG power devices in industry still isn’t evolving well as efficiency in three phase inverter setup over
rapidly, however slowly gaining momentum. a large power and switching frequency range. The
focus is to analyze the dependency of the power
electronics applications’ boundary constraints
varying positive and negative gate bias depending
on MOSFET or IGBT technology.
Ceramic capacitors are placed close to the
switching cell so that inductive loops within the
DC link connection are compensated. As state
of the art ceramic DC link capacitors show a great
dependence of their capacitance on the applied
DC link voltage a combination of capacitors was
chosen. In this case MEGACAP capacitors as well
as CeraLink capacitors both from Epcos/TDK are
used and connected in a series/parallel network.
Fig. 1: SiC MOSFETs’ anticipated area of application in This is necessary to allow for a wide input voltage
comparison with traditional Si transistors
range of the modules whilst having approximately
constant DC link capacitance close to the switches
amongst each other. These include semiconductor and furthermore to enable high input voltages.
switching frequency, the applications output current
ripple as well as the conducting inductivity thus the
overall volume of the power electronic application.
As switching loss analysis has already been
investigated in previous publications (e.g. [7]) it
shall not be the considered the scope of this paper,
compare [8].
2 The Compared Power Electronic
Transistors
2.1 Setup and characteristics
Fig. 2: Water cooled half bridge module fitted with either
The power module used in this comparison was SiC MOSFETs or Si IGBTs and air cooled diodes
introduced in [9] and a SiC Schottky diode was
added as freewheeling diode for the use with
Table 1 lists the relevant component characteristics
IGBTs. Due to the fact that the inherent body diode
of both the compared transistors. The SiC MOSFET
exhibits a large forward voltage, the SiC Schottky
C3M0065100K from Wolfspeed and the Si IGBT
freewheeling diode is also used in combination with
FGH40N120AN from ON Semiconductor are
the SiC MOSFETs, as this reduces conduction
chosen for this comparison, as their respective
losses during deadtime and improves switching
blocking voltage and maximum continuous current
behavior.
are comparable. The listed collector to emitter
The power module (compare Fig. 2) consists of two saturation voltage of the IGBT VCE,sat is measured
active power semiconductors in combination with at IC = 40 A collector to emitter current and with a
the afore mentioned freewheeling diodes, as well gate to emitter voltage of VGE = 15 V. This would
as a low inductive gate driver circuit. Switching correspond to an ohmic resistance of 65 mΩ which
commands are sent via fiber optic wire from a is comparable to the MOSFET. The capacitances
FPGA command system and are protected against of the SiC MOSFET are around 80 % lower than
interlock by an analog deadtime implementation those of the Si IGBT.
circuit. The deadtime is chosen according to the
lowest suggested values of the datasheets. The The command system implements a three phase
gate driver voltages are generated by modular 50 Hz current control with variable switching
isolated DC/DC converters in order to enable frequency. A passive ohmic inductive load is used.
Tab. 1: Comparison of relevant component characteristics according to their respective datasheets
ON Semiconductor FGH40N120AN Wolfspeed C3M0065100K
IGBT MOSFET
Blocking voltage VCE = 1200 V VDS = 1000 V
Maximum continuous current IC = 40 A ID = 35 A
Conduction characteristic VCE,sat = 2.9 V RDS,on = 65 mΩ
Cies = 3200 pF, Ciss = 660 pF,
Input and output capacitance
Coes = 370 pF Coss = 60 pF
Rise and fall time tr = 20 ns, tf = 40 ns tr = 10 ns, tf = 8 ns
2.2 Analysis of switching performance
and comparison
To fully comprehend the impact the component
characteristics (Tab. 1) have on the achievable
performance and efficiency of these two
semiconductor technologies one must first
investigate the switching characteristics. For this
investigation drain-source as well as gate-source
voltages of the SiC MOSFET and collector-emitter
as well as gate-emitter voltages of the Si IGBT
are measured during switching transients using
PMK’s BumbleBee differential voltage probe. Both
turn-on and turn-off transient measurements are
Fig. 3: Drain-source and collector-emitter voltage
achieved interconnecting the half bridge module in
curves during turn-off
a synchronous buck converter setup with 800 V DC
link voltage.
Figure 3 shows the drain-source and collector-
emitter voltages during turn-off. It can be seen
that the slew rate of the Si IGBT transient voltage
is considerably lower than that of the SiC MOSFET.
This can be attributed to the lower electron mobility
of silicon when compared to silicon carbide as well
as larger parasitic capacitance. This larger slew rate
will drastically reduce the switching losses within
the transistor. However, it also causes far greater
parasitic ringing which will have detrimental effect
on EMC. A further consequence of larger parasitic
capacitance can also be seen in the delay between
the two voltage transients, as voltage across the Fig. 4: Gate-source and gate-emitter voltage curves
IGBT can only increase after the Miller capacitance during turn-off
has been fully discharged. This effect is also
visualized in Fig. 4 where the Miller plateau seen in
gate-emitter voltage of the Si IGBT is considerably In Fig. 5 larger slew rate and shorter transient
longer than that of the SiC MOSFET. delay are observed in the SiC MOSFET, as well
as greater parasitic ringing, whereas Fig. 6 shows
The component characteristics have a similar effect the considerably larger Miller plateau of the Si IGBT
on the turn-on transient voltages of both transistors. during turn-on.
power (<3 kW) than the Si IGBT inverter. Using
a thermal camera, the case temperatures of the
transistors are also measured. The IGBT reached
a temperature of 75 ◦C, the maximum temperature
allowed during these tests, due to the risk of thermal
damage being too great at higher temperatures.
The case of the SiC MOSFET on the other hand
only reached a temperature of 62 ◦C. Same cooling
was used in all cases. During further tests the
maximum switching frequency of the SiC inverter
was established to be at 150 kHz, three times as
high as that of the IGBT inverter.
Fig. 5: Drain-source and collector-emitter voltage
curves during turn-on
Fig. 7: Measured efficiency over output power,
Fig. 6: Gate-source and gate-emitter voltage curves fPWM = 50 kHz
during turn-on
The correlation of switching frequency and
3 Efficiency Comparison efficiency was also investigated over a large
frequency range (10 kHz-125 kHz) at a reduced
The afore introduced half bridge modules are output power Pout of 10 kW to allow for such a large
configured as a three phase six switch two level comparative range. The limiting factor being the
inverter with a DC link voltage of 800 V. The input case temperature of the Si IGBTs. Figure 8 shows
and output power of the inverter is measured using that with rising switching frequency the difference in
a power measurement device. The efficiency is efficiency between the two inverters also increases
calculated as quotient of output and input power from less than one percentage point at 10 kHz to
disregarding the power for signal electronics and almost three percentage points at 125 kHz. The
gate drive circuitry. far smaller drop off in efficiency of the SiC inverter
demonstrates that it is able to achieve switching
In Fig. 7 efficiency is plotted over output power, frequencies of up to 400 kHz at this power rating
both inverters running with a switching frequency as seen in Fig. 8. At this point it shall be noted
of 50 kHz. It can be seen that the SiC MOSFET that air wound inductors are used. This eliminates
inverter achieves a higher efficiency than the core losses from the efficiency calculation. Further
Si IGBT over the whole power range, whilst also losses (such as ohmic resistance of the bus bars)
demonstrating a smaller drop off in efficiency at low are approximated to be constant.
Fig. 8: Measured efficiency over switching frequency, Fig. 9: Measured efficiency over switching frequency
Pout = 10 kW fPWM including indication of the transistors’ case
temperature
How the differing efficiency translates to thermal
load on the transistor itself is also investigated the increased losses within the inductivity due to
using a thermal camera. Both inverter are operated increasing size are far lower than the increase in
at a load of 20 kW and are cooled identically switching losses due to higher switching frequency
by a copper block which is perfused by water required to achieve the same reduction in output
(about 15 ◦C). The results seen in Fig. 9 clearly current ripple. However, an increase in inductance
show the effects of larger switching losses as well will mean an increase in the size of the inductivities’
as inferior thermal conductivity of the Si IGBT dimensions and thus an increase in the size of
compared to its SiC counterpart. With the IGBT the whole inverter, compare the lower part of
demonstrating an increase in package temperature Fig. 10. On the other hand, increasing switching
of more than 40 ◦C over the comparative range, frequency, also delivers the desired reduction in
with the SiC MOSFET only exhibiting an increase output current ripple without increasing overall size,
of 13 ◦C. but the increased switching losses has a detrimental
effect on efficiency.
This will not only allow far greater switching
frequencies to be achieved, as already proven in Prioritizing high efficiency over size by decreasing
Fig. 8, but also allow for further optimization of switching frequency whilst increasing the size of
heatsink design as well opening up alternatives to the inductor works well with both inverters, with
the conventional water-cooled approach. the SiC MOSFET based inverter only achieving
a slightly higher efficiency of 97.7 % compared
As overall size and efficiency are important criteria with 97.0 % of the Si IGBT based inverter. When
in inverter design, the relationship between these prioritizing size however, the difference between
aspects is investigated. For this a constant output the two transistors becomes far more obvious. The
current ripple of 4 A is chosen in the following much greater achievable switching frequency of the
elaboration (depicted in the upper part of Fig. 10). SiC MOSFET means that we can decrease the size
To achieve this current ripple, one of two factors of the overall system by over 20 % compared to a
can be changed: either switching frequency or the Si IGBT based inverter.
inductance of the conducting inductivity.
Increasing the inductance will decrease output
current ripple whilst minimizing the detrimental
effects on efficiency. This is due to the fact that
voltage which accompany the SiC technology are
highlighted.
Not only the correlation of efficiency, output
power and switching frequency is taken into
account. Moreover, the package temperature of
the power electronic components is visualized and
in one example the achievable inverter size is
approximated in accordance to the size of the
necessary inductor.
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