Circuits, Systems, and Signal Processing (2022) 41:2486–2502

https://doi.org/10.1007/s00034-021-01901-8

Design of Active Inductor-Based VCO with Wide Tuning

Range for RF Front End

Lakshmi Nediyara Suresh1 · Bhaskar Manickam1

Received: 3 October 2020 / Revised: 25 October 2021 / Accepted: 31 October 2021 /

Published online: 22 November 2021
© The Author(s), under exclusive licence to Springer Science+Business Media, LLC, part of Springer Nature 2021

Abstract
An active inductor based on gyrator-C structure is proposed in this work. Voltage-
controlled oscillator with wide frequency range and significant RF output power based
on the proposed active inductor is presented in this work. The design of the voltage-
controlled oscillator is carried out using UMC 180 nm RFCMOS technology. Post-
layout simulations are carried out using Spectre RF in Cadence virtuoso. Voltage-
controlled oscillator can be tuned from 500 MHz to 2.8 GHz suitable for multi-band
transceiver design. The proposed design attains a frequency tuning range of 139.4%.
VCO operates at a supply voltage of 1.8 V with a power consumption of 5.3 mW.
Phase noise at a frequency offset of 1 MHz from carrier frequency of 2.4 GHz is − 94
dBc/Hz. VCO occupies an active area of 200 × 150 µm2 . Figure of merit values that
take into consideration RF output power and frequency tuning range are comparable
to other VCO designs proposed in the literature. PVT analysis of the designed VCO
is presented in this work.

Keywords Gyrator-C · Active inductor · Tuning range · Oscillator · Self-resonant

frequency

1 Introduction

A wideband voltage-controlled oscillator (VCO) is required for frequency translation

in reconfigurable multi-standard RF transceivers. Multi-band transceivers demand the
co-existence of several network standards like GSM, CDMA, WLAN, WPAN, which
span multiple frequency bands ranging from 800 MHz to 2.5 GHz. The synthesis of

B Lakshmi Nediyara Suresh

lachuns123@gmail.com
Bhaskar Manickam
bhaskar@nitt.edu

1 Department of Electronics and Communication Engineering, National Institute of Technology,

Tiruchirappalli, India
Circuits, Systems, and Signal Processing (2022) 41:2486–2502 2487

active inductor [3,22] and its applications is an area of active research. Active inductors
play a vital role in the design of reconfigurable RF blocks, which include filter [17,20],
phase shifter [33], low noise amplifier [12], etc.
Synthesis of feedback systems is addressed in [28,29]. VCO structure has a feedback
loop consisting of resonator and cross-coupled pair. Conventionally, passive LC VCOs
are used due to their superior phase noise performance. But they have a limited tuning
range determined by maximum to minimum capacitance ratio of varactors. This limited
tuning range restricts VCO tuning range to an approximate limit of 30% [18]. Switched
capacitors [2] and switched coupled inductors [4] are used for wide tuning range with
critical control mechanisms. Active inductor-based VCO is a promising candidate in
the design of wide tuning range VCOs [9,13,26,30], but higher power consumption
than passive implementation is a drawback. However, frequency tuning range with
active inductors exceeds 100% [15]. DC to RF power conversion efficiency based
on active inductor depends on the voltage swing of the chosen circuit topology [31].
Cascoding in active inductor topology reduces effective series resistance of active
inductor, but this will also limit the frequency range up to which the circuit behaves
inductive. In addition to that, cascoding technique limits the voltage swing of VCO,
which in turn reduces the carrier signal power.
On-chip spiral inductor is presented in [1], which deals with square spiral inductor
realized using copper, aluminum and carbon nanotube. For spiral inductors, inductance
depends on electromagnetic field, whereas for active inductors, inductance depends on
transconductance of MOSFETs. Spiral inductor attains a low-quality factor, whereas
active inductor attains a high and tunable quality factor. Q factor is less than 10 in
[1], whereas Q as high as 250 is obtained for the proposed active inductor. The induc-
tance of spiral inductor depends on the geometry and the number of turns, whereas
the inductance of active inductor is independent of geometry. Spiral inductor has
fixed inductance depending on the number of turns. Large inductance value can be
obtained by stacking several spiral inductors, this in turn increases spiral-substrate
capacitance. Otherwise, the number of turns of the spiral needs to be increased which
consumes more area. At high frequency, skin effect leads to resistive loss in the spi-
ral. This degrades the quality factor. Spiral inductors also suffer from disadvantages
like capacitive and inductive cross-coupling with the substrate. Hence, compared to a
spiral inductor, active inductor provides the advantages, such as tunable inductance,
large and tunable quality factor, tunable self-resonant frequency (maximum frequency
up to which input impedance is inductive) and consumes small area. Hence, in low
power applications spiral inductors are attractive than active inductors, whereas for
reconfigurable RF transceivers that need to be operated on multi-standard and multiple
frequency bands, an active inductor-based design block is a suitable design choice.
A gyrator loaded by capacitor (intrinsic parasitic capacitance of transconductor in
gyrator network) emulates inductive impedance. Positive transconductor and nega-
tive transconductor connected in negative feedback form a gyrator. The positive and
negative transconductors can be realized using either n-type or p-type transconductor
stages [24]. Common gate or common drain configurations can be used to realize pos-
itive transconductor, and common source configuration can be used to realize negative
transconductor. Depending on the circuit topology, voltage swing of active inductor is
limited since the transconductors in gyrator network have to be driven into saturation.
2488 Circuits, Systems, and Signal Processing (2022) 41:2486–2502

Class AB active inductor with enhanced voltage swing is presented in [25,32]. The
input admittance of active inductor emulates input admittance of the shunt RLC tank
circuit.
The purpose of the study is to design an active inductor topology that attains a good
signal handling capability and wide inductance tuning range, which can be utilized in
the design of VCO. In the proposed circuit, instead of depending on cascode active
inductor topology, a tunable resistor is added in the feedback path of active inductor
topology with enhanced signal handling capability. This adds design freedom to tune
the self-resonant frequency of active inductor. VCO is designed with the proposed
active inductor as the pseudo-differential tank circuit, and the oscillations of VCO
are initiated and sustained by CMOS cross-coupled pair. The proposed CMOS cross-
coupled pair VCO attains large carrier signal power and moderate phase noise for a
wide tuning range. This leads to an efficient VCO design for multi-band transceivers.
Two limitations of active inductor are limited input signal handling capability which
depends on circuit’s topology and design freedom in tuning self-resonant frequency.
These two factors affect performance of VCO designs utilizing active inductors. The
major contributions of this research work are listed below.

1. A reconfigurable active inductor is proposed in this work. Two inductors are con-
nected in parallel either of which operates in an interleaved manner for low and high
input voltage range. Tuning range is enhanced by adding reconfigurable resistor
in feedback path of each active inductor. Compared to the active inductor design
proposed in [25], an additional design freedom is added by adding a reconfigurable
resistor. For active inductor proposed in [25] inductance and tuning range is deter-
mined by gm of transistors, which in turn depends on biasing. In the proposed
design, varying the feedback resistance using control voltage of the transistor con-
nected across it, both inductance and tuning range can be tuned without limiting the
signal swing. Q of the active inductor can also be increased by adding resistor in
feedback path. Hence, an additional design freedom is added to an existing active
inductor design that resulted in an active inductor design with high Q and signal
handling capability.
2. The proposed active inductor is used in the design of the proposed active inductor-
based VCO. High Q active inductor is utilized to minimize the resonator losses as
low as possible. The resistive losses are compensated by the CMOS cross-coupled
pair.
3. The proposed active inductor-based VCO attains a frequency tuning range ( FTR)
of 139.4% and output power of 7.54 dBm, which is a significant output power to
drive successive blocks of RF transceiver like power amplifier without drivers.
4. Three figures of merits (FOM) are used for comparing the performance of the
proposed active inductor-based VCO with the state-of-the-art VCO designs. Con-
sidering the frequency tuning range, output power and phase noise, the overall
performance of the VCO utilizing the proposed active inductor is on par with state-
of-the-art designs. FOM2 of the proposed design considering the output power is
the best compared to existing designs.
5. Active inductor proposed in this work is a promising candidate in VCO design for
reconfigurable RF front end operating in 2.4 GHz ISM band and its application can
Circuits, Systems, and Signal Processing (2022) 41:2486–2502 2489

Fig. 1 Basic gyrator-C-based active inductors: a CG -CS (n − p), b CG-CS ( p − n), c Equivalent RLC
network

be extended to reconfigurable bandpass filter design, since RLC equivalent network

of the active inductor is a shunt RLC circuit.

This paper is organized as follows. Section 2 presents the details regarding the
topology of the proposed active inductor for VCO design. Section 3 describes the
proposed active inductor-based VCO design based on the proposed active inductor.
Section 4 consolidates the post-layout simulation results of VCO. The performance
metrics are compared with the performance of other VCO designs reported in the
literature. Section 5 concludes the work.

Notations used CG-common gate, CS-common source. n-p-N type P type, p − n − P

type N type. gmi -transconductance of transistor i, Cgsi -gate-source capacitance of tran-
sistor i. R p -shunt resistance,C p -shunt capacitance, L-Inductance, Rs -series resistance.
Vin -threshold voltage of NMOS, Vthn -threshold voltage of NMOS, Vthp - threshold
voltage of PMOS, Vth -threshold voltage of MOSFET, Vdsat -drain-source saturation
voltage of MOSFET, Vdd -supply voltage, Vc -control voltage of transistor connected
in shunt with resistor. ωo -self-resonant frequency of active inductor, Itail -tail current.

2 Proposed Active Inductor

2.1 Basic Model of Gyrator-C-Based Active Inductor

CG-CS (n − p) and CG-CS ( p − n) basic gyrator-C-based active inductors [24]

are shown in Fig. 1a, b, respectively. M1 is the positive transconductor and M2 is the
negative transconductor in these circuits. Both circuits have the same input admittance
corresponding to the input admittance of the RLC network shown in Fig. 1c . The input
admittance in both these cases is given below

gm1 gm2
Yin = sC gs1 + gm1 + (1)
sC gs2 + go1

Equivalent RLC network parameters of the given active inductors are as in (2)–(5)
2490 Circuits, Systems, and Signal Processing (2022) 41:2486–2502

1
Rp = (2)
gm1
C p = C gs1 (3)
C gs2
L= (4)
gm1 gm2
go1
Rs = (5)
gm1 gm2

where gm1 is transconductance of the positive transconductor and gm2 is transcon-

ductance of the negative transconductor. Cgsi is the gate-source capacitance of
transconductor i. goi is output conductance of the transconductor i. These circuits
have limited input signal swing.

2.2 Modeling and Operation of the Proposed Active Inductor

The active inductor shown in Fig. 2a and presented in [25] is a combination of active
inductors in Fig. 1a, b and attains it rail to rail voltage swing. This circuit has enhanced
signal handling capability compared to Fig. 1a, b. Let Vthn = | Vth p | = Vth and Vdsat
be the drain source saturation voltage of current sources. For Vin ≤ Vdd - (Vdsat +Vth
), gyrator-C network formed by M1 and M2 forms an RLC resonating network. For
Vin ≥ (Vdsat -Vth ), gyrator-C network formed by M3 and M4 forms an RLC resonating
network. Hence, active inductors formed by M1 and M2 , and M3 and M4 operate in an
interleaved manner. For (Vdsat -Vth ) ≤ Vin ≤ Vdd - (Vdsat +Vth ), both M3 and M4 are in
saturation. The input impedance of this circuit is the parallel combination of impedance
of the two active inductors. Transistors are sized, and bias currents I1 and I2 are set
such that the inductance is same for both the cases ( C gs2 /gm1 gm2 = C gs4 /gm3 gm4 ) , to
have a symmetrical voltage swing [25]. The proposed active inductor shown in Fig. 2b
attains enhanced tunability compared to active inductor in Fig. 2a, since reconfigurable
resistors R1 , R2 are added in the feedback path of each gyrator network. The addition
of resistor in feedback path of active inductor reduces series resistance and increases
inductance [8]. The design freedom can be enhanced by making it tunable [19]. Since
inductance changes with feedback resistance, resistor values R1 and R2 are chosen to
be of the same value in order to get the same inductance for inductor formed by M1
and M2 , and M3 and M4 . The feedback resistance is made tunable using transistors
(M5 , M6 ) controlled by Vc in shunt with the resistor. The active inductor with rail
to rail voltage swing is chosen since the limited voltage swing of the active inductor
limits the swing of VCO. As control voltage varies, feedback resistance and inductance
varies, which in turn helps to tune the self-resonant frequency of active inductor. Thus
a reconfigurable wide tuning range active inductor that operates over a large input
voltage range can be achieved. Hence, the proposed topology is a suitable candidate
for reconfigurable VCO design with large signal swing. The schematic circuit of
generalized gyrator-C network with resistor R in feedback shown in Fig. 3 is used
to determine the RLC equivalent circuit elements of proposed active inductor circuit
given in Fig. 2b. Corresponding to the schematic, two active inductors operate in an
interleaved manner. When the input voltage Vin is high, M1 and M2 with feedback
Circuits, Systems, and Signal Processing (2022) 41:2486–2502 2491

Fig. 2 Active inductor topology: a Active inductor with enhanced swing, b Proposed active inductor

Fig. 3 Schematic of gyrator-C network with feedback resistor

resistor R1 forms the gyrator-C network. When the input voltage Vin is low, M3 and
M4 with feedback resistor R2 forms gyrator-C network. The input admittance of
generalized gyrator C network with feedback resistor is given in (6)

G m1 G m2
Y = G o2 + sC2 +   (6)
 
sCi2 R + 1
sCi2 G o1 + Co1 +  1 
R+ sC1
i2

Equivalent RLC network parameters of the proposed active inductor circuit can be
modeled from (6) and are given in (7)–(10).

Co1 + Ci2 (1 + G o1 R)
L= (7)
G m1 G m2
G o1 − ω2 Ci2 Co1 R
Rs = (8)
G m1 G m2
2492 Circuits, Systems, and Signal Processing (2022) 41:2486–2502

Fig. 4 Inductor characteristics: a Variation of inductance of active inductor with control voltage, b Variation
of quality factor of active inductor with control voltage

1
Rp = (9)
G o2
C p = C2 (10)

Here, G m1 is transconductance of positive transconductor and G m2 is transconductance

of negative transconductor. G o2 is total conductance at node 2. C2 is total capacitance
at node 2. G o1 is total conductance at node 1. Ci2 is input capacitance of negative
transconductor. Co1 is output capacitance of positive transconductor. When Vin is high,
the parameters R p , C p , L and Rs are determined by M1 (positive transconductor), M2
(negative transconductor) and feedback resistor R1 . On the other hand, when Vin is
low, R p , C p , L and Rs are determined by M3 (positive transconductor ), M4 (negative
transconductor) and feedback resistor R2 .
Frequency up-to which circuit shows inductive behavior is the self-resonant fre-
quency of active inductor. The self-resonant frequency is given in (11)

1
ωo = (11)
LC p

Figure 4a shows variation of inductance with control voltage. Inductance depends

on the imaginary part of input impedance of the active inductor. As control voltage
increases, effective feedback resistance decreases and inductance decreases, which, in
turn, increases the self-resonant frequency as shown in Fig. 4a. At a control voltage
of 1.8 V with feedback resistor R1 =R2 =1KΩ, self-resonant frequency of 3.5 GHz
is obtained. Figure 4b shows quality factor (Q) of the designed active inductor. Q
factor depends on the ratio of imaginary part to real part of input impedance. The
proposed design attains a maximum Q (Q max )of 250. The performance parameters
of the proposed active inductor is compared with active inductor designs reported in
[5,25] and is given in Table 1. Active inductor with the same signal handling capability
is presented in [25]. The proposed active inductor has higher Q and self-resonant
frequency than that presented in [25]. Reconfigurable resistor adds additional design
freedom to tune the self-resonant frequency and quality factor, whereas it was limited
to the transconductance design parameter in [25]. [5] presents a digitally tunable active
inductor where the self-resonant frequency can be tuned by varying the digital control
Circuits, Systems, and Signal Processing (2022) 41:2486–2502 2493

Table 1 Performance
Parameter [25] [5] This
comparison of active inductor
work
L max 1 (nH) 51.5 90 30
Q max 2 126 60 250
Self-resonant frequency (GHz) 2.6 3 3.5
Pdc 3 (mW) 2.7 3.5 2.5

L max 1 -Maximum inductance,Q max 3 -Maximum quality factor, Pdc

3 -DC power consumption

Fig. 5 Prototype of VCO

world at the cost of more power consumption. Q of the proposed active inductor is
higher than that of the active inductor design presented in [5].

3 VCO Based on Active Inductor

Figure 5 shows the prototype of the proposed active inductor-based voltage-controlled

oscillator. Tank circuit of the VCO is designed using the proposed active inductor. A
pseudodifferential active inductor formed by two single ended, identical grounded
active inductors is used to realize the RLC tank circuit of VCO. Cross-coupled neg-
ative transconductance stage(G m stage) is required to start and sustain oscillations.
CMOS cross-coupled pair (Mna -Mnb ,Mpa -Mpb ) is used to realize negative transcon-
ductance stage since it provides twice the voltage swing compared to NMOS only
structure [7] for the same power consumption. Cross-coupled pair is designed to pro-
vide large gain to initiate oscillation, and negative conductance of this stage should
compensate for the resistive losses of tank circuit to sustain oscillation. Itail deter-
mines the transconductance of negative transconductance stage. As a rule of thumb,
in order to sustain oscillation, negative resistance of the CMOS cross-coupled pair
(Rneg ) should be three times the largest parallel resistance of the tank circuit [31].
Hence, Rneg =-2/G m where transconductance G m is sum of the transconductance of
NMOS (gmn =gmna +gmnb ) and PMOS ( gmp =gmpa +gmpb ) cross- coupled pairs.
2494 Circuits, Systems, and Signal Processing (2022) 41:2486–2502

Fig. 6 Proposed VCO

Fig. 7 Layout of VCO

Figure 6 shows the proposed CMOS cross-coupled pair configured active inductor-
based VCO. M1a –M8a and M1b –M8b constitute the active inductor that determines the
frequency of oscillation. Frequency of oscillation can be tuned depending on control
voltage Vc of M5a , M6a , M5b and M6b transistors in shunt with feedback resistors R1a ,
R2a , R1b and R2b respectively. M7a , M8a , M7b and M8b provide bias current for active
inductor. CMOS cross-coupled pair (M9a , M9b and M10a , M10b ) realizes negative
transconductance stage. M11 provides tail current to cross-coupled pair. Transistors of
cross-coupled pair are appropriately sized to have transconductance sufficient enough
to start-up oscillation. The differential output of VCO is given by Vout+ -Vout− .
Layout is shown in Fig. 7. The three performance parameters taken into consider-
ation for the design of VCO are tuning range, phase noise and output power.
Circuits, Systems, and Signal Processing (2022) 41:2486–2502 2495

3.1 Tuning Range

Tuning range of VCO depends on active inductor. Resonant frequency of VCO (ω) is
given below

1
ω= (12)
LCtot

Here, the total capacitance Ctot = C p + C x ; where C p is shunt capacitance of equiva-

lent RLC circuit of active inductor; C x is the parasitic capacitance of the cross-coupled
pair. Inductance (L) can be tuned by varying the control voltage Vc . As the control
voltage increases, frequency of oscillation increases.

3.2 Phase Noise

The single side band phase noise spectral density in dBc/Hz due to noise current source
at a frequency offset of Δω from the carrier frequency is given by Hajmeere-lee model
[6], as given below
⎛ ⎞
i n2 Δf
⎜ 2 ⎟
Γrms
L(Δω) = 10log ⎜
⎝ . ⎟ (13)
2
qmax 2Δω2 ⎠

Here, L(Δω) is the phase noise at frequency offset Δω from carrier frequency ω;
i n2 Δf is the power spectral density of current noise source; qmax is the maximum
charge swing at output node; Γrms is the root mean square value of impulse sensi-
tivity function associated with the noise source. i n2 Δf is superposition of noise

2
power spectral density of active inductor (i nAI Δf ) and CMOS cross-coupled pair

2
(i nGm Δf ) as given below

i n2 Δf = i nAI
2
Δf + i nGm
2
Δf (14)

Noise associated with the active inductor consists of thermal noise due to resis-
tors (4KT/R) and transistors. Noise associated with each transistor consists of two
components—channel-induced drain current noise and gate-induced noise due to the
rapid fluctuation of channel charge as given in (15).

2
i nM Δf = 4KTδgds + 4KTγ gm (15)
2496 Circuits, Systems, and Signal Processing (2022) 41:2486–2502

Fig. 8 Performance of VCO : a Transient response, b Frequency of oscillation and DC power consumption
over the tuning range, c Phase noise at 1 MHz offset from 2.4 GHz, d Output power and phase noise over
tuning range

where gds and gm are output conductance and transconductance, respectively. K -

Boltzmann constant, T -Absolute temperature, drain current noise coefficient γ = 2
and gate noise coefficient δ = 4 for short channel devices [15].

3.3 Output Power

CMOS cross-coupled pair compensates for resistive losses of tank circuit and pro-
vides high output power. The proposed VCO design attains reasonably good voltage
swing since the resonator network formed by active inductor has good signal handling
capability. Hence, most of the DC input power is converted to RF power, leading to
high efficiency.

4 Results and Discussion

VCO is designed using UMC 180 nm 6-metal 1-poly RFCMOS technology. The circuit
occupies an active area of 200×150µm2 . Post-layout simulations are carried out using
Cadence Spectre RF, with BSIM 3.3 device model. Post-layout simulation results are
given in Fig. 8. Transient response is shown in Fig. 8a. Figure 8b shows the tuning
range of VCO and DC power consumption over this range. As the control voltage Vc
Circuits, Systems, and Signal Processing (2022) 41:2486–2502 2497

Fig. 9 Corner analysis: a Variation in frequency of oscillation with process corners, b Variation in phase
noise with process corners, c Frequency tuning characteristics for various process corners

is varied from 0 to − 1.8 V; frequency of oscillation increases from 500 MHz to 2.8
GHz resulting in a frequency tuning range of 139.4%. DC power consumption varies
from 4.3 to 5.3 mW over the tuning range. At a control voltage of 1.2 V frequency
of oscillation is 2.4 GHz. Figure 8c shows the phase noise at 1 MHz offset from
the carrier frequency 2.4 GHz. Periodic steady state (PSS) and Pnoise analysis are
performed to obtain the phase noise of VCO at 1 MHz frequency offset from 2.4 GHz,
and it is observed to be − 94 dBc/Hz. Figure 8d shows variation of phase noise and
output power over the entire tuning range. Phase noise decreases from − 90 to − 97
dBc/Hz with increase in signal power. This is in accordance with the theory that phase
noise improves with increase in carrier signal power [14]. Output signal power in dBm
shown in Fig. 8d is obtained from PSS analysis. Output power is significantly high for
the proposed oscillator design and it is found to be 7.54 dBm at 1.2 V when carrier
frequency is 2.4 GHz.
Figure 9 shows the variation in frequency of oscillation and phase noise with differ-
ent process corners of NMOS and PMOS transistors- slow–slow (SS), slow–fast (SF),
typical (TYP), fast–slow (FS) and fast–fast (FF). Frequency varies from 2.2 to 2.8
GHz as given in Fig. 9a. Phase noise at 1 MHz offset from carrier frequency 2.4 GHz
is found to vary from − 92 to − 95.5 dBc/Hz with process corners as shown in Fig. 9b.
Frequency tuning characteristics for different process corners are shown in Fig. 9c.
Variation of phase noise with temperature and supply voltage is shown in Fig. 10a,
2498 Circuits, Systems, and Signal Processing (2022) 41:2486–2502

Fig. 10 Phase noise variation with temperature and supply voltage: a Phase noise variation with temperature
at 1 MHz offset from carrier frequency 2.4 GHz, b Phase noise variation with supply voltage at 1 MHz
offset from carrier frequency 2.4 GHz

b, respectively. The PVT analysis shows that the circuit is robust with variation in
process, voltage, and temperature.
The designed VCO is compared with state-of-the-art VCO designs reported in
references and is given in Table 2. Three figures of merits are used for performance
comparison.
FOM1 is given in (16), based on power consumption and phase noise at an offset
Δ ω from frequency of oscillation ω [27].

 ω   
Pdc
FOM1 = L(Δω) − 20log + 10log (16)
Δω 1mW

FOM2 given in (17) considers efficiency of VCO, taking output power Pout into
consideration [21]
 
Pout
FOM2 = FoM1 − 10log (17)
1mW

FOM3 given in (18) considers frequency tuning range of VCO [27]

   
ω FTR Pdc
FOM3 = L(Δω) − 20log + 10log (18)
Δω 10 1mW

Here L(Δω) is the phase noise of VCO, FTR is the frequency tuning range in %,
Pout is the output power in dBm, Pdc is the DC power consumption in mW.
Comparison of the proposed active inductor-based VCO with LC active and LC
passive VCO is given in Table 2. LC passive VCO possess superior phase noise perfor-
mance. LC active VCO provides wider tuning range compared to LC passive VCO. It
is observed that, the proposed active inductor-based VCO based on the reconfigurable
active inductor with enhanced tunability and significant signal handling capability
resulted in a VCO design with wide tuning range, high output power, and moderate
Table 2 Performance comparison of CMOS voltage-controlled oscillator

Parameter [15] [31] [21] [16]a [11] [10] [23]a This worka

Technology (nm) 180 130 180 45 180 180 90 180

VCO Topology LC active LC active LC active LC active LC active LC active LC passive LC active
Tuning range(MHz) 500–3000 833–3720 954.3–1009.4 1100–1800 834– 997 500–3600 6400–10000 500–2800
ω (MHz) 2900 1260 978 1800 900 1480 8100 2400
Phase noise (dBc/Hz) − 102 − 104 − 109 − 98.37 − 127.8 − 100 − 102 − 94
Pdc (mW) 28 13 19 1.1 37.4 24.3 3.12 5.3
Circuits, Systems, and Signal Processing (2022) 41:2486–2502

FTR (%) 142.8 127 5.6 48.27 17.44 151.2 44.8 139.4
Output power(dBm) − 20 − 0.9 2.54 – − 21 − 19.89 – 7.54
FOM1 (dBc/Hz) − 156.8 − 154.9 − 156 − 163.1 − 171.15 − 149.5 − 175.3 − 154.4
FOM2 (dBc/Hz) − 136.8 − 153.96 − 158.6 − − 150.2 − 131.7 – − 161.9
FOM3 (dBc/Hz) − 179.9 − 176.9 − 151 − 176.7 − 176.2 − 172.4 − 188.4 − 177.3
a Post-layout simulation result
2499
2500 Circuits, Systems, and Signal Processing (2022) 41:2486–2502

phase noise at low power consumption. There exists a trade-off in attaining maxi-
mum frequency of oscillation and low phase noise. The resonator circuit can handle
large input voltage swing since two active inductors operate in an interleaved manner
depending on input signal swing. Hence, the active inductor design proposed in this
research work can be utilized in VCO design in RF front end. FOM1 that takes into
account phase noise and power consumption is − 154.4 dBc/Hz which is comparable
to existing to LC active VCO designs. It is observed from the performance comparison
table that, FOM2 which considers output power is the best for the proposed design
(− 161.9 dBc/Hz). FOM3 that considers tuning range is − 177.3 dBc/Hz for the pro-
posed VCO is a good figure of merit compared to other active LC VCO deigns. The
three figures of merits prove that the proposed active inductor-based VCO is on par
with other active LC VCO designs reported in the literature. Pros and cons of the work
are listed below.
Pros of the work
1. The proposed reconfigurable active inductor adds an additional design freedom
to an active inductor design with large signal handling capability. High Q active
inductor with wide tuning range is designed that can be utilized in the resonator of
VCO. Lesser the Q of the resonator due to resistive losses, more power is need to
boost the transconductance of the cross-coupled pair to compensate for the losses.
Hence, high Q resonator reduces the power requirement of the proposed active
inductor-based VCO.
2. The proposed active inductor-based VCO provides wide tuning range and high
output power, which leads to high RF efficiency. Due to an active resonator, signal
strength is high enough to drive the successive block in RF front end like mixer,
and drivers for power amplifier can be avoided.
3. For reconfigurable RF front end that needs to be operated on wide frequency band,
the proposed active inductor-based VCO is a suitable candidate compared to passive
LC VCO, since the tuning range is limited for passive LC VCO. To increase the
tuning range, switch capacitor arrays or capacitor banks need to be added which
consumes more area and increases the power consumption. Otherwise, switched
coupled inductors need to be used for increasing the tuning range, which in turn
consumes area.
Cons of the work
1. As the VCO design is an active LC VCO, phase noise is moderate. Passive LC VCO
has superior phase noise performance compared to active LC VCO in general.

5 Conclusion

In this paper, an active inductor with good input signal handling capability and wide
tuning range is proposed. A voltage-controlled oscillator based on active inductor is
designed for wideband transceiver architecture. VCO based on the proposed inductor
operates in wide frequency range with a frequency tuning range of 139.4%. VCO pro-
vides significant output power that is capable of driving other blocks like mixer. FOM
based on tuning range and output power efficiency is comparable to other state-of-the-
Circuits, Systems, and Signal Processing (2022) 41:2486–2502 2501

art designs in the literature. VCO occupies an active area (200 × 150 µm2 ) suitable
for compact RF front end. It can be used for applications that require reconfigurable
RF front ends, such as Internet of Things. The study is limited to the circuit design
of active inductor considering the signal handling capability and tuning range of the
active inductor and its application in VCO design. Phase noise reduction techniques
can be applied to the proposed design as a future work. As a future scope, the appli-
cation of the proposed active inductor can be extended to the realization of bandpass
filter, impedance matching load in low noise amplifier(LNA) design and phase shifter.
Funding There is no funding associated with this work.

Data Availability This manuscript has no associated data.

Declarations

Conflict of interest The authors declare that there are no conflicts of interest to disclose.

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