applied

sciences
Article
A Four-Port Dual-Band Dual-Polarized Antenna for Ku-Band
Satellite Communications
Son Trinh-Van1 , Woo Yong Yang 2 , Hyung Won Cho 2 and Keum Cheol Hwang 1, *

1 Department of Electrical and Computer Engineering, Sungkyunkwan University,

Suwon 440-746, Republic of Korea; jsonbkhn@gmail.com
2 The UK Office (Hanwha Phasor), Hanwha System, Seoul 04541, Republic of Korea;
eric.yang@hanwha-phasor.com (W.Y.Y.); hyungweon.cho@hanwhasystems.com (H.W.C.)
* Correspondence: khwang@skku.edu; Tel.: +82-31-290-7978

Abstract: This paper presents the development of a four-port dual-band dual-polarized antenna for
transmitting/receiving (Tx/Rx) applications in Ku-band satellite communications. The antenna con-
sists of two antenna elements sharing a common radiating aperture, a low-band element formed by an
L-probe proximity-fed square-ring radiator for operation at the Rx band of 10.7–12.75 GHz, and a high-
band element realized by a stacked-patch radiator for operation at the Tx band of 13.75–14.5 GHz.
Within a compact multilayer structure, the antenna achieves wide dual-band operation, with each
band having the ability to simultaneously operate with dual polarization. An antenna prototype is
fabricated and tested to verify its performance. The experimental results show that the proposed
antenna achieves an impedance bandwidth of 9.28–12.96 GHz (13.21–15.32 GHz), an isolation value
between two orthogonal polarizations of 12 dB (12.4 dB), and a peak gain of 6.63 dBi (5.42 diBi) at
the low band (high band). Tx/Rx isolation of more than 14 dB is achieved in both the Rx band and
Tx band.

Keywords: dual-band antenna; dual-polarized; Ku-band; satellite communications; shared aperture

1. Introduction
Citation: Trinh-Van, S.; Yang, W.Y.; Recently, there has been an increase in the demand for high-speed internet connectivity
Cho, H.W.; Hwang, K.C. A Four-Port across large coverage areas, both in the defense and commercial sectors [1–5]. Satellite
Dual-Band Dual-Polarized Antenna communication (SATCOM) Internet is one of the most effective solutions to meet such
for Ku-Band Satellite Communications. demands, providing communication and broadcast services to thousands of terminals
Appl. Sci. 2024, 14, 2730. https:// across a wide geographical area. This potential market is attracting commercial companies
doi.org/10.3390/app14072730 like Starlink, OneWeb, and Kuiper, leading to a new race to launch space-based broadband
Academic Editor: Nuno Silva
connectivity with SATCOM systems. In general, to ensure an optimal link and high
data rate between the satellite and the user, a mechanically scanned or phased-array
Received: 26 December 2023 user terminal is required to track low-Earth-orbit (LEO) satellites (for fixed and mobile
Revised: 15 January 2024 users) and geostationary equatorial orbit (GEO) (for mobile users) or medium-Earth-orbit
Accepted: 5 March 2024 (MEO) satellites.
Published: 25 March 2024
Traditionally, mechanically steered parabolic dishes or waveguide slot arrays have
been used for SATCOM antennas due to their low cost and high efficiency [6]. However,
these antennas are bulky and have a large swept volume. Although fixed-beam antenna
Copyright: © 2024 by the authors.
arrays can offer a low-profile solution [7], they still require mechanical scanning both in
Licensee MDPI, Basel, Switzerland. the azimuth and elevation planes, so they consume a significant volume. A low profile is
This article is an open access article also achieved by combining electronic steering in elevation with mechanical beam steering
distributed under the terms and in azimuth [8,9]. However, it still requires a motor with a high torque capacity, which has
conditions of the Creative Commons moving parts and requires extensive maintenance.
Attribution (CC BY) license (https:// To overcome the above-mentioned issues, phased-array antennas (PAAs) are preferred,
creativecommons.org/licenses/by/ since they can direct their main beam anywhere without requiring physical rotation. This
4.0/). concept involves arranging several small elements in such a way that controls the radiation

Appl. Sci. 2024, 14, 2730. https://doi.org/10.3390/app14072730 https://www.mdpi.com/journal/applsci

Appl. Sci. 2024, 14, 2730 2 of 13

phase of each element individually. A progressive phase shift is applied between the
elements in order to determine the beam pointing angle. Additionally, it also allows the
tracking of multiple satellites or objects simultaneously. The main drawback of conven-
tional PAAs is the need for separate transmit (Tx) and receive (Rx) antennas due to their
different operating frequencies, which significantly increases the actual occupied space and
manufacturing cost. Therefore, it is necessary to combine the Tx and Rx antennas into a
single radiating aperture to minimize the overall system volume. However, the design of
such a shared-aperture dual-band dual-polarized (DBDP) antenna is challenging. The com-
bination of two orthogonally arranged dual-band single-polarized antennas is the most
common method used to realize a shared-aperture DBDP antenna [10–14].
Nevertheless, this approach has two inherent drawbacks. First, it is difficult to control
the two frequency bands independently since they use the same design parameters. Second,
a mechanism to distinguish between Tx and Rx operations would also be required. A more
efficient solution is to implement four different excitation ports, two for the low-band (LB)
element and two for the high-band (HB) element, for the realization of a DBDP antenna with
a shared aperture. This approach provides better electrical performance and reduces the
manufacturing complexity. Despite these advantages, only a few four-port shared-aperture
antennas with DBDP operation have been reported in the literature [15–17].
In this paper, a four-port shared-aperture DBDP antenna is proposed based on the mul-
tilayer printed circuit board (PCB) technology for Tx/Rx Ku-band satellite communication
applications.
The antenna consists of an LB L-probe proximity-fed square-ring antenna and a HB
stacked-patch antenna, sharing a common radiating aperture. Each antenna has two
excitation ports for dual-polarized operation. The proposed antenna is designed and
optimized to cover two frequency bands, 10.7–12.75 GHz (Ku-band downlink) for Rx mode
and 13.75–14.5 GHz (Ku-band uplink) for Tx mode, with a frequency ratio of approximately
1.2. Moreover, the antenna has four ports, making it easy to tune the two operating
frequency bands at different ratios.

2. Antenna Design and Working Mechanism

2.1. Antenna Configuration
The geometry of the proposed four-port shared-aperture DBDP antenna and its side
view are shown in Figure 1a,b, respectively. The antenna is designed with a multilayer PCB
process. It has six substrate layers (sub-1–sub-6), five prepreg layers (prepreg 1–5), and seven
copper layers (M1–M7). The substrates are Taconic TLE-95 (ε 1 = 2.95 and tanδ1 = 0.0028).
The prepreg materials have the same thickness of h p and are based on Rogers RO4450T
material (ε 2 = 3.23 and tanδ2 = 0.0039).
The dual-band operation of the proposed antenna is achieved by two separate an-
tennas, one LB antenna and one HB antenna, sharing a common radiating aperture. The
LB antenna consists of a square-ring radiator with four chamfered corners printed on the
copper layer M1. Two L-probe feeds, each comprising a stripline embedded in copper layer
M2 and an LB feed, are implemented and positioned in the x- and y-directions to excite
the square ring for dual polarization at the low band. In order to improve the impedance
matching of the LB antenna, three protruding stubs are attached directly to the corners of
the square ring. The HB antenna is a stacked-patch radiator, consisting of two identical
plus-shaped parasitic patches printed on the copper layers M2 and M3 and a plus-shaped
driven patch printed on the copper layer M4. Two capacitive feeds are used and positioned
in the x- and y-directions to excite the driven patch for dual polarization at the high band.
Four identical open rectangular slots are etched symmetrically on each parasitic patch
to lower the operational frequency band of the HB antenna. In addition, four identical
arc-shaped slots are etched symmetrically on the driven patch to improve the impedance
matching. Due to the inherently poor isolation of the capacitive feeding, the HB antenna is
equipped with a shorting pin to improve the isolation between two orthogonal polariza-
tions. The proposed antenna is excited by four excitation ports P1–P4 through four 50-Ω
Appl. Sci. 2024, 14, 2730 3 of 13

coaxial cables. Each coaxial cable is connected to a feed via a grounded coplanar waveg-
uide (GCPW) consisting of a feedline and a bottom ground plane on the copper layer M7,
a top ground plane on the copper layer M6, and a series of orange blind vias penetrating
through the substrate layer sub-6. A cavity formed by a square ring printed on the copper
layer M5 and a series of green blind vias surrounding the antenna is used to define the
array lattice size for array construction. The size of the cavity is 10.3 mm × 10.3 mm. This
corresponds to an array with spacing of 0.43λ12.75GHz and 0.50λ14.5GHz between elements
at 12.75 and 14.5 GHz, respectively, allowing a maximum achievable scan angle of ±70◦
for both operating bands. The detailed configurations of each copper layer are shown in
Figure 2, with the optimized design parameters summarized in the caption. The antenna
was optimized using Ansys HFSS 2020 R2.

LB antenna
sub-1(h1)

prepreg 1 (hp)
sub-2 (h2)
LB feed vias
prepreg 2 (hp)
sub-3 (h3)
HB antenna
prepreg 3 (hp)
sub-4 (h4)
HB feed vias

shorting pin prepreg 4 (hp)

sub-5 (h5)

prepreg 5 (hp)
top ground

cavity
sub-6 (h6)

P3 P4 feedline

z bottom ground
y
x P2 P1 coaxial cable

dielectric prepreg copper feeding

LB feed vias (a) HB feed vias
M1
M2
M3
cavity vias
M4
M5 coaxial cable
M6 z
M7
feeding vias (b) shorting pin y x
Figure 1. Geometry of the proposed four-port shared-aperture DBDP antenna. (a) 3D exploded view.
(b) Side view. (h1 = 0.13 mm, h2 = 0.25 mm, h3 = 1.58 mm, h4 = h5 = 0.51 mm, h6 = 0.25 mm, and
h p = 0.076 mm).
Appl. Sci. 2024, 14, 2730 4 of 13

wsub r0 a
a5 a3 b2
b3 w0
b1
a2 a0 a6 b0 b4
a7 b5 r1
a4 a8
a1
z y z y z y r2
x x x r3
(a) (b) (c)
w ca v z y z y
ws z y x x w
s via
f 2
s via 2
x
fTx
s via
f Rx d1
w1
s via 1 ps d0
s via

(d) (e) (f)

Figure 2. Detailed configurations of each copper layer. (a) Copper layer M1. (b) Copper layer M2.
(c) Copper layer M4. (d) Copper layer M5. (e) Copper layer M6. (f) Copper layer M7. (Unit: mm,
a0 = 4.82, a1 = 0.735, a2 = 0.45, a3 = 1.135, a4 = 0.5, a5 = 0.35, a6 = 0.8, a7 = 0.86, a8 = 0.6, b0 = 3.66,
b1 = 1.16, b2 = 0.25, b3 = 0.65, b4 = 3.63, b5 = 1.16, r0 = 0.73, r1 = 0.34, r2 = 0.44, r3 = 0.64, w0 = 0.2,
ws = 0.6, wcav = 10.3, ps = 0.3, svia = 1.645, svia1 = 1.115, f Rx = 2.46, f Tx = 1.46, svia2 = 0.7, d0 = d1 = 1.2,
w f = 0.6, w1 = 1, wsub = 14.57, α = 76◦ ).

2.2. Working Mechanism

The proposed antenna is developed based on the design concept presented in [15].
This antenna, however, has a large size and a high frequency ratio of approximately 2.
Therefore, this work aims to design a compact shared-aperture DBDP antenna with a
smaller frequency ratio in order to meet the requirements for the downlink and uplink
bands of the Ku band. The following sections describe the key steps involved in designing
the proposed antenna.
Figure 3a illustrates the design steps for the LB antenna. Two reference designs,
including Ant.1 with no protruding stub and Ant.2 with one protruding stub, are also
investigated to compare with the proposed one. The simulated |S11 | and |S21 | are shown in
Figure 3b,c, respectively. As observed, the protruding stubs help to improve the impedance
matching of the LB antenna significantly (see Figure 3b). Moreover, high isolation between
two orthogonal polarizations |S21 | of the LB antenna is retained mostly by implementing
three protruding stubs (see Figure 3c). Referring to Figure 3a, the LB antenna exhibits a
−10 dB impedance bandwidth (IBW) covering the entirety of the desired frequency band of
10.7–12.75 GHz. The simulated current distributions observed at two resonant frequencies
of 10.2 GHz and 13 GHz when the LB antenna is excited by P1 are illustrated in Figure 3d.
At each resonant frequency, two currents appear, one in the clockwise direction and another
in the anti-clockwise direction, demonstrating the fundamental TM11 mode operating over
the operating bandwidth [18,19].
Appl. Sci. 2024, 14, 2730 5 of 13

stubs
Ant. 1 Ant. 2 Proposed
(a)
0 10.7 - 12.75 0
Ant. 1
-5 GHz -5 Ant. 2
Proposed

S 21 (dB)
S 11 (dB)

-10 -10
10.7 - 12.75
-15 -15 GHz
-20 -20
Ant. 1
-25 Ant. 2 -25
Proposed
-30 -30
9 10 11 12 13 14 9 10 11 12 13 14
Frequency (GHz) Frequency (GHz)
(b) (c)

10.2 GHz 13 GHz

P1 P1
(d)
Figure 3. (a) Design evolution of the LB antenna. (b) |S11 |. (c) |S21 |. (d) Simulated current distributions
at 10.2 GHz and 13 GHz.

The HB antenna is designed according to the procedure shown in Figure 4, and the
simulated |S33 | of the HB antenna according to each procedure is shown in Figure 5 (the
S-parameters of the LB antenna are not shown here for brevity). In addition, the simulation
showed that the HB antenna design procedure had a minor impact on the LB antenna’s
performance. As a starting point, a capacitive fed microstrip patch antenna (HB-Ant.1) is
selected and designed as shown in Figure 4a. The miniaturization of the patch is achieved
by cutting off four small squares from each corner. As shown in Figure 5, HB-Ant.1 exhibits
poor impedance matching for the frequency of interest from 13.75 to 14.5 GHz. Thus, it is
necessary to improve the impedance matching of HB-Ant.1. To improve the impedance
matching, a plus-shaped parasitic patch is introduced and located above the driven patch
of HB-Ant.2. As illustrated in Figure 5, the impedance matching of HB-Ant.2 is improved
for the frequency range of 14.1–15.2 GHz. In HB-Ant.3, one more plus-shaped parasitic
patch is added, significantly improving the reflection coefficients (see Figure 5). At this
point, instead of increasing the patch sizes to extend the impedance matching range further
Appl. Sci. 2024, 14, 2730 6 of 13

to cover the frequency of 13.75–14.5 GHz, four rectangular slots are cut off from each
parasitic patch of HB-Ant.4. The |S33 | of HB-Ant.4 indicates that it operates from 13.6
to 15.7 GHz (see Figure 5). Moreover, the reflection coefficient of the driven patch is
significantly enhanced by applying arc-shaped slots in the final step. Through the inclusion
of additional slots, the compactness of the HB antenna can be achieved without increasing
its physical dimensions. According to Figure 5, the proposed HB antenna shows a −10 dB
IBW, covering the entire desired frequency range of 13.75–14.5 GHz.

z
(a) y (b)
x
rectangular slots

(d) (c)

arc-shaped slot

(e)
Figure 4. Step-by-step antenna design for the HB antenna (other components are hidden for clear
illustration). (a) HB-Ant.1. (b) HB-Ant.2. (c) HB-Ant.3. (d) HB-Ant.4. (e) Proposed HB antenna.

An overview of the shorting pin’s function is also presented in Figure 6. As shown

in Figure 6a, the HB antenna’s resonance does not change significantly as a result of the
shorting pin. With the shorting pin, the HB antenna still exhibits a wideband operating
frequency band covering the entire designed frequency range of the Tx band. According to
Figure 6b, employing the shorting pin in the HB antenna improves the isolation between
its two orthogonal polarization ports (|S43 |). The current distribution of the HB antenna
without and with the shorting pin at 14 GHz is illustrated in Figure 6c. Without the shorting
pin, an induced current will occur when the HB antenna is excited through P3 and flow
to P4 (see Figure 6c), thus resulting in poor isolation between two ports. The function of
the shorting pin is to enhance the isolation by shorting this induced current. As shown in
Figure 6c, a strong current is generated in the shorting pin, while the induced current on
the feed via P4 becomes weaker.
Appl. Sci. 2024, 14, 2730 7 of 13

0
13.57 - 14.5
GHz
-5

S 33 (dB)
-10

-15

-20 HB-Ant.3
HB-Ant.1 HB-Ant.4
HB-Ant.2 Proposed
-25
13 13.5 14 14.5 15 15.5 16
Frequency (GHz)
Figure 5. Simulated |S33 | for each step of the design procedure for the HB antenna.

0 0
wo/shorting pin 13.57 - 14.5
-5 w/shorting pin -5 GHz
13.57 - 14.5
S 33 (dB)

S 43 (dB)
-10 GHz -10
-15 -15
-20 -20 wo/shorting pin
w/shorting pin
-25 -25
13 13.5 14 14.5 15 15.5 16 13 13.5 14 14.5 15 15.5 16
Frequency (GHz) Frequency (GHz)
(a) (b)
z
y
x

P4 P3 P4 shorting pin P3
(c)
Figure 6. Effect of the shorting pin on the (a) |S33 | and (b) |S43 | of the HB antenna. (c) Simulated
current distribution of the HB antenna with and without the shorting pin when P3 excited.

3. Experimental Results and Discussion

Based on the optimized design parameters, the proposed four-port shared-aperture
DBDP antenna is fabricated with six Taconic TLE-95 substrate layers of different thicknesses.
These substrate layers are bonded together using Roger RO4450T bonding sheets with the
same thickness of 0.0762 mm. Figure 7a shows a photograph of the fabricated antenna. The
antenna is 14.55 mm × 14.55 mm × 3.61 mm in size. Each excitation port of the antenna is
connected with a semi-flexible pigtail cable (part number: SR047-KF1). The S-parameters
were tested using the ShockLine™ MS46122B vector network analyzer from Anritsu. The
far-field radiation characteristics of the fabricated antenna were measured in an RF anechoic
Appl. Sci. 2024, 14, 2730 8 of 13

chamber, as shown in Figure 7b. A dual-polarized rectangular horn antenna was utilized
to attain the co-polarized/cross-polarized waves.

Top view Back view

(a)

standard horn

antenna under test

(b)
Figure 7. Photograph of the (a) fabricated antenna and (b) far-field measurement environment.

Figure 8a–c illustrate a comparison of the simulated and measured S-parameters of

the proposed antenna. The simulation and measurement are found to be in reasonable
agreement. The measured results are slightly shifted down approximately 300 MHz in
comparison with the simulated results. Multilayer PCB manufacturing uncertainties are
primarily responsible for this discrepancy. Referring to Figure 8a, the measured −10 dB IBW
of 9.28–12.96 GHz (33.10%) is achieved for both P1 and P2 of the LB antenna, covering the
entire Rx band of 10.7–12.75 GHz. Within the −10 dB IBW, the measured isolation between
two orthogonal polarizations (|S21 |) of the LB antenna is larger than 12 dB. Referring to
Figure 8b, the measured −10 dB IBWs of the LB antenna are 13.21–15.88 GHz for P3 and
13.10–15.32 GHz for P4. Although the antenna is symmetrical, there is a discrepancy in the
measured results of |S33 | and |S44 |, which is mainly caused by imperfections in the hand
soldering process. An overlapped IBW of 13.21–15.32 GHz (14.79%) is achieved for the
HB antenna, covering the whole Tx band of 13.75–14.5 GHz. Within the overlapped IBW,
the measured isolation between two orthogonal polarizations (|S43 |) of the HB antenna is
larger than 12.4 dB. The Tx/Rx isolations, namely |S31 | and |S41 |, are depicted in Figure 8c.
As observed, the measured Tx/Rx isolation of more than 14 dB is achieved in both the Rx
band and Tx band.
Appl. Sci. 2024, 14, 2730 9 of 13

0 S 11 - mea.
S 22 - mea.

S-parameters (dB)
-5 S 11 , S 22 - sim.
-10
-15
-20
-25 10.7 - 12.75 S 21 - sim.
GHz S 21 - mea.
-30
9 10 11 12 13 14 15 16
Frequency (GHz)
(a)
0 13.75 - 14.5
0
13.75 - 14.5

S-parameters (dB)
S-parameters (dB)

-5 GHz -5 GHz
10.7 - 12.75
-10 -10 GHz

-15 -15
S 33 , S 44 - sim.
-20 S 33 - mea. -20
S 44 - mea.
-25 S 43 - sim. -25 S 31 - sim. S 41 - sim.
S 43 - mea. S 31 - mea. S 14 - mea.
-30 -30
9 10 11 12 13 14 15 16 9 10 11 12 13 14 15 16
Frequency (GHz) Frequency (GHz)
(b) (c)
Figure 8. (a) S-parameters of the LB antenna. (b) S-parameters of the HB antenna. (c) Tx/Rx isolation.

The simulated and measured realized gains of the proposed antenna are depicted in
Figure 9a,b. Within the Rx band, the measured realized gain ranges from 4.18 to 6.02 dBi
for P1 and from 4.62 to 6.63 dBi for P2. Meanwhile, the measured realized gains of 4.99 to
5.42 dBi and 4.65 to 5.23 dBi are achieved for P3 and P4, respectively, within the Tx band.
The measured and simulated radiation patterns of the proposed antenna in the xz-plane
(ϕ = 0◦ ) and yz-plane (ϕ = 90◦ ) at 11.75 GHz (for Rx band) and 14 GHz (for Tx band) are
plotted in Figure 10a–d. The measured cross-polarization in the boresight direction remains
≥20.1 dB and ≥18.4 dB in the Rx band and Tx band, respectively. It is also seen from
Figure 10 that the off-axis cross-polarization in the E-plane is high, especially at 14 GHz
(Tx band). Sequential rotated feeding can be applied to overcome this problem when
designing a Tx/Rx dual-polarized phased-array antenna based on the proposed design.
We compare the proposed antenna with previously reported DBDP antennas in Table 1.
Compared to previous antennas [13,15,16], the proposed antenna has a smaller frequency
ratio and a smaller size. The designs in [13,14] are two-port DBDP antennas, whereas the
rest, including the proposed design, are four-port DBDP antennas. Although the design
in [17] has a higher gain, the proposed antenna offers much wider IBWs and is much
more compact.
Appl. Sci. 2024, 14, 2730 10 of 13

8 8

Realized Gain (dBi)

Realized Gain (dBi)

6 6
4 4
10.7 - 12.75 13.75 - 14.5
2 GHz 2 GHz

0 0
mea. P1 mea. P3
-2 mea. P2 -2 mea. P4
sim. P1, P2 sim. P 3 , P 4
-4 -4
9 10 11 12 13 14 13 13.5 14 14.5 15 15.5 16
Frequency (GHz) Frequency (GHz)
(a) (b)
Figure 9. Simulated and measured realized gains of (a) LB antenna and (b) HB antenna.

simulation
measurement
o o

-30
o q=0
30
o Co-pol. -30
o q=0
30
o

o
Cross-pol. o o
Cross-pol. o
-60 60 -60 60

-90
o -30 -20 -10 0 90 o o -30 -20 -10 0 90 o
-90

o o
o 120 o 120
-120 -120
o o
o 150 o 150
-150 o -150 o
180 180
E-plane E-plane
o o
o q=0 o Co-pol. o q=0 o
-30 30 -30 30
o Cross-pol. o o Cross-pol. o
-60 60 -60 60

-90
o -30 -20 -10 0 90 o-90 o -30 -20 -10 0 90 o

o o
o 120 o 120
-120 -120
o o
o 150 o 150
-150 o -150 o
180 180
H-plane H-plane
(a ) (b)
o o
Figure 10. Cont.o
-30
q=0
30
o Co-pol. -30
o q=0
30
o

Cross-pol. Cross-pol.
o o o o
-60 60 -60 60

-90
o -30 -20 -10 0 90 o o -30 -20 -10 0 90 o
-90
 o o
o 120 o 120
-120 -120
o o
o 150 o 150
-150 o -150 o
180 180
Appl. Sci. 2024, 14, 2730
H-plane H-plane 11 of 13

(a ) (b)
o o

-30
o q=0
30
o Co-pol. -30
o q=0
30
o

Cross-pol. Cross-pol.
o o o o
-60 60 -60 60

-90
o -30 -20 -10 0 90 o o -30 -20 -10 0 90 o
-90

o o
o 120 o 120
-120 -120
o o
o 150 o 150
-150 o -150 o
180 180
E-plane E-plane
o o
o q=0 o Co-pol. o q=0 o
-30 30 -30 30
o Cross-pol. o o Cross-pol. o
-60 60 -60 60

o -30 -20 -10 0 90 o o -30 -20 -10 0 o

-90 -90 90

o o
o 120 o 120
-120 -120
o o
o 150 o 150
-150 o -150 o
180 180
H-plane H-plane
(c ) (d)
Figure 10. Simulated and measured normalized radiation patterns of the proposed antenna at
(a) 11.75 GHz, fed by P1, (b) 11.75 GHz, fed by P2, (c) 14 GHz, fed by P3, and (d) 14 GHz, fed by P4.

Table 1. Comparison with previous DBDP antennas.

IBWs Peak Gain Frequency Tx/Rx

Ref. Size [λ2L ]
LB/HB [%] LB/HB [dBi] Ratio Isolation
[13] ⋆ 23.7/12.8 N.M. 1.7 >17 0.64 × 0.64
[14] ⋆ 22/16 5.0/5.0 1.5 N.M. 50 × 0.50
[15] 28.3/29.4 7.04/6.94 2.0 N.M. 0.83 × 0.83 ⋆⋆
[16] ⋆ 8.0/10.0 8.7/8.3 2.0 N.M. 0.75 × 0.75
[17] 13.7/5.3 ⋆⋆ 8.0/7.0 1.2 >17 0.86 × 0.86
Proposed 33.10/14.79 6.63/5.42 1.2 >14 0.56 × 0.56
λ L is the free space wavelength at the center frequency of the low band. ⋆ Simulated. ⋆⋆ Estimated from graph.
N.M.: not mentioned.

4. Conclusions
A four-port shared-aperture DBDP antenna for Tx/Rx Ku-band SATCOM applications
was developed, fabricated, and tested. The antenna was designed using the multilayer
printed circuit board technology, consisting of an LB L-probe proximity-fed square-ring
antenna and a HB stacked-patch antenna, sharing a common aperture. Through the
experiment, an IBW of 9.28–12.96 GHz (13.21–15.32 GHz), an isolation value between two
orthogonal polarizations of 12 dB (12.4 dB), and a peak gain of 6.63 dBi (5.42 diBi) were
achieved at the low band (high band). The Tx/Rx isolation was more than 14 dB in both the
Appl. Sci. 2024, 14, 2730 12 of 13

Rx band and Tx band. The proposed antenna exhibits dual-band dual-polarized operation,
covering two frequency bands, 10.7–12.75 GHz (Ku-band downlink) for Rx mode and
13.75–14.5 GHz (Ku-band uplink) for Tx mode; therefore, it can be directly used as a basic
unit to develop a Ku-band Tx/Rx phased array for LEO SATCOM applications. This is
considered the future work of the proposed design. In addition, the design can easily be
scaled to operate in the Ka band (27.5–31 GHz for uplink and 17.7–21.2 GHz for downlink).

Author Contributions: Conceptualization, S.T.-V., W.Y.Y., H.W.C. and K.C.H.; methodology, S.T.-V.
and K.C.H.; software, S.T.-V.; validation, S.T.-V., W.Y.Y., H.W.C. and K.C.H.; formal analysis, S.T.-V.;
investigation, S.T.-V. and K.C.H.; resources, S.T.-V. and K.C.H.; data curation, S.T.-V., W.Y.Y., H.W.C.
and K.C.H.; writing—original draft preparation, S.T.-V.; writing—review and editing, S.T.-V., W.Y.Y.,
H.W.C. and K.C.H.; visualization, S.T.-V.; supervision, K.C.H.; project administration, K.C.H. All
authors have read and agreed to the published version of the manuscript.
Funding: This work was supported by the Research Fund of Hanwha Systems (Hanwha Phasor).
Institutional Review Board Statement: Not applicable.
Informed Consent Statement: Not applicable.
Data Availability Statement: All data have been included in study.
Conflicts of Interest: Authors Woo Yong Yang and Hyung Won Cho were employed by the The
UK Office (Hanwha Phasor). The authors declare that this study received funding from Hanwha
Systems (Hanwha Phasor). The funder was not involved in the study design, collection, analysis,
interpretation of data, the writing of this article or the decision to submit it for publication.

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