Model XC1900A-03 Rev D

Hybrid Coupler
3 dB, 90°
Description
The XC1900A-03 is a low profile, high performance 3dB hybrid coupler in a
new easy to use, manufacturing friendly surface mount package. It is
designed for DCS and PCS band applications. The XC1900A-03 is
designed particularly for balanced power and low noise amplifiers, plus
signal distribution and other applications where low insertion loss and tight
amplitude and phase balance is required. It can be used in high power
applications up to 150 Watts.

Parts have been subjected to rigorous qualification testing and they are
manufactured using materials with coefficients of thermal expansion (CTE)
compatible with common substrates such as FR4, G-10, RF-35, RO4350
and polyimide. Available in both 5 of 6 tin lead (XC1900A-03P) and 6 of 6
tin immersion (XC1900A-03S) RoHS compliant finishes.
Electrical Specifications **
Features: Insertion Amplitude
Frequency Isolation VSWR
• 1400-2000 MHz Loss Balance
• DCS and PCS MHz dB Min dB Max Max : 1 dB Max
• High Power 1400-1500 23 0.08 1.15 0 to -0.4
• Very Low Loss 1700-2000 25 0.15 1.15 ± 0.13
• Tight Amplitude Balance 1805-1880 27 0.12 1.12 ± 0.10
• High Isolation 1930-1990 27 0.12 1.12 ± 0.10
• Production Friendly Operating
Phase Power ΘJC
• Tape and Reel Temp.
• Available in Lead-Free (as Degrees Avg. CW Watts ºC/Watt ºC
illustrated) or Tin-Lead 90 ± 2.0 150 28 -55 to +95
• Reliable, FIT=0.53 90 ± 2.0 150 28 -55 to +95
90 ± 2.0 150 28 -55 to +95
90 ± 2.0 150 28 -55 to +95
**Specification based on performance of unit properly installed on Anaren Test Board 58481-0001 with small
signal applied. Specifications subject to change without notice. Refer to parameter definitions for details.

Top View (Near-Side) Side View Bottom View (Far-Side)

.069±.014
[1.75±0.35]
.560±.010 4X .040±.004
[14.22±0.25] [1.02±0.10] Pin 1
Pin 1 Pin 2 Pin 2

Orientation GND
Marker Denotes .350±.010 .220±.004
4X .059±.004 SQ
Pin 1 [8.89±0.25] [5.59±0.10]
[1.50±0.10]

Pin 4 Denotes GND Pin 3 Pin 3 .430±.004

[10.92±0.10] Pin 4
Array Number

Dimensions are in Inches [Millimeters] *For RoHS Compliant Versions order with S suffix
XC1900A-03* Mechanical Outline Tolerances are Non-Cumulative

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This datasheet has been downloaded from http://www.digchip.com at this page

 Model XC1900A-03
Rev D

Hybrid Coupler Pin Configuration

The XC1900A-03 has an orientation marker to denote Pin 1. Once port one has been identified the other ports are
known automatically. Please see the chart below for clarification:

Configuration Pin 1 Pin 2 Pin 3 Pin 4

Splitter Input Isolated -3dB ∠θ − 90 -3dB ∠θ
Splitter Isolated Input -3dB ∠θ -3dB ∠θ − 90
Splitter -3dB ∠θ − 90 -3dB ∠θ Input Isolated
Splitter -3dB ∠θ -3dB ∠θ − 90 Isolated Input

*Combiner A ∠θ − 90 A ∠θ Isolated Output

*Combiner A ∠θ A ∠θ − 90 Output Isolated
*Combiner Isolated Output A ∠θ − 90 A ∠θ
*Combiner Output Isolated A ∠θ A ∠θ − 90

*Note: “A” is the amplitude of the applied signals. When two quadrature signals with equal amplitudes are
applied to the coupler as described in the table, they will combine at the output port. If the amplitudes are
not equal, some of the applied energy will be directed to the isolated port.

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Insertion Loss and Power Derating Curves

Typical Insertion Loss Derating Curve for XC1900A-03 Power Derating Curve for XC1900A-03
0 225
typical insertion loss (f=1880MHz) power handling at 1880MHz
-0.02 typical insertion loss (f=1990MHz) 200 power handling at 1990MHz
typical insertion loss (f=2000MHz) power handling at 2000MHz
-0.04
175

-0.06
150
I n s e rt io n L o s s (d B )

-0.08

P o w e r (W a t t s )
125
-0.1
100
-0.12
75
-0.14

50
-0.16

-0.18 25

-0.2 0
-100 -50 0 50 100 150 200 250 300 350 0 25 50 75 100 125 150 175 200 225
Temperature of the Part (ºC) Base Plate Temperature (ºC)

Insertion Loss Derating: Power Derating:

The insertion loss, at a given frequency, of a group of The power handling and corresponding power derating
couplers is measured at 25°C and then averaged. The plots are a function of the thermal resistance, mounting
measurements are performed under small signal surface temperature (base plate temperature),
conditions (i.e. using a Vector Network Analyzer). The maximum continuous operating temperature of the
process is repeated at 95°C, 150°C, and 200°C. A best- coupler, and the thermal insertion loss. The thermal
fit line for the measured data is computed and then insertion loss is defined in the Power Handling section of
plotted from -55°C to 300°C. the data sheet.

As the mounting interface temperature approaches the

maximum continuous operating temperature, the power
handling decreases to zero.

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Typical Performance (-55°C, 25°C and 95°C): 1700-2000 MHz

Return Loss for XC1900A-03 (Feeding Port 1) Return Loss for XC1900A-03 (Feeding Port 2)
0 0
- 55ºC - 55ºC
-5 25ºC -5 25ºC
95ºC 95ºC
-10 -10

-15 -15
R e t u rn L o s s (d B )

R e t u rn L o s s (d B )
-20 -20

-25 -25

-30 -30

-35 -35

-40 -40

-45 -45

-50 -50
1700 1750 1800 1850 1900 1950 2000 1700 1750 1800 1850 1900 1950 2000
Frequency (MHz) Frequency (MHz)

Return Loss for XC1900A-03 (Feeding Port 4)

Return Loss for XC1900A-03 (Feeding Port 3) 0
0 - 55ºC
- 55ºC -5 25ºC
-5 25ºC 95ºC
95ºC -10
-10
-15
-15
R e t u rn L o s s (d B )

-20
R e t u rn L o s s (d B )

-20
-25
-25
-30
-30
-35
-35
-40
-40
-45
-45
-50
-50 1700 1750 1800 1850 1900 1950 2000
1700 1750 1800 1850 1900 1950 2000 Frequency (MHz)
Frequency (MHz)

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Typical Performance (-55°C, 25°C and 95°C): 1700-2000 MHz

Coupling for XC1900A-03 (Feeding Port 1) Insertion Loss for XC1900A-03 (Feeding Port 1)
-2.8 0
- 55ºC - 55ºC
25ºC 25ºC
95ºC -0.02 95ºC
-2.9

-0.04

-3

I n s e rt io n L o s s (d B )
-0.06
C o u p lin g (d B )

-3.1 -0.08

-0.1
-3.2

-0.12

-3.3
-0.14

-0.16
1700 1750 1800 1850 1900 1950 2000 1700 1750 1800 1850 1900 1950 2000
Frequency (MHz) Frequency (MHz)

Phase Balance for XC1900A-03 (Feeding Port 1) Isolation for XC1900A-03 (Feeding Port 1)
93 0
- 55ºC - 55ºC
25ºC -5 25ºC
95ºC 95ºC
92
-10

-15
P h a s e B a la n c e (D e g re e s )

91
-20
I s o la t io n (d B )

90 -25

-30
89
-35

-40
88
-45

87 -50
1700 1750 1800 1850 1900 1950 2000 1700 1750 1800 1850 1900 1950 2000
Frequency (MHz) Frequency (MHz)

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Definition of Measured Specifications

Parameter Definition Mathematical Representation

Vmax
The impedance match of VSWR =
VSWR the coupler to a 50Ω Vmin
(Voltage Standing Wave Ratio) system. A VSWR of 1:1 is Vmax = voltage maxima of a standing wave
optimal. Vmin = voltage minima of a standing wave

The impedance match of

the coupler to a 50Ω VSWR + 1
Return Loss system. Return Loss is Return Loss (dB)= 20log
an alternate means to VSWR - 1
express VSWR.
The input power divided Pin
Insertion Loss by the sum of the power Insertion Loss(dB)= 10log
at the two output ports. Pcpl + Pdirect
The input power divided Pin
Isolation by the power at the Isolation(dB)= 10log
isolated port. Piso
The difference in phase
Phase Balance angle between the two Phase at coupled port – Phase at direct port
output ports.

The power at each output

Pcpl Pdirect
10log and 10log
Amplitude Balance divided by the average ⎛ Pcpl + Pdirect ⎞ ⎛ Pcpl + Pdirect ⎞
power of the two outputs. ⎜ ⎟ ⎜ ⎟
⎝ 2 ⎠ ⎝ 2 ⎠

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Notes on RF Testing and Circuit Layout

The XC1900A-03 Surface Mount Couplers require the use of a test fixture for verification of RF performance. This test
fixture is designed to evaluate the coupler in the same environment that is recommended for installation. Enclosed
inside the test fixture, is a circuit board that is fabricated using the recommended footprint. The part being tested is
placed into the test fixture and pressure is applied to the top of the device using a pneumatic piston. A four port Vector
Network Analyzer is connected to the fixture and is used to measure the S-parameters of the part. Worst case values
for each parameter are found and compared to the specification. These worst case values are reported to the test
equipment operator along with a Pass or Fail flag. See the illustrations below.

3 & 5 dB 10 & 20 dB Test Board

Test Board Test Board In Fixture

Test Station

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The effects of the test fixture on the measured data must be minimized in order to accurately determine the
performance of the device under test. If the line impedance is anything other than 50Ω and/or there is a discontinuity
at the microstrip to SMA interface, there will be errors in the data for the device under test. The test environment can
never be “perfect”, but the procedure used to build and evaluate the test boards (outlined below) demonstrates an
attempt to minimize the errors associated with testing these devices. The lower the signal level that is being
measured, the more impact the fixture errors will have on the data. Parameters such as Return Loss and
Isolation/Directivity, which are specified as low as 27dB and typically measure at much lower levels, will present the
greatest measurement challenge.

The test fixture errors introduce an uncertainty to the measured data. Fixture errors can make the performance of the
device under test look better or worse than it actually is. For example, if a device has a known return loss of 30dB and
a discontinuity with a magnitude of –35dB is introduced into the measurement path, the new measured Return Loss
data could read anywhere between –26dB and –37dB. This same discontinuity could introduce an insertion phase
error of up to 1°.

There are different techniques used throughout the industry to minimize the affects of the test fixture on the
measurement data. Anaren uses the following design and de-embedding criteria:

• Test boards have been designed and parameters specified to provide trace impedances of 50
±1Ω. Furthermore, discontinuities at the SMA to microstrip interface are required to be less than
–35dB and insertion phase errors (due to differences in the connector interface discontinuities
and the electrical line length) should be less than ±0.25° from the median value of the four
paths.

• A “Thru” circuit board is built. This is a two port, microstrip board that uses the same SMA to
microstrip interface and has the same total length (insertion phase) as the actual test board. The
“Thru” board must meet the same stringent requirements as the test board. The insertion loss
and insertion phase of the “Thru” board are measured and stored. This data is used to
completely de-embed the device under test from the test fixture. The de-embedded data is
available in S-parameter form on the Anaren website (www.anaren.com).

Note: The S-parameter files that are available on the anaren.com website include data for frequencies that are
outside of the specified band. It is important to note that the test fixture is designed for optimum performance through
2.3GHz. Some degradation in the test fixture performance will occur above this frequency and connector interface
discontinuities of –25dB or more can be expected. This larger discontinuity will affect the data at frequencies above
2.3GHz.

Circuit Board Layout

The dimensions for the Anaren test board are shown below. The test board is printed on Rogers RO4350 material
that is 0.030” thick. Consider the case when a different material is used. First, the pad size must remain the same to
accommodate the part. But, if the material thickness or dielectric constant (or both) changes, the reactance at the
interface to the coupler will also change. Second, the linewidth required for 50Ω will be different and this will introduce
a step in the line at the pad where the coupler interfaces with the printed microstrip trace. Both of these conditions will
affect the performance of the part. To achieve the specified performance, serious attention must be given to the
design and layout of the circuit environment in which this component will be used.

If a different circuit board material is used, an attempt should be made to achieve the same interface pad reactance
that is present on the Anaren RO4350 test board. When thinner circuit board material is used, the ground plane will
be closer to the pad yielding more capacitance for the same size interface pad. The same is true if the dielectric
constant of the circuit board material is higher than is used on the Anaren test board. In both of these cases,
narrowing the line before the interface pad will introduce a series inductance, which, when properly tuned, will
compensate for the extra capacitive reactance. If a thicker circuit board or one with a lower dielectric constant is used,

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the interface pad will have less capacitive reactance than the Anaren test board. In this case, a wider section of line
before the interface pad (or a larger interface pad) will introduce a shunt capacitance and when properly tuned will
match the performance of the Anaren test board.

Notice that the board layout for the 3dB and 5dB couplers is different from that of the 10dB and 20dB couplers. The
test board for the 3dB and 5dB couplers has all four traces interfacing with the coupler at the same angle. The test
board for the 10dB and 20dB couplers has two traces approaching at one angle and the other two traces at a different
angle. The entry angle of the traces has a significant impact on the RF performance and these parts have
been optimized for the layout used on the test boards shown below.

10 & 20dB Test Board 3 & 5dB Test Board

Testing Sample Parts Supplied on Anaren Test Boards

If you have received a coupler installed on an Anaren produced microstrip test board, please remember to remove the
loss of the test board from the measured data. The loss is small enough that it is not of concern for Return Loss and
Isolation/Directivity, but it should certainly be considered when measuring coupling and calculating the insertion loss
of the coupler. An S-parameter file for a “Thru” board (see description of “Thru” board above) will be supplied upon
request. As a first order approximation, one should consider the following loss estimates:

Frequency Band Avg. Ins. Loss of Test Board @ 25°C

800 – 1000 MHz ~ 0.07dB
1700 – 2300 MHz ~ 0.12dB

For example, a 1900MHz, 10dB coupler on a test board may measure –10.30dB from input to the coupled port at
some frequency, F1. When the loss of the test board is removed, the coupling at F1 becomes -10.18dB (-10.30dB +
0.12dB). This compensation must be made to both the coupled and direct path measurements when calculating
insertion loss.

The loss estimates in the table above come from room temperature measurements. It is important to note that the
loss of the test board will change with temperature. This fact must be considered if the coupler is to be evaluated at
other temperatures.

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Peak Power Handling

High-Pot testing of these couplers during the qualification procedure resulted in a minimum breakdown voltage of
1.7KV (minimum recorded value). This voltage level corresponds to a breakdown resistance capable of handling at
least 12dB peaks over average power levels, for very short durations. The breakdown location consistently occurred
across the air interface at the coupler contact pads (see illustration below). The breakdown levels at these points will
be affected by any contamination in the gap area around these pads. These areas must be kept clean for optimum
performance. It is recommended that the user test for voltage breakdown under the maximum operating conditions
and over worst case modulation induced power peaking. This evaluation should also include extreme environmental
conditions (such as high humidity).

Orientation Marker

A printed circular feature appears on the top surface of the coupler to designate Pin 1. This orientation marker is not
intended to limit the use of the symmetry that these couplers exhibit but rather to facilitate consistent placement of
these parts into the tape and reel package. This ensures that the components are always delivered with the same
orientation. Refer to the table on page 2 of the data sheet for allowable pin configurations.

Test Plan

Xinger II couplers are manufactured in large panels and then separated. A sample population of parts is RF small
signal tested at room temperature in the fixture described above. All parts are DC tested for shorts/opens. (See
“Qualification Flow Chart” section for details on the accelerated life test procedures.)

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Power Handling

The average power handling (total input power) of a Xinger coupler is a function of:

• Internal circuit temperature.

• Unit mounting interface temperature.
• Unit thermal resistance
• Power dissipated within the unit.

All thermal calculations are based on the following assumptions:

• The unit has reached a steady state operating condition.

• Maximum mounting interface temperature is 95oC.
• Conduction Heat Transfer through the mounting interface.
• No Convection Heat Transfer.
• No Radiation Heat Transfer.
• The material properties are constant over the operating temperature range.

Finite element simulations are made for each unit. The simulation results are used to calculate the unit thermal
resistance. The finite element simulation requires the following inputs:

• Unit material stack-up.

• Material properties.
• Circuit geometry.
• Mounting interface temperature.
• Thermal load (dissipated power).

The classical definition for dissipated power is temperature delta (ΔT) divided by thermal resistance (R). The
dissipated power (Pdis) can also be calculated as a function of the total input power (Pin) and the thermal insertion loss
(ILtherm):

ΔT ⎛ − ILtherm
⎞
Pdis = = Pin ⋅ ⎜⎜1 − 10 10 ⎟
⎟ (W ) (1)
R ⎝ ⎠
Power flow and nomenclature for an “X” style coupler is shown in Figure 1.

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PIn POut (RL) POut (ISO)

Input Port Pin 1 Isolated Port

Coupled Port Pin 4 Direct Port

POut(CPL) POut (DC)

Figure 1

The coupler is excited at the input port with Pin (watts) of power. Assuming the coupler is not ideal, and that there are
no radiation losses, power will exit the coupler at all four ports. Symbolically written, Pout(RL) is the power that is
returned to the source because of impedance mismatch, Pout(ISO) is the power at the isolated port, Pout(CPL) is the
power at the coupled port, and Pout(DC) is the power at the direct port.

At Anaren, insertion loss is defined as the log of the input power divided by the sum of the power at the coupled and
direct ports:

Note: in this document, insertion loss is taken to be a positive number. In many places, insertion loss is written as a
negative number. Obviously, a mere sign change equates the two quantities.

⎛ Pin ⎞
IL = 10 ⋅ log10 ⎜ ⎟ (dB)
⎜P ⎟ (2)
⎝ out ( CPL ) + Pout ( DC ) ⎠
In terms of S-parameters, IL can be computed as follows:

IL = −10 ⋅ log10 ⎛⎜ S31 + S41 ⎞

2 2
⎟ (dB) (3)
⎝ ⎠
We notice that this insertion loss value includes the power lost because of return loss as well as power lost to the
isolated port.

For thermal calculations, we are only interested in the power lost “inside” the coupler. Since Pout(RL) is lost in the
source termination and Pout(ISO) is lost in an external termination, they are not be included in the insertion loss for
thermal calculations. Therefore, we define a new insertion loss value solely to be used for thermal calculations:

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⎛ Pin ⎞
ILtherm = 10 ⋅ log10 ⎜ ⎟ (dB )
⎜P + P + P + P ⎟ (4)
⎝ out ( CPL ) out ( DC ) out ( ISO ) out ( RL ) ⎠

In terms of S-parameters, ILtherm can be computed as follows:

ILtherm = −10 ⋅ log10 ⎛⎜ S11 + S 21 + S 31 + S 41 ⎞⎟

2 2 2 2
(dB ) (5)
⎝ ⎠
The thermal resistance and power dissipated within the unit are then used to calculate the average total input power
of the unit. The average total steady state input power (Pin) therefore is:

ΔT
Pdis R
Pin = = (W )
⎛ − ILtherm
⎞ ⎛ − ILtherm
⎞ (6)
⎜1 − 10 10 ⎟ ⎜1 − 10 10 ⎟
⎜ ⎟ ⎜ ⎟
⎝ ⎠ ⎝ ⎠
Where the temperature delta is the circuit temperature (Tcirc) minus the mounting interface temperature (Tmnt):

ΔT = Tcirc − Tmnt ( oC ) (7)

The maximum allowable circuit temperature is defined by the properties of the materials used to construct the unit.
Multiple material combinations and bonding techniques are used within the Xinger II product family to optimize RF
performance. Consequently the maximum allowable circuit temperature varies. Please note that the circuit
temperature is not a function of the Xinger case (top surface) temperature. Therefore, the case temperature cannot
be used as a boundary condition for power handling calculations.

Due to the numerous board materials and mounting configurations used in specific customer configurations, it is the
end users responsibility to ensure that the Xinger II coupler mounting interface temperature is maintained within the
limits defined on the power derating plots for the required average power handling. Additionally appropriate solder
composition is required to prevent reflow or fatigue failure at the RF ports. Finally, reliability is improved when the
mounting interface and RF port temperatures are kept to a minimum.

The power-derating curve illustrates how changes in the mounting interface temperature result in converse changes
of the power handling of the coupler.

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Mounting Coupler Mounting Process

In order for Xinger surface mount couplers to work The process for assembling this component is a
optimally, there must be 50Ω transmission lines leading conventional surface mount process as shown in Figure
to and from all of the RF ports. Also, there must be a 1. This process is conducive to both low and high volume
very good ground plane underneath the part to ensure usage.
proper electrical performance. If either of these two
conditions is not satisfied, electrical performance may not
meet published specifications.

Overall ground is improved if a dense population of

plated through holes connect the top and bottom ground
Figure 1: Surface Mounting Process Steps
layers of the PCB. This minimizes ground inductance
and improves ground continuity. All of the Xinger hybrid
Storage of Components: The Xinger II products are
and directional couplers are constructed from ceramic
available in either an immersion tin or tin-lead finish.
filled PTFE composites which possess excellent electrical
Commonly used storage procedures used to control
and mechanical stability having X and Y thermal
oxidation should be followed for these surface mount
coefficient of expansion (CTE) of 17-25 ppm/oC.
components. The storage temperatures should be held
between 15OC and 60OC.
When a surface mount hybrid coupler is mounted to a
printed circuit board, the primary concerns are; ensuring Substrate: Depending upon the particular component,
the RF pads of the device are in contact with the circuit the circuit material has an x and y coefficient of thermal
trace of the PCB and insuring the ground plane of neither expansion of between 17 and 25 ppm/°C. This coefficient
the component nor the PCB is in contact with the RF minimizes solder joint stresses due to similar expansion
signal. rates of most commonly used board substrates such as
RF35, RO4350, FR4, polyimide and G-10 materials.
Mounting Footprint Mounting to “hard” substrates (alumina etc.) is possible
To ensure proper electrical and thermal performance depending upon operational temperature requirements.
there must be a ground plane with 100% The solder surfaces of the coupler are all copper plated
solder connection underneath the part with either an immersion tin or tin-lead exterior finish.

Solder Paste: All conventional solder paste formulations

will work well with Anaren’s Xinger II surface mount
components. Solder paste can be applied with stencils or
syringe dispensers. An example of a stenciled solder
paste deposit is shown in Figure 2. As shown in the
figure solder paste is applied to the four RF pads and the
entire ground plane underneath the body of the part.

.430 Multiple
[10.92] plated thru holes
to ground

4X .040 .220
[1.02] [5.59]

4X .065 SQ
4X 50 Ω
[1.65]
Transmission
Line

Dimensions are in Inches [Millimeters]

XC1900A-03* Mounting Footprint

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Reflow: The surface mount coupler is conducive to most of

today’s conventional reflow methods. A low and high
temperature thermal reflow profile are shown in Figures 5
and 6, respectively. Manual soldering of these components
can be done with conventional surface mount non-contact
hot air soldering tools. Board pre-heating is highly
recommended for these selective hot air soldering
methods. Manual soldering with conventional irons should
be avoided.

Figure 2: Solder Paste Application

Coupler Positioning: The surface mount coupler can

be placed manually or with automatic pick and place
mechanisms. Couplers should be placed (see Figure 3
and 4) onto wet paste with common surface mount
techniques and parameters. Pick and place systems
must supply adequate vacuum to hold a 0.50-0.55
gram coupler.

Figure 3: Component Placement

Figure 4: Mounting Features Example

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Figure 5 – Low Temperature Solder Reflow Thermal Profile

Figure 6 – High Temperature Solder Reflow Thermal Profile

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Qualification Flow Chart

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 Model XC1900A-03 Rev D

Application Information

The XC1900A-03 is an “X” style 3dB (hybrid) coupler. Port configurations are defined in the table on page 2 of this data
sheet and an example driving port 1 is shown below.

Ideal 3dB Coupler Splitter Operation

1V 1 4 0.707V∠θ (-3dB)

Isolated Port 2 3 0.707V ∠θ -90 (-3dB)

The hybrid coupler can also be used to combine two signals that are applied with equal amplitudes and phase
quadrature (90º phase difference). An example of this function is illustrated below.

Ideal 3dB Coupler Combiner Operation

0.707V∠θ 1 4 Isolated Port

0.707V ∠θ -90 2 3 1V∠Φ

3dB couplers have applications in circuits which require splitting an applied signal into 2, 4, 8 and higher binary
outputs. The couplers can also be used to combine multiple signals (inputs) at one output port. Some splitting and
combining schemes are illustrated below:

2-Way Splitter/Combiner Network

Input * 50Ω
Amplitude and
Termination
Phase tracking
Devices
* 50Ω Output
Termination

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 Model XC1900A-03
Rev D

4-Way Splitter/Combiner Network

* 50Ω * 50Ω
Termination Term.
Amplitude and
Phase tracking
Input Devices
* 50Ω
Termination

* 50Ω
Termination Amplitude and
Phase tracking Output
Devices
* 50Ω
* 50Ω Termination
Term.

n
The splitter/combiner networks illustrated above use only 3dB (hybrid) couplers and are limited to binary divisions (2
number of splits, where n is an integer). Splitter/combiner circuits configured this way are known as “corporate”
networks. When a non-binary number of divisions is required, a “serial” network must be used. Serial networks can be
designed with [3, 4, 5, ….., n] splits, but have a practical limitation of about 8 splits.

A 5dB coupler is used in conjunction with a 3dB coupler to build 3-way splitter/combiner networks. An ideal version of
this network is illustrated below. Note what is required; a 50% split (i.e. 3dB coupler) and a 66% and 33% split (which is
actually a 4.77dB coupler, but due to losses in the system, higher coupler values, such as 5dB, are actually better
suited for this function). The design of this type of circuit requires special attention to the losses and phase lengths of
the components and the interconnecting lines. A more in depth look at serial networks can be found in the article
“Designing In-Line Divider/Combiner Networks” by Samir Tozin, which describes the circuit design in detail and can be
found in the White Papers Section of the Anaren website, www.anaren.com.

3-Way Splitter/Combiner

5 dB (4.77) 1/3 Pin 1/3 Pin 3 dB coupler * 50Ω

Pin coupler Termination
G=1

* 50Ω 2/3 Pin 1/3 Pin

Termination 2/3 Pin

1/3 Pin

G=1

3 dB coupler 5 dB (4.77) * 50Ω

coupler Termination

1/3 Pin
* 50Ω 1/3 Pin
Termination Pout
G=1

*Recommended Terminations
Power (Watts) Model
8 RFP-060120A15Z50
15 RFP-250375A4Z50
50 RFP-375375A6Z50
100 RFP-500500A6Z50

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 Model XC1900A-03 Rev D

Reflections From Equal Unmatched Terminations

Referring to the illustration below, consider the following reflection properties of the 3dB coupler. A signal applied to
port 1 is split and appears at the two output ports, ports 3 & 4, with equal amplitude and in phase quadrature. If ports
3 & 4 are not perfectly matched to 50Ω there will be some signal reflected back into the coupler. If the magnitude and
angle of these reflections are equal, there will be two signals that are equal in amplitude and in phase quadrature (i.e.
the reflected signals) being applied to ports 3 & 4 as inputs. These reflected signals will combine at the isolated port
and will cancel at the input port. So, terminations with the same mismatch placed at the outputs of the 3dB coupler will
not reflect back to the input port and therefore will not affect input return loss.

Γ× 0.707V ∠θ
Γ (0.5V ∠2θ + 0.5V ∠2θ -180) = 0V
0.707V∠θ (-3dB)
4
1V 1 Termination = ZL
ZL − Z 0
Γ=
ZL + Z 0
Isolated Port 2 Termination = ZL
3
0.707V ∠θ -90 (-3dB)
|Γ (0.5V ∠2θ -90 + 0.5V ∠2θ -90)| = |Γ|
Γ× 0.707V ∠θ -90

The reflection property of common mismatches in 3dB couplers is very beneficial to the operation of many networks.
For instance, when splitter/combiner networks are employed to increase output power by paralleling transistors with
similar reflection coefficients, input return loss is not degraded by the match of the transistor circuit. The reflections
from the transistor circuits are directed away from the input to the termination at the isolated port of the coupler.

This example is not limited to Power Amplifiers. In the case of Low Noise Amplifiers (LNA’s), the reflection property of
3dB couplers is again beneficial. The transistor devices used in LNA’s will present different reflection coefficients
depending on the bias level. The bias level that yields the best noise performance does not also provide the best
match to 50 Ω. A circuit that is optimized for both noise performance and return loss can be achieved by combining
two matched LNA transistor devices using 3dB couplers. The devices can be biased for the best noise performance
and the reflection property of the couplers will provide a good match as described above. An example of this circuit is
illustrated below:

LNA Circuit Leveraging the Reflection Property of 3dB Couplers

Input 50Ω
Termination

50Ω Output
Termination

Energy reflected from LNA Amplitude and phase tracking

devices biased for optimum LNA devices biased for
noise performance optimum noise performance

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 Model XC1900A-03
Rev D

Signal Control Circuits Utilizing 3dB Couplers

Variable attenuators and phase shifter are two examples of signal control circuits that can be built using 3dB couplers.
Both of these circuits also use the reflection property of the 3dB coupler as described above. In the variable attenuator
circuit, the two output ports of a 3dB coupler are terminated with PIN diodes, which are basically a voltage variable
resistor at RF frequencies (consult the literature on PIN diodes for a more complete equivalent circuit). By changing the
resistance at the output ports of the 3dB coupler, the reflection coefficient, Γ, will also change and different amounts of
energy will be reflected to the isolated port (note that the resistances must change together so that Γ is the same for
both output ports). A signal applied to the input of the 3dB coupler will appear at the isolated port and the amplitude of
this signal will be a function of the resistance at the output ports. This circuit is illustrated below:

Variable Attenuator Circuit Utilizing a 3dB Coupler

Γ× 0.707V ∠θ
0.707V∠θ (-3dB)
4
Input 1

Vdc PIN Diodes

Output 2
3
0.707V ∠θ -90 (-3dB)
|Γ (0.5V ∠2θ -90 + 0.5V ∠2θ -90)| = |Γ|
and Γ× 0.707V ∠θ -90
|Output| = | Γ|×|Input|

If Γ=0, no energy is reflected from the PIN diodes and S21 = 0 (input to output). If | Γ | =1, all of the energy is reflected
from the PIN diodes and |S21| = 1 (assuming the ideal case of no loss). The ideal range for Γ is –1 to 0 or 0 to 1, which
translate to resistances of 0Ω to 50Ω and 50Ω to ∞Ω respectively. Either range can be selected, although normally 0Ω
to 50Ω is easier to achieve in practice and produces better results. Many papers have been written on this circuit and
should be consulted for the details of design and operation.

Another very similar circuit is a Variable Phase Shifter (illustrated below). The same theory is applied but instead of PIN
diodes (variable RF resistance), the coupler outputs are terminated with varactors. The ideal varactor is a variable
capacitor with the capacitance value changing as a function of the DC bias. Ideally, the magnitude of the reflection
coefficient is 1 for these devices at all bias levels. However, the angle of the reflected signal does change as the
capacitance changes with bias level. So, ideally all of the energy applied to port 1, in the circuit illustrated below, will be
reflected at the varactors and will sum at port 2 (the isolated port of the coupler). However, the phase angle of the signal
will be variable with the DC bias level. In practice, neither the varactors nor the coupler are ideal and both will have
some losses. Again, many papers have been written on this circuit and should be consulted for the details of design and
operation.

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 Model XC1900A-03 Rev D

Variable Phase Shifter Circuit Utilizing a 3dB Coupler

Γ× 0.707V ∠θ
0.707V∠θ (-3dB)
4
Input 1

Vdc Varactor Diodes

Output 2
3
* |Γ (0.5V ∠2θ -90 + 0.5V ∠2θ -90)| =| Γ| 0.707V ∠θ -90 (-3dB)
Γ× 0.707V ∠θ -90

* The phase angle of the signal exiting port 2 will vary with the phase angle of Γ, which is the reflection
angle from the varactor. The varactors must be matched so that their reflection coefficients are equal.

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Model XC1900A-03
Rev D

Packaging and Ordering Information

Parts are available in both reel and tube. Packaging follows EIA 481-2. Parts are oriented in tape and reel as
shown below. Minimum order quantities are 2000 per reel and 30 per tube. See Model Numbers below for
further ordering information.

XX XXXX X - XX X

Xinger Coupler Frequency (MHz) Size (Inches) Coupling Value Plating Finish

0450 = 410-480 A = 0.56 x 0.35 03 = 3dB P = Tin Lead

0900 = 800-1000 B = 1.0 x 0.50 05 = 5dB S = Immersion Tin
1900 = 1700-2000 E = 0.56 x 0.20 10 = 10dB
XC 2100 = 2000-2300 L = 0.65 x 0.48 20 = 20dB
2500 = 2300-2700 M= 0.40 x 0.20 30 = 30dB
3500 = 3300-3700 P = 0.25 x 0.20

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