232 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 58, NO.

1, JANUARY 2011

Input Power Factor Compensation Algorithms Using

a New Direct-SVM Method for Matrix Converter
Hoang M. Nguyen, Hong-Hee Lee, Member, IEEE, and Tae-Won Chun

Abstract—An input filter is necessary for a matrix converter Input filter designs for MC guarantee near-unity power factor
(MC) system to improve the input current quality with low har- operation on the power supply side by improving the main input
monic components, as well as to reduce the input voltage distortion current quality, which has sinusoidal waveforms containing low
supplied to the MC. However, the input filter’s characteristics
make the input power factor (IPF) obtained at unity only in the harmonic components, and by reducing distortion of input volt-
presence of high output loads, and the IPF degrades significantly ages that are supplied to the MC module [5], [6]. The presence
under light-load conditions. In this paper, we propose a new direct of input filter in the direct ac–ac power conversion, which has
space vector modulation (DSVM) method to achieve the required no energy storage, can cause the instability during operations
displacement angle between the input voltage and input current [7]–[11]. In [12] and [13], the input filter design for sliding
of the MC. A new switching strategy is introduced based on the
maximum compensated angle. Then, power factor compensation mode controlled MC considered the maximum allowable dis-
algorithms using the new DSVM method to achieve the maximum placement angle introduced by the filter and the controllable
IPF are presented, in which compensation algorithm I is based on IPF capability, as well as the ripple presented in capacitor
using the input filter and power supply parameters to estimate voltages. The authors in [14] proposed an integration of MC
the optimal compensated angle. Compensation algorithm II is with filters that provides lower electromagnetic interference,
subsequently proposed using a proportional–integral controller
to overcome drawbacks presented in compensation algorithm I. lower common-mode current, and lower shaft voltage.
Simulation and experimental results are shown to validate the However, the basic hardware limitation of the input filter
effectiveness of the proposed compensation algorithms. still exists, which results in a displacement angle between input
Index Terms—Direct space vector modulation (DSVM) method, line-to-neutral voltage and input line current at the main power
input filter, matrix converter (MC). supply. Consequently, the IPF at the power supply could be far
from the desired power factor of unity. In particular, in the case
I. I NTRODUCTION of low-output-power condition, the IPF at the power supply
would decrease significantly.

I N THE PAST two decades, the evolution of power device

technology and the development of large-power integrated
circuits have revised the direct ac–ac power conversion tech-
In order to overcome this problem, we proposed a new
direct space vector modulation (DSVM) method based on the
maximum controllable displacement angle between the input
nologies. These types of converter fulfill all requirements of current and input voltage of the MC. The new DSVM method
conventionally used rectifier/dc link/inverter structures and pro- was developed by using a new pulsewidth modulation (PWM)
vide efficient ways to convert electric power for motor drives, switching pattern which authors already introduced the basic
uninterruptible power supplies, variable-frequency generators, idea succinctly in [15]. In this paper, two IPF compensation
and reactive energy control [1]–[4]. algorithms using the new DSVM are proposed to improve
A matrix converter (MC) delivers the following advantages: the IPF of the MC. First, a compensation algorithm based on
1) sinusoidal input and output current waveforms; the calculation of the optimal compensated angle is analyzed
2) controllable input power factor (IPF); and discussed. This compensation algorithm provides a fast
3) operations in all four quadrants of the torque–speed plane response which allows high IPF achievement. However, the
due to the regenerative capability; accuracy of this algorithm depends on some of the MC’s
4) high reliability and long life due to the absence of bulky hardware configuration parameters, including power supply and
electrolytic capacitors; LC filter values.
5) smaller and lighter design than other regeneration invert- Subsequently, another compensation algorithm is suggested
ers with equivalent power ratings. to overcome the drawbacks presented in the first compensation
algorithm. This compensation algorithm is based on a
Manuscript received July 31, 2009; revised November 28, 2009; accepted proportional–integral (PI) controller usage for power factor
January 5, 2010. Date of publication March 8, 2010; date of current version
December 10, 2010. This work was supported in part by the University control. Aside from the flexible adjustment capability of the
of Ulsan, by Ulsan Metropolitan City, and by the Ministry of Knowledge power factor that this compensation algorithm can provide,
Economy of the Korean Government through the Network-based Automation its performance is independent on the MC’s hardware
Research Center at the University of Ulsan.
The authors are with the School of Electrical Engineering, University of configuration.
Ulsan, Ulsan 680-749, Korea (e-mail: nmhoang@mail.ulsan.ac.kr; hhlee@ System responses in both transient and steady-state oper-
mail.ulsan.ac.kr; twchun@mail.ulsan.ac.kr). ations are presented in this paper. Comparison between the
Color versions of one or more of the figures in this paper are available online
at http://ieeexplore.ieee.org. two IPF compensation algorithms is also analyzed and summa-
Digital Object Identifier 10.1109/TIE.2010.2044736 rized in details. Simulation studies and experiments using the

0278-0046/$26.00 © 2011 IEEE

NGUYEN et al.: INPUT POWER FACTOR COMPENSATION ALGORITHMS USING A NEW DSVM METHOD FOR MC 233

Fig. 1. MC equivalent circuit of a single-phase input filter.

proposed theories on a three-phase inductive load (RL) model

are conducted. The results obtained are presented illustrating Fig. 2. Structure of a three-phase ac–ac MC.
better performances of the proposed compensation algorithms
using the new DSVM method. Furthermore, experimental re- displacement angle:
sults on a 5-hp induction motor control, applying the power  
ωCVs 1
factor compensation algorithm based on a PI controller, are δ = ϕis /ii = tan−1 . . (8)
(1 − ω 2 LC) Is
given to validate the feasibility and effectiveness of the pro-
posed compensation algorithms for dynamic load conditions. From (8), the displacement angle depends on three parame-
ters: the L and C values of the input filter and the fundamental
input current amplitude Is , which depends on the output load
II. I NPUT F ILTER A NALYSIS OF MC conditions of the MC. In order to minimize this displacement
The input filter is designed to filter high harmonic compo- angle, small LC values are used in the practical MC system.
nents of the input current and to reduce the input voltage dis- However, with small LC values, the main power supply current
tortion supplied to the MC. The following are basic equations contains higher harmonic components√resulting from the well-
regarding input and output voltages, and currents of the MC: known higher cutoff frequency 1/(2π LC).
To achieve a maximum IPF at the main power supply, the
va = Vm cos(ωt) SVM method, which guarantees the input current and input
vb = Vm cos(ωt − 2π/3) (1) voltage of the MC to be in phase, must be revised to compensate
for the displacement angle caused by the input filter.
vc = Vm cos(ωt − 4π/3)
→
−vi = 2(va + vb ej2π/3 + vc ej4π/3 )/3 = Vi ejαi (2)
III. N EW DSVM M ETHOD
vo = 2(vA + vB ej2π/3 + vC ej4π/3 )/3 = Vo ejαo (3)
ii = 2(ia + ib ej2π/3 + ic ej4π/3 )/3 = Ii ejβi (4) A. Space Vectors of MC
io = 2(iA + iB ej2π/3 + iC ej4π/3 )/3 = Io ejβo . (5) The three-phase MC module includes nine bidirectional
switches, as shown in Fig. 2. There are 27 possible switching
From Fig. 1, the following equations are obtained: configuration (SC) states. However, only 21 SCs of them can
be used to implement the modern control algorithms for the
vi = vs − L(dis /dt) MC such as the SVM and direct torque control methods.
if c = C(dvi /dt) As shown in Table I, the following are observed.
is = if c + ii . (6) 1) Group I (±1, ±2, . . . , ±9) consists of the SCs which have
two output phases connected to the same input phase.
Considering the power factor on the power supply side, the 2) Group II (0a , 0b , 0c ) consists of the SCs which have all
power supply frequency in (1) will be taken as the fundamental output phases connected to a common input phase. For
frequency. The MC control algorithm guarantees the load con- each SC, the corresponding line-to-neutral voltage vector
trol requirements and unity power factor at the power supply and input line current vector have fixed directions as
side as well. Then, (6) can be rewritten as follows: represented in Fig. 3.
3) Group III consists of six other SCs which have the output
Vs = Vs ej0 phases connected to different input phases. The output
Vi = Vs − jωLIs voltage vector and input current vector have variable
 −1 ωLIs
= Vs2 + (ωLIs )2 e−j tan ( Vs ) directions and can rarely be used.
Ii = (1 − ω 2 LC)Is − jωCVs
  ωCV 
2 −j tan−1 s
B. New DSVM Method for MC
= [(1 − ω LC)Is ] + (ωCVs ) e
2 2 (1−ω 2 LC)Is
.
(7) For the sake of explaining the new DSVM method, we
assume both the desired output voltage and the input voltage
Finally, the input current of the power supply always space vectors to be located in sector 1 without missing the
leads the input current of the MC with the following generality of the analysis (0 ≤ αo ≤ π/3 and kv = 1, −π/6 ≤
234 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 58, NO. 1, JANUARY 2011

TABLE I By similar analysis, the SCs selected to obtain vo are +1 (d3 )
P OSSIBLE SC S AND V ECTORS U SED IN THE MC
and −3 (d4 ) with the following duty ratios:
   
2q sin kv π3 − αo sin π6 − αi − δ − (ki − 1) π3
d3 = √
3 cos(δ)
(12)
 π  π 
2q sin kv 3 − αo sin 6 + αi − δ − (ki − 1) 3
π
d4 = √ .
3 cos(δ)
(13)

Finally, zero SC (d5 ) is applied to complete the sampling

period

d5 = 1 − (d1 + d2 + d3 + d4 )
(2kv −1)π (ki −1)π
2q cos αo − 6 cos αi − δ − 3
= √ (14)
3 cos(δ)

where d1 , d2 , d3 , d4 , and d5 are the duty ratios of four

active SCs (in this case, −7, +9, +1, and −3) and zero SC,
respectively, and q = vo /vm is the voltage transfer ratio.
Table II shows all switching patterns that can be used in the
DSVM method if the desired input current vector is located in
the same sector with the input voltage vector.
Case 2: The displacement angle places the desired input
current vector and input voltage vector in different sectors.
Without missing the generality of the analysis, we assume that
the input voltage vector is located in sector 1 (−π/6 ≤ αi ≤
αi ≤ π/6 and ki = 1), as shown in Fig. 3, where kv and ki are π/6) and that the desired input current vector is located in
the output voltage vector sector and the input voltage vector sector 6 (−π/2 ≤ βi ≤ −π/6) with the different phase angle
sector, respectively. δ shown in Fig. 4.
Case 1: The displacement angle places the desired input Similar to Case 1, to generate the desired voltage vector vo
current vector and the input voltage vector in the same sector and to maintain the desired different phase angle δ between the
(−π/6 ≤ βi ≤ π/6) in Fig. 3(b). The desired output voltage input current vector and the input voltage vector, the suitable
vector vo is generated from two vectors vo and vo . To match SCs selected are −7, +8, +1, −2, and zero SCs with the
the vector direction as vo , among the six possible SCs (±7, ±8, following duty ratios:
±9) that have the output voltage vector in the same direction of    
2q sin αo −(kv −1) π3 sin π6 + αi −δ−(ki −2) π3
vo , only two higher voltage magnitude vectors are considered to d1 = √
generate which must maintain the input current vector direction 3 cos(δ)
ii to be inside sector 1 and lag behind the input voltage vector (15)
with a certain angle δ = αi − βi . In order to approach the given   π 
conditions, SCs −7 and +9 are selected to drive the MC, and 2q sin αo −(kv −1) 3 sin 6 − αi −δ−(ki −2) 3
π π
d2 = √
from Fig. 3, the duty ratios of SCs −7 (d1 ) and +9 (d2 ) should 3 cos(δ)
satisfy the following relationship: (16)
    
d1
= sin πd2+β 2q sin kv π3 −αo sin π6 + αi −δ−(ki −2) π3
6 −βi )
sin( π ( 6 i) (9) d3 = √ (17)
d1 (−2vab /3) + d2 (2vca /3) = vo . 3 cos(δ)
 π   
Solving (9) under the given condition δ = αi − βi , the duty 2q sin kv 3 −αo sin π6 − αi −δ−(ki −2) π3
d4 = √ (18)
ratios for SCs −7 and +9 are respectively as follows: 3 cos(δ)
   
2q sin αo −(kv −1) π3 sin π6 − αi −δ−(ki −1) π3 2q cos αo − (2kv6−1)π cos αi −δ− (ki−2)π
d1 = √ 3
3 cos(δ) d5 =1− √ (19)
3 cos(δ)
  π (10)
2q sin αo −(kv −1) 3 sin 6 + αi −δ−(ki −1) 3
π π
where d1 , d2 , d3 , d4 , and d5 are the duty ratios of four
d2 = √ .
3 cos(δ) active SCs (in this case, −7, +8, +1, and −2) and zero SC,
(11) respectively.
NGUYEN et al.: INPUT POWER FACTOR COMPENSATION ALGORITHMS USING A NEW DSVM METHOD FOR MC 235

Fig. 3. (a) Output line-to-neutral voltage vector. (b) Input line current vector.

TABLE II
S WITCHING PATTERNS FOR THE I NPUT VOLTAGE AND C URRENT V ECTORS L OCATED IN THE S AME S ECTOR

Fig. 4. (a) Output line-to-neutral voltage vector. (b) Input current and voltage vectors at different sectors.

Table III shows all possible switching patterns if the desired As seen from (20), corresponding to each voltage transfer
input current vector and input voltage vector are located in ratio q, there exists a possible maximum compensated displace-
different sectors. ment angle between the desired input current vector and the
input voltage vector. However, the new DSVM method is only
C. Maximum Compensated Angle validated if the input voltage vector leads the input current
vector to one sector, i.e., δ ≤ π/3. The maximum compensated
The duty ratio of the zero SC has to be positive to validate angle is given by
the DSVM method. Considering the fact that d5 ≥ 0, we obtain
(20) from (14) and (19)   √ √

√ cos−1 2q
√
3
, 4
3
≤q≤ 2
3
δmax = √ (21)
3 cos(δ)
q≤ . (20) π/3, 0<q≤ 3
4 .
2
From
√ (20), the maximum voltage transfer ratio is inferred to Fig. 5 shows the relationship between the maximum com-
be 3 cos(δ)/2,
√ and the well-known maximum voltage transfer pensated angle and the voltage transfer ratio representing the
ratio becomes 3/2 at δ = 0. reference output voltage of the load demands.
236 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 58, NO. 1, JANUARY 2011

TABLE III
S WITCHING PATTERNS FOR THE I NPUT VOLTAGE AND C URRENT V ECTORS L OCATED IN THE D IFFERENT S ECTORS

Fig. 6. Block diagram of compensation algorithm I based on the optimal

compensated angle.
Fig. 5. Maximum compensated angle versus voltage transfer ratio.

IV. IPF C OMPENSATION A LGORITHMS

U SING THE N EW DSVM M ETHOD
A. Compensation Algorithm Based on the
Optimal Compensated Angle
From the filter characteristics explained in Section II, the
main input current variation directly impacts the displacement
angle between the main input phase voltage and the main input
current.
To overcome this effect, a power factor compensation algo-
rithm based on the optimal compensated angle (compensation Fig. 7. Block diagram of compensation algorithm II based on using a PI
algorithm I) is proposed. The optimal compensated angle is controller.
calculated in (8); its value depends on the main input current B. Compensation Algorithm Based on a PI Controller
and main input voltage magnitudes, fundamental frequency,
and LC values. To overcome the drawbacks in compensation algorithm I,
Fig. 6 shows the proposed online block diagram. The com- another IPF compensation algorithm using a PI controller is
pensated displacement angle is periodically updated at each proposed: compensation algorithm II.
compensating period Tpf (Tpf  Ts ), where Ts is the PWM In compensation algorithm II, a sine value of the displace-
sampling period. ment angle ψi between the input line-to-neutral voltage vector
The advantages of compensation algorithm I are as follows. and the corresponding input line current vector is chosen to
control the IPF in Fig. 7. The unity power factor at the power
1) It improves the main power factor produced by the new supply side of the MC is intrinsically satisfied if the value of
DSVM method without changing the output control. sin(ψi ) is maintained close to zero.
2) It reduces the main input current magnitude due to a The proposed PI controller is noted by
smaller reactive power supply.  
Ki
In compensation algorithm I, the estimation of the compen- δcomp = Kp + Δe (22)
s
sated angle is dependent on many parameters that impact on the
accuracy of the optimal compensated angle estimation. where Δe = sin(ψi )ref − sin(ψi )est .
NGUYEN et al.: INPUT POWER FACTOR COMPENSATION ALGORITHMS USING A NEW DSVM METHOD FOR MC 237

Fig. 8. Input/output waveforms of the MC at reference output voltage

q = 0.7, fo = 70 Hz with the conventional DSVM method (pf = 0.912).

Fig. 10. Input waveforms of the MC at reference output voltage q = 0.4,

fo = 40 Hz. (a) Conventional DSVM method (pf = 0.589). (b) Compensa-
tion algorithm I (pf = 1.0). (c) Compensation algorithm II (pf = 1.0).

Fig. 9. Input/output waveforms of the MC at reference output voltage

q = 0.7, fo = 70 Hz with compensation algorithm II (pf = 1.0).

The limiter of the PI controller is always updated with

the new maximum compensated angle δmax from (21). The
compensated angle is within [0, δmax ], and it will be inserted
as the desired displacement angle between the input current
and the voltage in the new DSVM method. In the case of the
IPF compensation, the stable response is more important than
the fast response. Thus, PI gains selected for the proposed
algorithm are recommended: The kp gain is low, and the ki
gain is not so large to avoid large overshoot and to have stable Fig. 11. Input/output waveforms of the MC at reference output voltage
performance. Under these constraints, the PI gains are selected q = 0.25, fo = 25 Hz with the new DSVM method (pf = 0.445).
based on the computer simulation in Section V.
With this proposed algorithm, the compensating period re- The input filter was designed with L = 1.4 mH, C =
quired to maintain a high power factor is as the same as the 22.5 μF, and Y connection. The PWM frequency was 10 kHz,
PWM sampling period (Tpf = Ts ). As the output changes, the and simulation results were obtained under a constant V /f
PI still functions well in terms of the steady-state and dynamic condition.
performance achievement of the input side. Furthermore, this Fig. 8 shows the simulation results of the conventional
proposed algorithm is independent on the input filter and power DSVM method at reference output voltage q = 0.7, fo =
supply parameters, which are quite sensitive during practical 70 Hz. The input current of the MC is in phase with the
operations. input line-to-neutral voltage. However, the main power supply
current lags behind the input line-to-neutral voltage due to
V. S IMULATION R ESULTS AND D ISCUSSIONS the input low-pass filter, as explained in Section II. The IPF
obtained here is only 0.912.
Simulation was carried out on a three-phase RL load using In order to determine the differences between the conven-
PSIM 6.0 software. The simulation parameters for the RL load tional DSVM method and our new DSVM method, the same
were as follows: reference output voltage was applied using compensation al-
1) power supply (line-to-neutral voltage): 100 V/60 Hz; gorithm II shown in Fig. 9. Due to the optimal compensated
2) three-phase RL load: 26 Ω, 12 mH. angle achieved by the PI controller, the main input current at the
238 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 58, NO. 1, JANUARY 2011

Fig. 12. IPFs and compensated angles under constant V /f . (a) IPF. (b) Compensated angle.

Fig. 13. Waveforms of the filter and MC at reference output voltage

q = 0.15, fo = 15 Hz with the maximum compensated angle in compensation
algorithm II.

Fig. 15. Input/output waveforms of the MC during the compensation

process at reference output voltage q = 0.7, fo = 70 Hz. (a) Compensation
algorithm I. (b) Compensation algorithm II.
Fig. 14. IPF and input/output waveforms of the MC as the load changes from
q = 0.4, fo = 40 Hz to q = 0.7, fo = 70 Hz.

power supply is in phase with the line-to-neutral input voltage, than the maximum allowable compensated angles shown in
while the MC satisfies the reference output voltage presented in Fig. 5. Therefore, the desired unity power factor can be
the figure by the output currents and line-to-line output voltage. obtained.
Fig. 10(a)–(c) shows the steady-state performances at refer- The simulation results of reference output voltage q = 0.25,
ence output voltage q = 0.4, fo = 40 Hz of the conventional fo = 25 Hz are shown in Fig. 11. As plotted in Fig. 5, the
DSVM method, compensation algorithm I, and compensation required optimal compensated angle is higher than the max-
algorithm II, respectively. The input currents and voltage wave- imum compensated angle π/3. Consequently, the maximum
forms in Fig. 10(a) present the IPF at 0.589. The input currents compensated angle is applied to compensate the IPF, and the
and voltage waveforms in Fig. 10(b) and (c) describe the MC highest IPF (0.445) can be obtained, while only 0.27 is obtained
operations with the optimal compensated angle for compensa- with no compensation in the conventional DSVM method.
tion algorithms I and II, respectively. The power factors for both Observing our new DSVM method’s results, we can see that
algorithms are achieved at unity. the magnitude of the input current decreases significantly due
As easily seen in the two cases (q = 0.7, fo = 70 Hz and q = to the minimum reactive power supply utilization resulting from
0.4, fo = 40 Hz), the optimal compensated angles are smaller the proposed IPF compensation algorithms.
NGUYEN et al.: INPUT POWER FACTOR COMPENSATION ALGORITHMS USING A NEW DSVM METHOD FOR MC 239

TABLE IV
C OMPARISON B ETWEEN C OMPENSATION A LGORITHMS I AND II

Fig. 18. Input/output waveforms of the MC at reference output voltage

q = 0.7, fo = 70 Hz with the conventional DSVM method.

Fig. 16. Compensated angle response at reference output voltage q = 0.7,

fo = 70 Hz for different group values of PI gains.

Fig. 19. Input/output waveforms of the MC at reference output voltage q =

0.7, fo = 70 Hz with the proposed compensation algorithm II.

Fig. 20. Steady state of the MC’s input waveforms at reference output voltage
q = 0.4, fo = 40 Hz. (a) Conventional DSVM method. (b) Compensation
algorithm II.

Fig. 17. Block diagram of the MC hardware configuration and laboratory in Fig. 12(a). With the high output voltage region q ≥ 0.35,
MC prototype. the IPFs obtained from compensation algorithms I and II are
significantly increased to almost unity compared to the lower
Comparison of the IPFs obtained from the conventional values obtained from the conventional DSVM method. The
DSVM method and the compensation algorithms using our difference of the optimal compensated angles obtained from
new DSVM method is shown in Fig. 12. With a constant the two compensation algorithms is very small, approximately
V /f for the RL load, the higher output load corresponds to 0.02 rad shown in Fig. 12(b).
the higher voltage transfer ratio (q). The compensated angles For the middle output voltage region 0.2 ≤ q < 0.35, the
at steady state from both compensation algorithms are shown IPF is much higher compared to the one obtained from the
240 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 58, NO. 1, JANUARY 2011

Fig. 21. Input/output waveforms of the MC at reference output voltage

q = 0.25, fo = 25 Hz with the proposed compensation algorithm II.

conventional DSVM, due to using the maximum compensated

angle π/3, to both compensation algorithms.
At very low output voltage region q < 0.2, the output load
currents are getting smaller, and the input currents (ii ) of the
MC synthesized from the output load currents are much smaller
than the filter current (ifc ) shown in Fig. 1. Moreover, the filter
current is the dominant component of the main input current
(is ). In this case, it is not expected for the compensated angle to
significantly improve the main IPF regardless of the maximum
compensated angle obtained from compensation algorithm II.
As explained in Section II, compensation algorithm I is based
on the approach of unity power factor assumption, and com-
pensation algorithm I can only give the higher power factor due
Fig. 22. Input/output waveforms of the MC at reference output voltage
to a certain compensated angle calculated from (8), shown in q = 0.7, fo = 70 Hz during the compensation process. (a) Compensation
Fig. 12(b). For more illustrations, simulation result in Fig. 13 algorithm I. (b) Compensation algorithm II.
shows the waveforms of the filter and MC at reference output
voltage q = 0.15 with the maximum compensated angle in
compensation algorithm II. Because the fundamental waveform
of the MC’s input current (ii_fund ) is so small compared
to the fundamental waveform of the filter capacitor’s current
(ifc_fund ), the main input current (is ≈ ifc_fund + ii_fund ) can-
not have a major effect to improve the IPF.
Fig. 14 shows the input/output waveforms of the MC using
a PI controller under the new DSVM method during the load
change from reference output voltage q = 0.4, fo = 40 Hz
to q = 0.7, fo = 70 Hz. The main input current is controlled
in phase with the line-to-neutral input voltage after a short
transient period, and the IPF is still kept at unity after the load
change condition. This operation clearly illustrates the effective
performance of the proposed algorithm under dynamic load
conditions.
Compensation algorithm I using the compensated angle
Fig. 23. Input/output waveforms of the MC as the load changes from q = 0.4,
calculated from (8) and compensation algorithm II using a fo = 40 Hz to q = 0.7, fo = 70 Hz.
PI controller were both simulated with the reference output
voltage q = 0.7, fo = 70 Hz. The compensation algorithms selected PI gains used in both simulation and experiment are
were applied at 20 ms for two cases, and the simulation results kp = 0.05 and ki = 70.
are shown in Fig. 15(a) and (b), respectively. Comparison
conclusions are summarized in Table IV.
VI. E XPERIMENTAL V ERIFICATION
Fig. 16 shows the compensated angle responses for different
control gains of the PI controller at the reference output voltage To validate the proposed theories and simulation, an ex-
q = 0.7, fo = 70 Hz. As can be seen from Fig. 16, the small perimental test of our new DSVM method was carried out
PI gains are recommended for practical applications, and the using a three-phase power supply 100 V/60 Hz (line-to-neutral
NGUYEN et al.: INPUT POWER FACTOR COMPENSATION ALGORITHMS USING A NEW DSVM METHOD FOR MC 241

Fig. 24. Input/output waveforms of the induction motor operating at speed 1200 r/min and load torque 2 N · m. (a) Conventional DSVM method (pf = 0.804).
(b) Compensation algorithm II (pf = 1.0).

voltage), an input filter (L = 1.4 mH, C = 22.5 μF, and The main input current ias leads the input line-to-neutral volt-
Y connection), and a three-phase RL load (R = 26 Ω and age due to the input filter effect. Fig. 19 shows the experimental
L = 12 mH), which are identical to the simulation parameters. results using compensation algorithm II. Moreover, it is evident
Moreover, the MC module implemented was an FM35R12KE3 that the main input current ias is in phase with the line-to-
insulated-gate bipolar transistor module produced by Eupec. neutral voltage vas , and a unity power factor is obtained. In
Fig. 17 shows the block diagram of the MC hardware config- addition to the power factor improvement, it is observed from
uration and laboratory MC prototype. The system configuration Figs. 18 and 19 that the sinusoidal output current waveforms
consists of two 32-bit DSPs type TMS320F2812 operating remain unchanged in the new DSVM method. Furthermore,
at a clock frequency of 150 MHz, which communicate with Fig. 20(a) and (b) shows the MC’s input waveforms at the refer-
each other through a dual-port RAM (CY7C028). One of ence output voltage q = 0.4, fo = 40 Hz with the conventional
the DSPs (DSP1) is the main controller which executes the and new DSVM methods, respectively. Lastly, the IPF of unity
main control programs, including reading A/D converters, en- obtained by using the new DSVM method is identical to the
coding pulses, processing the MC’s proposed compensation simulation results.
algorithms, and communicating with PC through serial commu- Fig. 21 shows the input/output waveforms of the MC at
nication interface (SCI), and another DSP (DSP2) is dedicated reference output voltage q = 0.25, fo = 25 Hz. The maximum
to PWM signal generation. It receives the final switching pat- compensated angle π/3 was applied, and the simulation results
terns every sampling period from the main controller (DSP1) were verified by experiment.
through a dual-port RAM and subsequently generates space Fig. 22(a) and (b) shows the experimental results during the
vectors and their duty ratios according to the new DSVM compensating transient state of compensation algorithms I and
method. A complex programmable logic device (CPLD) board II, respectively. The output current representing the output load
is implemented using an Altera EPM7218 for four-step com- remains unchanged during the compensation process.
mutation and logical protection circuits and safe operation Fig. 23 shows the MC dynamic performance as the load
modes: The CPLD receives the space vector patterns changed changes, which is represented by changing the reference output
from the DSP2 and the output current directions which are voltage from q = 0.4, fo = 40 Hz to q = 0.7, fo = 70 Hz.
used for the well-known four-step current commutation [16]; The input currents are kept in phase with the line-to-neutral
the protection signal from the overvoltage and overcurrent voltage as the load changes. This is a very important operation
protection circuits will input the CPLD for safe operation. as applied to the motor control area.
Furthermore, the CPLD can receive the command from the Our new DSVM method was applied to induction motor con-
main controller (DSP1) for safe, starting-up, turning-off, and trol to validate the effectiveness of the compensation algorithm,
running modes of the MC system. An analog board uses a 12-b and in this case, compensation algorithm II was examined
four-channel A/D converter (AD7864AS) to measure input in the experiment. The indirect field-oriented control method
currents and voltages for higher accuracy; the load currents are in combination with the new DSVM method for MC was
measured by internal 12-b A/D channels of the DSP1. All con- implemented on the same MC hardware design to control a
trol modes and commands of the main controller are monitored 5-hp induction motor. The motor parameters are given in the
by GUI in PC through an SCI standard, and data can be viewed Appendix.
by either a PC through a controller–area network or an oscil- The experiment was carried out at a rotor speed of 1200 r/min
loscope. The new DSVM strategies were operated at a PWM and a load torque of 2 N · m in Fig. 24. With the same load
frequency of 10 kHz with a double-side switching pattern. conditions, which are represented by motor speed and stator
The experimental results shown in Fig. 18 are the input/ current waveforms, the new DSVM method [Fig. 24(b)] can
output waveforms of the MC at the reference output voltage improve the IPF to be unity as compared to 0.804, obtained by
q = 0.7, fo = 70 Hz using the conventional DSVM method. the conventional DSVM method [Fig. 24(a)].
242 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 58, NO. 1, JANUARY 2011

Fig. 25. Input/output waveforms of the MC as the induction motor speed changes from 1000 to 500 r/min at load torque 3 N · m.

Fig. 25 shows the experimental results of the induction motor Mutual inductance 153 mH.
operation when the rotor speed is gradually changed from 1000 Number of pole pair 2.
to 500 r/min at a constant load torque of 3 N · m. The proposed Rated speed 1730 r/min.
compensation algorithm guarantees the unity power factor at
the power supply side throughout dynamic performances. R EFERENCES
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NGUYEN et al.: INPUT POWER FACTOR COMPENSATION ALGORITHMS USING A NEW DSVM METHOD FOR MC 243

[13] S. F. Pinto and J. F. Silva, “Direct control method for matrix converters Hong-Hee Lee (S’88–M’91) received the B.S.,
with input power factor regulation,” in Proc. 35th IEEE Power Electron. M.S., and Ph.D. degrees in electrical engineering
Spec. Conf., Jun. 2004, vol. 3, pp. 2366–2372. from Seoul National University, Seoul, Korea, in
[14] K. Yamada, T. Higuchi, E. Yamamoto, H. Hara, T. Sawa, M. M. Swamy, 1980, 1982, and 1990, respectively.
and T. Kume, “Filtering techniques for matrix converters to achieve en- From 1994 to 1995, he was a Visiting Professor
vironmentally harmonics drives,” in Proc. 11th EPE, Dresden, Germany, with Texas A&M University, College Station. Since
2005. 1985, he has been with the Department of Electri-
[15] H.-H. Lee, H. M. Nguyen, and T.-W. Chun, “New direct-SVM method for cal Engineering, University of Ulsan, Ulsan, Korea,
matrix converter with main input power factor compensation,” in Proc. where he is currently a Professor with the School of
34th IEEE IECON, Nov. 10–13, 2008, pp. 1281–1286. Electrical Engineering. He is also the Director of the
[16] N. Burany, “Safe control of four quadrant switches,” in Conf. Rec. IEEE Network-based Automation Research Center, which
IAS Annu. Meeting, 1989, pp. 1190–1194. is sponsored by the Ministry of Knowledge Economy. His research interests are
power electronics, network-based motor control, and control networks.
Dr. Lee is a member of the Korean Institute of Electrical Engineers, the
Korean Institute of Power Electronics, and the Institute of Control, Robotics
and Systems.

Hoang M. Nguyen was born in Nha Trang, Vietnam, Tae-Won Chun was born in Korea in 1959. He
in 1979. He received the B.S. degree in electrical received the B.S. degree in electrical engineering
engineering from Ho Chi Minh City University of from Pusan National University, Busan, Korea, in
Technology, Ho Chi Minh City, Vietnam, in 2002 1981 and the M.S. and Ph.D. degrees in electrical
and the M.S. and Ph.D. degrees from the Univer- engineering from Seoul National University, Seoul,
sity of Ulsan, Ulsan, Korea, in 2005 and 2010, Korea, in 1983 and 1987, respectively.
respectively. Since 1986, he has been a Member of the Faculty
He is currently with the School of Electrical En- of the Department of Electrical Engineering, Univer-
gineering, University of Ulsan. His research interests sity of Ulsan, Ulsan, Korea, where he is currently a
are electrical drives, industrial networks, renewable Full Professor. His current research interests are con-
energy, and modern power converters, particularly trol of electrical machines, power converter circuits,
matrix converters. and industrial applications.