Hindawi

International Journal of Antennas and Propagation

Volume 2021, Article ID 3874794, 10 pages
https://doi.org/10.1155/2021/3874794

Research Article
Design and Impedance Modeling of a Compact Sequentially
Rotated Quasi-Lumped Antenna Array for Wi-Fi Applications

Yazeed M. I. Qasaymeh
Department of Electrical Engineering, College of Engineering, Majmaah University, Al-Majmaah 11952, Saudi Arabia

Correspondence should be addressed to Yazeed M. I. Qasaymeh; yazed100@gmail.com

Received 29 June 2021; Revised 28 August 2021; Accepted 1 September 2021; Published 18 September 2021

Academic Editor: Mohammad Alibakhshikenari

Copyright © 2021 Yazeed M. I. Qasaymeh. This is an open access article distributed under the Creative Commons Attribution
License, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is
properly cited.
In this study, a miniaturized 2 × 2 sequentially rotated (SR), circularly polarized (CP), and quasi-lumped antenna array that
resonates in the IEEE 802.11a band is introduced. The shorting pins technique is implemented to achieve circular radiation
patterns, and the resonating elements are excited using a SR quarter-wavelength feeding network. A resonance model of the four
radiating elements array is postulated to give a physical insight into the relative dimensions and to allow for a study of the
resonance characteristics and the effect of the shorting pins. An antenna model is simulated, fabricated, and measured to
authenticate this arrangement, giving results of |S11| < −10 dB and axial ratio (AR) < 3 dB for bandwidths of 3.85%
(5.645–5.867 GHz) and 1.54% (5.77–5.86 GHz) for right-hand circular polarization (RHCP). The size of the antenna array
structure is 0.696λ0 × 0.599λ0 × 0.0157λ0 at a center frequency of 5.8 GHz.

1. Introduction subject to a narrow bandwidth and phase differences due

to the differences in line lengths [7]. In contrast, a parallel-
Shortly after their introduction in the 1970s, microstrip fed topology gives a wider bandwidth, as it does not
patch antennas attracted considerable attention due to their produce high mismatch losses [8]. In addition, corporate
useful features, such as ease of analysis and fabrication, and parallel feeds have the advantage that the excitation
desirable radiation characteristics, low mass, cost efficiency, for such arrays has good design elasticity and ease of
straightforward integration, minimal profile, mechanical integration when two-dimensional arrays are created. The
robustness, and compatibility with MIMC designs [1, 2]. feed magnitude for each element can be easily controlled
However, they were considered to be impractical for mobile by employing either amplifiers or attenuators, and the
communications, particularly, for installation in mobile phase can be controlled using phase shifters.
handsets because of their large size. Thus, printed microwave Numerous single-feed CP microstrip antenna methods
quasi-lumped antennas appeared to be the most suitable have been documented in the literature, such as a U-slot
contenders to replace classical patch antennas, as they were microstrip antenna [9], an equilateral triangular single-feed
not subject to size limitations [3, 4]. circularly polarized microstrip antenna [10], a CP annular-
Single-element antennas are normally low-gain de- ring microstrip antenna developed using a pair of inserted
vices that need to be arrayed to achieve high gain per- slits [11], and a square-ring microstrip antenna with a
formance. Several types of feed have been used in linear truncated corner design [12]. Other designs have been based
arrays to meet this requirement, including series [5] and on the carving of a split ring resonator on a patch [13],
parallel microstrip feeds [6]. In general, a series feed etching of asymmetrical slots into a single probe-feed square
topology results in a more concise network that involves microstrip [14], the use of an L-shaped microstrip connected
shorter transmission lengths, fewer junctions, and lower to a tag-chip and terminated by a shorting pin coupled to a
insertion losses compared to a parallel feed; however, it is microstrip antenna [15], and the use of a circular eccentric
2 International Journal of Antennas and Propagation

ring to generate CP radiation [16]. The aforementioned yielding a wide axial ratio band. Other techniques for slicing
methods aimed to achieve CP by altering the physical di- slots or adding tails are less beneficial in terms of size mitigation
mensions of the antenna. compared to the use of shorting pins, which is the most ef-
In this paper, a quasi-lumped resonator antenna array is fective solution for the size miniaturization of patch antennas.
presented with the aim of achieving size reduction and The impact of the shorting pins depends on several parameters,
circular radiation performance. In conventional approaches, such as the number and radii of the posts and the thickness of
if the single element is small, the array is also small. In view the microstrip antenna. The shorting pins are directly con-
of this, a corporate feed design for a quasi-lumped array nected between the radiating element and the ground plane.
antenna network will be implemented to reduce the foot-
print of the array configuration.
The three major objectives of this study are (i) to provide 2.3. Feeding Topology. Single feeds for CP radiation opera-
an array equivalent impedance circuit used to give a fre- tion can be categorized into two types: single port/single feed
quency resonance preliminary indication, (ii) to generate CP and single port/double feed. The single-port feed technique
radiation, and (iii) to design a miniaturized array structure is easy to construct [20]. SR arrays were first proposed in [21]
with the smallest possible size. The antenna array equivalent to improve the performance of CP arrays. Microstrip an-
circuit is built based on array feed and elements’ dimensions. tenna arrays use SR to achieve improvements in polarization
CP is achieved using the technique of loading of shorting purity, impedance matching, and pattern symmetry across
pins, while a quarter-wavelength transformer technique is wider bandwidths [22, 23].
used to feed the antenna array to improve the gain and In general, corporate feed networks are used to provide
reduce the overall array size. power splits of two-element numbers. The quarter-wave-
In Section 2, the proposed antenna array excitation length approach is a technique that is used to match the
technique and feeding method are introduced. The im- transmission line to the antenna using a transformer. The
pedance resonance model of the array is described in Section impedance characteristics of the transmission line are re-
3. A discussion and the experimental results from the stricted in terms of the range and values; thus, the quarter-
proposed array are presented in Section 4. wavelength transformer technique is most suitable when
microstrip transmission lines are used [24, 25].
The proposed design is shown in Figure 3. Our array
2. Configuration of the Antenna Array consists of four radiating quasi-lumped elements fed by a
2.1. Quasi-Lumped Geometry. The components required for 2 × 2 SR microstrip in the front layer and a ground plane in
the approximate microwave operation of lumped elements in the bottom layer. The shorting pins are located at the center
microstrip structures are microstrip shorts and stubs with of the resonating elements. A quarter-wave impedance
dimensions shorter than a quarter wavelength at the resonating transformer of 70.7 Ω is used to convert a 100 Ω for each
frequency. These are referred to as quasi-lumped. A primary radiating element fed by a 50 Ω line, thus creating the
benefit of this method is the simplicity of transition into an conditions for impedance matching.
equivalent circuit, wherein capacitors and inductors constitute
a microstrip structure. Figure 1 shows the structure of a single 3. Array Equivalent Impedance Model
quasi-lumped element resonator. The equivalent lumped ele-
ments of the single proposed antenna are shown in Figure 2. A In this study, an equivalent impedance model is proposed in
winding-line inductor can be used to increase the capacitance. order to give a physical description of the resonance fre-
The inductor L is molded as a finger that is connected across the quency of the array. This model is used to validate the
capacitor C. The pad capacitors CP1 and CP2 connected to both simulated and measured resonance operations. The antenna
sides of the structure act as capacitors to the ground and can be array consists of a quarter-wavelength microstrip feed and
adjusted in order to tune the resonant frequency of the pro- resonating elements. The quarter-wave microstrip feed is
posed resonator. The details of the process of calculating the developed by cascading three segments with characteristic
operating frequency of the proposed single quasi-lumped impedance values of 50, 70.7, and 100 Ω, as shown in Fig-
antenna and its dimensions are discussed in [17]. Table 1 ure 3. The microstrip circuit consists of the equivalent ra-
presents the dimensions of a single quasi-lumped element at diation conductance Grm and the susceptance of the fringing
a resonant frequency of 5.8 GHz. field capacitance of the microstrip CL. The conductance Grm
and capacitance are calculated using (1 and (2), respectively,
as reported in [26]:
2.2. Shorting Pins. The use of shorting pins was first intro-
duced in [18] to achieve CP radiation. If the shorting pins are 160π2 h2
Grm � , (1)
situated near to the edges of a diagonal-fed patch, two resonant Z2cm λ20 εcm
modes are excited, one of which is higher than the other, and
hence, CP is achieved [19]. The polarization of microstrip where h is substrate height, Zcm is characteristic impedance
antennas can be tuned by adjusting the positions of the of the microstrip, and εcm is effective dielectric constant.
√���
shorting posts (pins) within the boundaries of the antenna. leq C εcm
Inserting the shorting posts into the antenna patch changes the Cl � , (2)
Zcm
smallest point of minor frequency to a higher frequency,
International Journal of Antennas and Propagation 3

w Interdigital Capacitor

Pad
Capacitor
Cp2

WL
L

Inductor L
Pad
Capacitor
Cp1
Y
Wc
Ge X
IL CL Z
Interdigital Capacitor
(a) (b)

Figure 1: Proposed quasi-lumped antenna: (a) dimensions used to calculate the lumped elements; (b) allocation of the lumped elements.

L
CP1 CP2

Figure 2: Equivalent circuit for the proposed antenna.

where leq is equivalent extralength of microstrip and C is The equivalent circuit for the shorting posts is repre-
velocity of light. sented in the form of shunt inductances in the locations at
The quasi-lumped equivalent lumped elements C, L, which the posts are connected to the ground. The inductive
CP1, and CP2 are derived from equations (3–5), respectively. reactance of the posts and the feed probe are calculated from
Table 2 presents the equivalent lumped elements for a single
377 2πh
quasi-lumped element at 5.8 GHz obtained using the XL � √�� tan􏼠 􏼡. (6)
equations reported in [27–29]: εr λ0

εr + 1 Figures 4(a) and 4(b) show the equivalent circuits for a

C � ε0 􏼒 􏼓􏼂(N − Δ)CL 􏼃, (3) single quasi-lumped array element with a shorting post and
2
with a single segment microstrip feed, respectively. The
2IL W impedance of the microstrip feeder segments and the four
L � 200 × 10− 9 IL 􏼢ln􏼠 􏼡 + 0.50049 + L 􏼣, (4) radiating elements are connected to form the array equiv-
WL + h 3IL
alent resonant circuit, which is used to study the resonance
behavior of the array.
⎢
⎡
⎢ 2.85εeff ⎤⎥⎥
CP � ⎢
⎢
⎢
⎢
⎣ 􏽱������������ ⎥⎥⎥⎥⎦
2 2
ln􏼔1 +(1/2) 8h/εeff 􏼁􏼔 8h/εeff 􏼁 + 8h/εeff 􏼁 + π 􏼕􏼕 4. Results and Discussion
1 Figure 5 shows the equivalent input impedance, modeled
× ,
25.4 × 10− 3 using Agilent Advanced Design System (ADS). The resis-
(5) tance and capacitance values representing the microstrip
segment feeder and the four resonating quasi-lumped ele-
where h is substrate thickness, IL is inductor length, wL is ments are given in Tables 2 and 3. The impedance model is
inductor width, N is the number of fingers, and Δ is the used to predict the resonance behavior of the antenna array
correction factor between 0.44 and 0.65 for N � 1 and 2, before modeling and fabrication. No radiation pattern is
respectively. expected from this model.
4 International Journal of Antennas and Propagation

Table 1: Parameters for the elements of the proposed quasi-lumped antenna.

Parameter Dimensions (mm)
We 0.35
IL 3.35
C 3.05
N 8
IL / 1.23
ge 1.23
WL 1.2
L 5.4
W 5.8
h 0813

Quasi Lumped Resonator

50 Ω Shorting 50 Ω
Pin
Shorting Pin

λ/4 λ/4 50 Ω
50 Ω 100 Ω
70.7 Ω λ/4 λ/4 70.7 Ω
70.7 Ω 70.7 Ω

50 Ω Shorting 50 Ω
Pin
Y

Shorting
50 Ω Pin
X
Feeding Network
Z
Figure 3: Proposed SR quarter-wavelength quasi-lumped antenna array.

Table 2: Equivalent lumped elements for the quasi-lumped resonator.

Value Parameter
C 0.347 PF
CP1 0.17 PF
CP2 0.17 PF
L 1.47 nH

Lshorting pin CL Grm

L CP2
CP1

(a) (b)

Figure 4: Equivalent circuits for (a) a single quasi-lumped element with a shorting pin at the center and (b) a microstrip feed.
International Journal of Antennas and Propagation 5

Figure 5: Equivalent impedance model for the antenna array, modeled in ADS.

Table 3: Equivalent lumped elements for the microstrip feeder segments.

Grm (Ω) CL [F] Characteristic impedance (Ω)
− 13
0.0682 4.3 × 10 100
0.0171 8.62 × 10− 13 50
0.0341 6.0967 × 10− 13 70.7

Figure 6 shows the fabricated quasi-lumped antenna increased, the resonance shifts to a higher frequency. When
array. The overall dimensions of the antenna array are the diameter of the shorting pin is 2 mm, the optimum
31 × 36 mm2. A single-port array consists of 2 × 2 SR quasi- return loss is reached at −24.94 dB.
lumped elements in the top layer and a ground plane in the Figure 10 presents the simulated and measured results
bottom layer. In the quarter-wave transformer method, an for the input return loss. The return loss calculated using
SR is used to feed the single-element resonators on a ADS modeling occurs at 5.8 GHz, with a minimal value of
RO4003 C microwave substrate with a relative permittivity −20.81 dB. Based on this result, we were able to validate the
of 3.38 and a thickness of 0.813 mm. The width and length of intended design before prototyping was carried out and
the quarter-wavelength transformer are 1.036 and 7.03 mm, measurements were taken. At this stage, various design
respectively, and the 100 Ω and 50 Ω impedance widths are parameters were not considered, such as the surrounding
equal to 0.46 mm and 1.898 mm, respectively. The reso- boundary of the antenna, soldering, and mismatches, and
nating elements are spaced at half wavelengths, and the main this can explain the slight difference between the modeled
purpose of the proposed antenna is to excite the elements results and the simulated and measured ones. The simulated
and generate a circular radiation output, therefore imple- return loss exhibited a bandwidth of 201 MHz in the range
menting the shorting pins loading technique. The shorting 5.66–5.861 GHz, with a minimal loss of 35.5 dB at 5.81 GHz.
pins are located at the center of the quasi-lumped element. The measured return loss exhibited a bandwidth of 222 MHz
Figure 7 shows the power flow simulated by the CST in the range 5.645–5.867 GHz, with a minimal loss of
software over the proposed antenna. The power flow from 35.68 dB at 5.814 GHz. We can therefore say that there is a
the feeding port over the quarter-wave feeding network to good agreement between the simulated and measured re-
the radiating quasi-lumped elements reaches a maximum at sults. The divergence in the results is primarily due to small
5.8 GHz, thus ensuring that the maximum power is radiated construction errors and the nonuniformity arising from the
from the quasi-lumped radiators. The magnitude of the manual soldering of the SMA connectors.
current density for an array element at 5.8 GHz is depicted in The simulated and measured E-co-polarization (RHCP)
Figure 8, and it can be observed that the current distribution and E-cross-polarization (LHCP) patterns in the xz plane at
is concentrated within the quasi-lumped fingers. the design frequency of 5.8 GHz are shown in Figure 11(b).
Figure 9 shows the effect of changes in the shorting Here, co-polarization (RHCP) is higher than cross polari-
diameter on S11 for a single resonance quasi-lumped ele- zation (LHCP) by 30 dB in the broadside direction. In
ment, simulated using CST Microwave Studio. It can be seen general, the variance between the simulated and measured
that the proposed resonator operates at 5.8 GHz without the antenna patterns is within 20 dB, and the measured patterns
use of shorting pins. In addition, when the diameter is agree well with the simulated ones. Figures 11(a) and 11(c)
6 International Journal of Antennas and Propagation

Figure 6: Photograph of the fabricated quasi-lumped antenna array.

VA/m2
1.44e+05
1.31e+05
1.18e+05
1.05e+05
91532
78456
65380
52304
39228
26152
13076
0

power (f=5.8) [1] (peak)

Component: Abs
Orientation: Outside
3D Maximum [V/m2]: 12.74e+06
Frequency: 5.8

Figure 7: Power flow in the proposed quasi-lumped antenna array at 5.8 GHz.

show the radiation patterns at the minimum and maximum effects of the SMA connectors. However, the measured and
resonance frequencies. simulated results show reasonable agreement.
Figure 12 shows the ARs for the proposed antenna, Figure 13 shows the simulated and measured gains for
which are determined by measuring the fields in the vertical the resonance bands. The maximum simulated gain is
and horizontal planes. The simulated and measured 3 dB AR around 8 dBi at 5.66 GHz, while the maximum measured
bandwidths for the proposed antenna are 5.79–5.86 GHz gain is 7.48 dBi at 5.66 GHz. This minor difference be-
and 5.77–5.86 GHz, respectively. The deviation between the tween the simulated and measured results can be at-
simulated and measured results may be caused by several tributed to fabrication tolerances and material losses. A
reasons such as the experiment accuracy, the experiment reflector can be used in the bottom plane to enhance the
range, the loss tangent of the substrate, and the parasitic gain performance.
 International Journal of Antennas and Propagation 7

dB (1 A/m)
58.4

54.7

51.1

47.5

43.8

40.2

36.6

32.9

29.3

25.6

22
18.4

surface current (f=5.8) [1] (peak)

3D Maximum [A/M]: 58.38 dB
Frequency: 5.8
Phase: 12

Figure 8: Surface current distribution for an array element at 5.8 GHz.

0
0
–5
–5 –10
Return Loss (dB)
Return Loss (dB)

d –15
–10
–20
–15 –25

–20 –30
–35
–25
–40
5.5 5.6 5.7 5.8 5.9
5.65 5.70 5.75 5.80 5.85 5.90 5.95 Frequency (GHz)
Frequency (GHz)
Simulated
d = 1 mm d = 1.6 mm
Measured
d = 1.2 mm d = 1.8 mm
Impedance Model
d = 1.4 mm d = 2 mm
Figure 10: Simulated and measured reflection coefficients.
Figure 9: Effects of changes in the diameter of the shorting pin
diameter on S11.
8 International Journal of Antennas and Propagation

0 0
10 330 30 10 330 30
0 0
–10 –10
–20 300 60 –20 300 60
–30 –30
–40 –40
–50 –50
270 90 270 90
–50 –50
–40 –40
–30 –30
–20 –20
240 120 240 120
–10 –10
0 0
10 210 150 10 210 150
180 180

E-Co-Simulated E-Co-Measured E-Co-Simulated E-Co-Measured

E-Cross-Simulated E-Cross-Measured E-Cross-Simulated E-Cross-Measured
(a) (b)
0
10 330 30
0
–10
–20 300 60
–30
–40
–50
270 90
–50
–40
–30
–20
240 120
–10
0
10 210 150
180

E-Co-Simulated E-Co-Measured
E-Cross-Simulated E-Cross-Measured
(c)

Figure 11: Simulated and measured E-co-polarized and E-cross-polarized patterns at the xz plane: (a) at 5.7 GHz; (b) at 5.8 GHz; (c) at
5.85 GHz.

6
Axial Ratio (dB)

5.74 5.76 5.78 5.80 5.82 5.84 5.86 5.88 5.90

Frequency (GHz)

Simulated
Measured
Figure 12: Simulated and measured 3 dB ARs.
International Journal of Antennas and Propagation 9

10
8

Gain (dBi)
6
4
2
0
5.64 5.66 5.68 5.70 5.72 5.74 5.76 5.78 5.80 5.82 5.84 5.86 5.88
Frequency (GHz)
Simulated
Measured

Figure 13: Simulated and measured gains.

Table 4: Comparison of the proposed antenna array with several 2 × 2 SR arrays reported in the literature at 5.8 GHz.
References Peak gain (dBi) Bandwidth (%) Cross polarization (dB) Efficiency Size based on wavelength
[23] 10.5 17.39 25 Not available 1.34λ0 × 1.34λ0
[29] 7.46 52 Not available Not available 1.868λ0 × 1.868λ0
[30] 7.25 28.5 15 Not available 0.64λ0 × 0.64λ0
[31] 8.25 29 12 Not available 0.86λ0 × 0.86λ0
Proposed work 7.48 3.85 20 80–84% 0.696λ0 × 0.599λ0

5. Conclusion [3] N. L. Nguyen, “A CPW-fed quasi-PIFA antenna using quasi-

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The author declares no conflicts of interest.
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