A Phase-Locked Loop with Embedded
Analog-to-Digital Converter for Digital Control
Sooho Cha, Chunseok Jeong, and Changsik Yoo
A phase-locked loop (PLL) is described which is
operable from 0.4 GHz to 1.2 GHz. The PLL has basically
the same architecture as the conventional analog PLL
except the locking information is stored as digital code. An
analog-to-digital converter is embedded in the PLL,
converting the analog loop filter output to digital code.
Because the locking information is stored as digital code,
the PLL can be turned off during power-down mode while
avoiding long wake-up time. The PLL implemented in a
0.18 m CMOS process occupies 0.35 mm2 active area.
From a 1.8 V supply, it consumes 59 mW and 984 W
during the normal and power-down modes, respectively.
The measured rms jitter of the output clock is 16.8 ps at
1.2 GHz.
Keywords: Phase-locked loop, digital control, CMOS.
I. Introduction
Phase-locked loops (PLLs) are widely used for various
purposes such as frequency multiplication, clock
synchronization, and zero-delay buffering [1]-[4]. Among
several types of PLL, the charge pump (CP) PLL shown in Fig.
1 is most widely used due to its simple architecture and
superior performance. In a CP PLL, the oscillation frequency
of the voltage controlled oscillator (VCO) is stored as analog
voltage on loop filter output. Therefore, during the powerdown mode, that is, while the output clock of the PLL is not
necessary, it is very difficult to turn the PLL off because it
would take a long time for the PLL to be re-locked due to the
loss of locking information stored as analog voltage. If the PLL
remains running during the power-down mode, the power
consumption would be substantial.
In order to alleviate this problem of the analog CP PLL, an
all-digital PLL might be used where the VCO is substituted by
a digitally controlled oscillator (DCO) and phase error is
detected by a time-to-digital converter (TDC) as shown in Fig.
2 [4]. A digital PLL can be turned off without any
CP
Manuscript received Aug. 10, 2006; revised Mar. 02, 2007.
This work was supported by Hynix Semiconductor. The CAD tools were provided by IDEC.
Sooho Cha (phone: + 82 31 208 7593, email: sooho.cha@samsung.com) was with the
Department of Electronics and Computer Engineering, Hanyang University, Seoul, S. Korea,
and is now with Advanced Technology Development Team, Memory Division, Samsung
Electronics, Hwaseong, Gyeonggi-do, S. Korea.
Chunseok Jeong (email: chunseok.jeong@hynix.com) was with the Department of
Electronics and Computer Engineering, Hanyang University, Seoul, S. Korea, and is now with
DRAM Design Team 4, Hynix Semiconductor Inc., Incheon, S. Korea.
Changsik Yoo (phone: +82 2 2297 3361, email: csyoo@hanyang.ac.kr) is with the
Department of Electronics and Computer Engineering, Hanyang University, Seoul, S. Korea.
ETRI Journal, Volume 29, Number 4, August 2007
UP
ClkRef
PFD
ClkFb
VCO
DN
ClkOut
1/N
Fig. 1. Conventional analog charge-pump PLL.
Sooho Cha et al.
463
ClkRef
TDC
Phase
error
ClkFb
Digital
controller
DCO
code
DCO
ClkOut
TDC: Time-to-digital converter
DCO: Digitally controlled oscillator
ClkRef
ClkFb
Tracking ADC loop
UP
PFD
DN
Comp
1/N
10-bit
digital code
Fig. 2. Conventional all-digital PLL.
concern about the wake-up time because all the locking
information is stored as digital code. The jitter and locking
accuracy (static phase error) of the digital PLL are determined
by the minimum time step provided by the TDC and DCO. To
minimize jitter and static phase error, the delay of the delay
cells must be minimized. However, for a given clock period,
the smaller delay of the delay cells means a larger number of
delay cells because the total delay (sum of delays of all delay
cells) must be equal to the clock period. Therefore, for a
smaller step with a given clock period, a larger silicon area and
higher power consumption are required because a larger
number of delay cells with smaller delay is required.
In this paper, an alternative technique is proposed which
allows an analog PLL to be turned off during power-down
mode without increasing wake-up time [5]. Neither DCO nor
TDC is used. Instead, a tracking analog-to-digital converter
(ADC) is embedded in the feedback loop of the analog CP
PLL. The operation principle of the PLL is described in the
next section, and the experimental results on a 0.4 GHz to 1.2
GHz PLL implemented in a 0.18 m CMOS process follow in
section III. Finally, the conclusion is given in section IV.
II. Analog PLL with Embedded ADC in Feedback
Loop for Digital Control
The proposed analog PLL has basically the same
architecture as the conventional analog CP PLL as shown in
Fig. 3. The difference is that the tracking ADC is embedded in
the feedback loop to convert the analog VCO control voltage to
digital code.
Because the PLL is digitally controlled, there is bang-bang
jitter. The ADC has 10-bit resolution to provide sufficient
locking accuracy and jitter performance. With higher resolution
of the ADC, better jitter performance can be achieved but the
area and power consumption will be larger. The resolution of
the ADC is determined so the bang-bang jitter due to the ADC
quantization is smaller than 20 ps. The ADC comprises a
comparator, counter, and digital-to-analog converter (DAC).
The loop filter output is compared with the output of the 10bit DAC, and the counter counts up or down accordingly. The
464
Sooho Cha et al.
Counter
VCO
1/N
DAC
ClkOut
Fig. 3. Proposed PLL with embedded ADC for digital control.
Power-down
Wake-up
DAC output
Loop filter output
Fig. 4. Effect of floating loop filter output during power-down
mode.
ADC itself is a feedback system; thus, there are two feedback
loops in the PLL. To prevent the ADC from affecting the
stability of the global feedback of the PLL, the loop bandwidth
of the ADC is designed to be much wider than that of the
global feedback loop of the PLL.
While the PLL is turned off (power-down mode), the phase
frequency detector (PFD) and CP are turned off, which means
the loop filter output is floating. Therefore, the loop filter output
can deviate from its originally locked value determined during
normal operation. Of course, the 10-bit digital code of the
DAC does not change even during the power-down mode,
which means the outputs of the DAC and the loop filter are not
equal during the power-down mode. When the PLL wakes up
from the power-down mode, the ADC loop tries to equalize the
DAC output to the loop filter output, although the DAC output
already has the correct value. Then, the PLL must be re-locked,
resulting in a long wake-up time as shown in Fig. 4. To avoid
this, a switch is inserted between the DAC and the loop filter as
shown in Fig. 5. The switch is turned on during the powerdown mode to prevent the loop filter output from floating and
to ensure that the loop filter output is the same as the DAC
output. The clock inputs of the 10-bit counter and DAC are
delayed in relation to that of the comparator to provide a
sufficient timing margin for each block.
Because the output of the DAC controls the VCO, the glitch
of the DAC which occurs when the control code changes must
be minimized (For example, with binary weighted architecture,
a large glitch occurs when the code changes from 0111111111
to 1000000000). Therefore, the 10-bit DAC is thermometer
ETRI Journal, Volume 29, Number 4, August 2007
Column
Up
Comp
CLK
10-bit
counter
Dn
td
Power-down
mode
td
VCO
Row
Decoder output
Loop filter
DAC
10-bit
digital code
DAC code
(a)
Fig. 5. Tracking ADC loop with switch between the DAC and
loop filter outputs to prevent the loop filter output from
floating during power-down mode.
Row
...0 1 1
Column
1 1 1 ... 1 1
Column
Row
Row
...1 1 1
Row first
Column
1 1 1 ... 1 1
Row
...0 1 1
Decoder output
Column first
Column
0 1 1 ... 1 1
Column
0 1 1 ... 1 1
DAC code
Row
...1 1 1
(a)
Column
1 1 1 . . . 1 1
Row
. . .0 1 1
(b)
Fig. 7. (a) Decoding scheme for glitch minimization and (b) the
principle of glitch minimization.
Column first
Row first
Column
1 1 1 . . . 1 1
Row
...1 1 1
Row
...0 1 1
Column
1 0 0 . . . 0 0
Row
...1 1 1
Column
1 0 0 . . . 0 0
(b)
Fig. 6. (a) Conventional decoding scheme of thermometer-coded
DAC and (b) the mechanism of large glitch generation.
ETRI Journal, Volume 29, Number 4, August 2007
coded and built with a segmented structure of 9 bits with unit
current cells and 1 bit with a binary scaled current cell [6]. The
array of the current cells is configured as a four by five matrix.
With a conventional decoding scheme, rows are selected first,
and then an appropriate number of columns is selected as
shown in Fig. 6(a). If mismatch exists between the delays of
the control signal paths for the row and column switch, the
thermometer-coded DAC still exhibits a large glitch when
another row is selected as shown in Fig. 6(b) [4], [5]. If the
column (row) control path is faster, one less (more) row is
temporarily selected, creating a large glitch. To avoid this
problem, the improved decoding scheme shown in Fig. 7(a)
proposed in [4] is adopted. This scheme uses different control
logics for even and odd rows and limits the glitch to less than
2-LSB even with delay mismatch between column and row
control signals as shown in Fig. 7(b).
Sooho Cha et al.
465
Even with the input offset of the comparator, the PLL can be
locked with some difference between the outputs of the loop
filter and DAC at the completion of locking. However, the
offset of the comparator must still be compensated. During the
power-down mode, the loop filter output is forced to be the
same as the DAC output to prevent the loop filter output
floating and to enable fast wake-up as explained above. If the
locked value of the loop filter output is different from the DAC
output due to the input offset of the comparator, the PLL should
be re-locked when the system exits from the power-down
mode as illustrated in Fig. 8.
As shown in Fig. 9(a), the comparator is composed of a preamplifier and a sense-amplifier type latch where the input
offset is compensated at the pre-amplifier output [7]. Because
the control voltage of the VCO can vary within a wide range,
the pre-amplifier of the comparator employs a rail-to-rail input
stage as shown in Fig. 9(b). The timing of the comparator is
DAC output
Power-down
Wake-up
Loop filter output
Comparator offset
Fig. 8. Offset of comparator results in long wake-up time.
VCM
Q1
IN
INB
VCM
Q2
Q1
Q2
+
+
Q2
Q2
VCM
Pre-amp
VCM
M7
Latch_clk
M8
M5
M6
M3
M4
LN
OutB
Rctrl
Out
In
(40/0.35)
InB
M2
M1
Latch_clk
(20/0.35)
LP
Delay cell
Latch
M1, M2: W = 20 m, L = 0.18 m
M3, M4: W = 5 m, L=0.18 m
(30/1)
Vctrl
M9
-
M5, M6, M7, M8: W = 10 m, L = 0.18 m
M3, M4: W = 10 m, L = 0.18 m
Rctrl
(a)
M11
M12
Vref
M10
M9
M2
InB
M3 M4
OutB
In
Vctrl
Out
VCO replica bias generator
M1
7-stage ring oscillator
(a)
M5
M6 M7
M8
2000
M1, M2: W = 20 m, L = 0.5 m M5, M6, M7, M8: W = 20 m, L = 0.5 m
M3, M4: W = 80 m, L=0.5 m M9, M10, M11, M12: W = 40 m, L = 0.5 m
1
2
Latch
Offset
compensation
0.6
0.7
0.8
0.9
1.1
1.2
1.3
1.4
VCO control voltage (V)
Latch
Fig. 9. (a) Offset compensated comparator, (b) pre-amplifier with
rail-to-rail input common-mode range and (W/L) of the
transistors (all in m), and (c) timing diagram.
Sooho Cha et al.
800
(C)
466
1200
400
Signal
amplification
Pre-charge
1600
Frequency (MHz)
(b)
NN
SS
FF
(b)
Fig. 10. (a) Voltage-controlled oscillator and (b) its voltage-tofrequency characteristics (NN: VDD=1.8 V, 25oC, normal
process corner; SS: VDD=1.6 V, 100oC, slow process
corner; and FF: VDD=2.0 V, 0oC, fast process corner).
ETRI Journal, Volume 29, Number 4, August 2007
Single-tone
frequency estimator
7.99999999999806e+007
Frequency in Hz
Vctrl
2
Simout_int
To workspace
Pulse
generator
REF
Up
In1
VCO
On
In2
PFD
Out1
RC charge pump
In
In1 Out1
VCO
Out
Comp & DAC1
Convert to
square wave
Divide frequency
by 10
Transport
delay
Single tone
frequency estimator
8.001289062511102e+008
Frequency in Hz
2
CLK
Osc under test
Target clock
10
Reset after N cycles
1.6110672298726
p-p jitter in ps
Cycles until reset
Jitter
Jitter measurement with respect to target clock
Jitter 1
Fig. 11. Simulation setup for behavioral simulation of PLL.
shown in Fig. 9(c), where 1 and 2 are non-overlapping
clocks. When 1 is high, the input and output nodes of the preamplifier are connected to the reference voltage and the offset
is stored on the capacitor. When 2 is high, the offset voltage is
cancelled, and only the input differential voltage is amplified.
When Latch is LOW, the output nodes of the latch are precharged to VDD and when Latch goes to HIGH, the latch
senses the output of the pre-amplifier.
The VCO shown in Fig. 10(a) is a fully differential ring
oscillator for low sensitivity to common-mode noise. The
oscillation frequency can be controlled from 350 MHz to 1.3
GHz at the nominal condition (NN) as shown in Fig. 10(b).
The active VCO control voltage range is from 0.6 V to 1.1 V,
which can be easily covered by the rail-to-rail input stage of the
comparator in Fig. 9(b). The replica bias ensures the constant
swing of the output of the VCO [2].
III. Experimental Results
Because the proposed PLL has two feedback loops, it is very
difficult to check the stability and determine the loop
parameters analytically. Therefore, behavioral simulation has
ETRI Journal, Volume 29, Number 4, August 2007
been performed by Matlab with the simulation setup shown in
Fig. 11 [8]. From the behavioral simulation results, the
bandwidth of the global feedback loop is set to 500 kHz, while
the DAC is clocked at a much faster rate of 40 MHz to let the
tracking ADC loop have a wider bandwidth. The locking
behavior of the PLL is simulated by HSPICE, and the resultant
waveforms are shown in Fig. 12. After the loop is locked, the
control code of the DAC changes by 1-LSB; thus, the VCO
control voltage ripples by about 2 mV, which will result in
20 ps peak-to-peak jitter. The wake-up time of the proposed
PLL is smaller than 0.1 s which is mainly determined by the
start-up time for the VCO to begin oscillating while for a
conventional analog PLL, the wake-up time would be in the
order of several tens of s.
The PLL has been implemented in a 0.18 m CMOS
process whose microphotograph is shown in Fig. 13, and the
active area is 1.0 mm0.35 mm. The measured rms jitter of the
1.2 GHz output clock is 16.8 ps, while the peak-to-peak jitter is
84 ps as shown in Fig. 14. The peak-to-peak jitter is larger than
the value obtained from the behavioral simulation in Fig. 12
because other sources of jitter exist such as the thermal noise of
transistors and power supply line noise. From a 1.8 V supply
Sooho Cha et al.
467
voltage, the PLL consumes 59 mW and 984 W during the
normal operation and power-down mode, respectively. In Table
1, the measured performance of the PLL is summarized.
Voltage (V)
0.85
Loop filter output
0.84
IV. Conclusion
0.83
DAC output
0.82
0.0
5.0
10.0
15.0
Time (s)
Fig. 12. Simulated locking behavior of PLL.
A 0.4 GHz to 1.2 GHz PLL has been developed which has
embedded ADC to store the locking information as digital code.
The embedded ADC has 10-bit resolution with tracking
architecture, where DAC is in the feedback loop. The DAC
employs a glitch minimizing decoding scheme for small jitter.
Because the locking information is stored as digital code, the
PLL can be turned off during power-down mode, while
avoiding long wake-up time.
References
Fig. 13. Microphotograph of PLL.
Fig. 14. Measured jitter histogram of 1.2 GHz output clock.
Table 1. Performance summary of PLL.
Technology
0.18 m 4-metal CMOS
Die area
1.0 mm 0.35 mm
Supply voltage
1.8 V
Power dissipation
59 mW : normal operation
984 W : power-down state
Jitter
RMS : 16.8 ps
Locking range
400 MHz 1.2 GHz
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Sooho Cha et al.
[1] T. Olsson and P. Nilsson, A Digitally Controlled PLL for Digital
SOCs, IEEE J. Solid-State Circuits, vol. 39, no. 5, May 2004, pp.
751-760.
[2] S. Kim, K. Lee, Y. Moon, D.-K. Jeong, Y. Choi, and H.-K. Lim,
A 960-Mb/s/pin Interface for Skew-Tolerant Bus Using Low
Jitter PLL, IEEE J. Solid-State Circuits, vol. 32, no. 5, May 1997,
pp. 691-700.
[3] J. Maneatis, Low-Jitter Process-Independent DLL and PLL
Based on Self-Biased Techniques, IEEE J. Solid-State Circuits,
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[4] J. Lin, B. Haroun, T. Foo, J.-S. Wang, B. Helmick, S. Randall, T.
Mayhugh, C. Barr, and J. Kirkpatrick, A PVT Tolerant 0.18
MHz to 600 MHz Self-Calibrated Digital PLL in 90 nm CMOS
Process, Dig. Tech. Papers, Int. Solid-State Circuits Conf., Feb.
2004, pp. 488-489.
[5] S. Cha, C. Jeong, C. Yoo, and J. Kih, Digitally Controlled Phase
Locked Loop with Tracking Analog-to-Digital Converter, Proc.
IEEE Asian Solid-State Circuits Conf., 2005, pp. 377-380.
[6] C.-H. Lin and K. Bult, A 10-bit, 500 Msample/s CMOS DAC in
0.6 mm2, IEEE J. Solid-State Circuits, vol. 33, no. 12, Dec. 1998,
pp. 1948-1958.
[7] B. Razavi and B.A. Wooley, Design Techniques for High-Speed,
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27, no. 12, Dec. 1992, pp. 1916-1926.
[8] Matlab Manual, Mathworks Inc., 2005.
ETRI Journal, Volume 29, Number 4, August 2007
Sooho Cha received the BS and MS degrees in
electronics and computer engineering from
Hanyang University, Seoul, Korea. He joined
the Memory Division of Samsung Electronics
as a member of research staff, Hwaseong,
Korea, in 2006. His main research interests
include mixed-mode CMOS circuit design and
high-speed interface circuit design.
Chunseok Jeong received the BS degree in
electronics engineering from the University of
Seoul, Korea, in 2003, and the MS degree in
electronics and computer engineering from
Hanyang University, Seoul, Korea, in 2005. He
joined the DRAM Design Team of Hynix
Semiconductor Inc., Ichon, Korea, where he is
involved in the development of 1G DDR3 SDRAM. His research
interests include phase-locked loop, delay-locked loop, data converters,
and thermal sensors.
Changsik Yoo received the BS (with the
highest honor), MS, and PhD degrees in
electronics engineering from Seoul National
University, Seoul, Korea, in 1992, 1994, and
1998, respectively. From 1998 to 1999, he was
with Integrated Systems Laboratory (IIS), Swiss
Federal Institute of Technology (ETH), Zurich,
Switzerland, as a member of research staff working on CMOS RF
circuits. From 1999 to 2002, he was with Samsung Electronics,
Hwaseong, Korea. Since 2002, he has been an associate professor of
Hanyang University, Seoul, Korea. He is the winner or co-winner of
several technical awards including the Samsung Best Paper Bronze
Award in the 2006 International SoC Design Conference, the Silver
Award in 2006 IDEC Chip Design Contest, the Best Paper Award in
the 2006 Silicon RF IC Workshop, and the Golden Prize for research
achievement in next generation DRAM design from Samsung
Electronics in 2002. His main research interests include CMOS RF
transceiver design, mixed mode CMOS circuit design, and high-speed
interface circuit design.
ETRI Journal, Volume 29, Number 4, August 2007
Sooho Cha et al.
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