er Ele Uren elt gs | th eee ~,, september 1975 35p up-to-date electronics , w#*¢. & «3902 — elektor september 1975 publisher's notices ‘boards. These boards can be ordered from our Conterbury offi Payment, including £ 0.15 p & p, must be in advance. Delivery time is approximately three weeks. Bank account number: A/C No, 11014587, sorting code 40-16-11 Midland Bank Ltd, Canterbury. All prices include VAT at the rate shown in brackets. Many Elektor circuits are accompanied by designs for printed circuits. For ‘those who do not fee! inclined to etch their own printed circuit boards, ‘a number of these designs ae also available as ready-etched and predrilied umber ive price % VAT Hen 5 110 (25) fers 058 (2a) mers 8 80 asl Stmtereo die preamp Hele (088 (a) eee tae otees fos § S40 te) ‘imorsion meter a1 188 1d converter 14a 3080 (8) tap enor 14571080 (8) minidram ayrator 1485q 2080 (25) tnindrum meres 14668-2055 (25) trina noe twee 2 105 (5) beet ‘oz 4-220, 8) qu amplifier 1091120, 5) slectronie loudspeaker 1827208025) rmostop te 2108 (8) cor power supply wes 4128 (8) dita rev counter (contol pb. only!) 1690 1 055 (8) car anihelt lar i024 140 (8) tos clock S314 lok circuit tora 1 148 (8) tos clock 6314 Spi boars 1078 1 088 (8) thos eock tebace iso 4 070 (8) minim tp. ia1a 2 070 (25) Imiigrum raffle circuit were 3 110 (25) sutomaticbssdrum tere 3 058 (2a) tnirodrure ter 2 098 (28) tera plier tess 1085 5) SSilessrecenerfor MW and LW B68 O28) So presne fos 4 18028) {ann miitron Silay wos. 2 140 (8) twin ed ploy 40022 1a (8) thin cade counter ws3 2 1a (8 recon 408, 2 050 (8) Glee preamp 76131 down 3 095 (28) ‘vax! depay a3 2180 (8) Site probe sows 2 185 (8) big bon 95 5028. 2128 (25) compressor Soren 3120 (28) found fo = «2 tad (25 {up tester soe 4 170 (8) {plan tester front pane 8076728 4 190 (8) pcb, and wiring tester 91065 (088 (8) Mhythm generotor M252 oo «8 (0a) 5) 17200 sren sie 5 075 (5) A209080 ser00 kode 2188 080 25) fatonen timer our 8 078 8) : 913-8078 8) NeW: reve umber iewe price % VAT sci aie 9759 6 120 (25) ‘imatore smpier fae S085 28) Tart dimer tar Oas (8} erste digital clock tua 8 V0 {8 Sire (2 bourse) me 8 1 18 Sr clock front pana (raneparent plastic 70363 6 090 (8) ELEKTOr Volume 1 — number 6 Editor W. van der Horst Deputy editor: P. Holmes Art editor C. Sinke Drawing office : L. Martin Subscriptions: Mrs. A. van Meyel UK. Staff: Editorial T. Emmens Advertising: P. Appleyard al offices, administration and adverts 6 Stour Street, Canterbury CT1 2X2. ‘Tal. Canterbury (0227) ~ 54430. ‘Telex: 965504, Elektor has been published every two months until ‘August 1975; it now appears monthly. ‘Copies can be ordered from our Canterbury office. ‘The subscription rate for 1975 is £ 3.60 (incl. p & pl: tha first iseue (Nov/Dec 1974) will be included in this at no additional cost. Single copies: £ 0.36 (not incl. p & p). ‘Subscription rates (airmail): ‘Australia/New Zealand — S issues £7.20 Europeen countries outside UK = issues £5.20 roy = Bissues £6.25 All other countries = Bisues £6.40 ‘Subscribers are requested to notify a change of ‘address four weeks in advance end to return envalope ‘bearing previous address. Members of the technical staff will be available to ‘answer technical queries (relating to articles published in Elektor) by telephone on Mondays from 14.00 to 16.30. Letters should be addressed to the department concerned: TQ = Technical Queries; ADV = Advertise. ‘ments; SUB = Subscriptions; ADM = Administration; ED = Editorial (articles submitted for publication etc); EPS = Eloktor printed circuit board service. ‘The circuits published are for domestic use only. The submission of designs or articles to Eloktor implics [permission to the publishers to alter and translate the text and design, and to use the contents in other Elektor publications and activities. The publishers annat guarante to return any material submited to All drawings. photographs, printed cireuit boards and articles published in Eloktor are copyright and may ‘not be reproduced or imitated in whole or part ‘without prior written permission of the publishers. Copyright © 1975 Elektor publishers Ltd — Canterbury. Printed in the Netherlands.i Se a ————— ‘This l= design for a high-quality 40 W audio amplifier besed on an sarlier 20 W design. The ampliti ‘2m wmusvel Gesign features end the construction is problem-free. ‘The cicuit will operate from 4.5 V battery ‘= small loudspeaker or headphones. The citcuit is not outstanding for its power or quality, but itis simple and relate. MTG 575 15s eb eee. Se) eeaea tesa oe “Onechig’diitl clocks, such as the MMB314 (Elektor i, p. 24) are excalient for simple time keep ‘eer, they 26 not ideal for criving external devices such as ime signals, ealendar, ete. This is where conven ‘Smet clock design with standard TTL ICs scores; it has a BCD-coded time output, as well pulse train Out. ‘buts giving repetition rates varying from one per second *0 one por day. phasing — G. Knapienski and F. Mitschke ....-.-.2-.0+s+seseuseseseeeees owadoys there are 9 greet number of methods of producing unusual electronic sound effects, A fevourite ‘sffect ig "Phasing’ and in this article this is accomplished, somewhat unusually, by using a ‘path filter. Various Kinds of psychedelic flashing light display can be generated easily using logic shift registers. Several Gecuits are discussed here of varying complexity. ‘ance Amplifier is a new type of operational amplifier. The gain can be Get=rmined externally by the value of the output load resistor, and by the so-called bios current. The latter ‘@akes it possible to control tha gain instantaneously over @ range of about 80 dB by an external potential. ‘Serce the OTA will be used in several Elektor projects, an explanation of the working principles of this device should prove useful. luring holidays can be something of a problem. However the device deseribed here wi ‘overcome this problem by automatically dispensing the required quantity of feed each day, tristable — J. Koper ...... 20... sees eee roll out the bandit — L. Wiechers .... seees eee pe ee ‘The cisaventage of the ‘three-eyed bandit’ (Elektor No. 2 page 238) Ws that a Hop button must be pressed to ‘Sop the three oscillators and obtain the final display. Thus the tension and anticipation obtained with @ real ‘one-ermed bandit as the number drums slowly grind to a halt is missing. The circutt described here overcomes ‘this by providing an output whose frequency slowly reduces until it finally stops. dual slope dvm — H.L. Krielen ..........- ‘A design is described for a basic 2% digit digital voltmeter using pe ‘nes full-scale sensitivity of 199 mV, but may be extended at the constructor’s option. tep — A. Schulz . nh nis dha i eeeee ee ered A LEP (light emitting pistol) with electronic time and hit indication offers the possibility of moving the age-old fair attraction ‘the shooting sal’ to the living-room. bicycle trafficator — N. Beun . . . 30 mhz amplifier —W. Kiimmel ....... 20... .0ceceueeeeeceeeececeeseaees ‘The input threshold voltage of TTL IC's lies between 1.8 end 3 volts. This simple transistor amplifier can be sed fo amplify relatively lowJevel signals to a level suitable for driving frequency counters and other equipment. ‘The Eurosil E11091C offers the possibility of a single-chip digital clock operating directly from a crystal refer~ ‘ence. Using this IC a compact clock can be constructed whieh is eminently suitable asa car clock or portable market ...| | \ iT Automatic tester for mono and stereo broadcast circuits For regular interchange of programmes, broadcasting organizations use special programme circuits placed at their dis- posal by the telephone administrations either as permanent connections or, if they cross national borders, as tempor- arily established transmission paths. Until now, assessment of the quality of such circuits was performed manually and met with some difficulties, es- pecially in the case of international programme transmission (non-uniform test methods, different test instruments), ‘as well as being unsatisfactory with regard to the time required for lining-up. With automatic tester K 1060, Siemens are now offering an instrument for auto- ‘matic quality supervision of mono and stereo programme circuits. The automatic tester, designed accord- ing to the most recent CCITT rec- ‘ommendation for programme circuits from 30 Hz to 16 Kliz, consists of a transmitter and receiver, each contain- ing a test and control unit, and a high- speed recorder. Among other things, the weighted (psophometric) and un- weighted noise, non-linear distortions, level step, frequency response, level difference and sum level, phase differ- ence and crosstalk can be assessed. Two main routines (mono and stereo) and nine sub-routines are available. The automatic test routine for mono cir- cuits, including printout of the record, takes only about 133 seconds, while that for stereo circuits takes about 370 seconds. ‘The test unit of the transmitter contains two spot-frequency generators and a function generator which supplies vari- SELERTOr ous spot-frequencies at certain input voltages; a frequency sweep from 30 Hz to 16 kHz is achieved by using a variable de voltage. This dc voltage increases exponentially, producing a logarithmic frequency sweep. In addition to this, beginning at 50 Hz. a pulse is added to the signal at each octave, the recorder registering the pulse as a frequency marker. The output voltages of the three generators are supplied individu- ally or in a combination, depending on the test mode, to an amplifier; they are then set automatically by means of vari- able attenuators to the exact value re- ‘quired in each case. The outputs for channels A and B are balanced and floating, For monitoring purposes, the test routine in progress can be followed over the built-in loudspeaker or over earphones. The control unit of the ‘transmitter, which is equipped with a clock generator, is used for automatic step-by-step execution of the test rou- tine ‘The test unit of the receiver has two balanced floating transformer inputs of identical design for channels A and BB. The test circuit for channel A has attenuators in front of and behind the amplifier which can be cut in auto- matically according to the test mode. ‘These are followed by various filters for weighted and unweighted measure- ment of noise, non-linearity and cross- talk. The signal is applied via a further amplifier to the rectifier, whose output voltage is passed through a logarithmic network to obtain levelinear indication con the recorder. Here again, the test routine can be monitored over the built- in loudspeaker or over earphones. The test circuit for channel B is similar in design to that for channel A. As in the case of the transmitter, the control cir- cuit of the receiver is equipped with a clock generator, and in addition to this it has automatic start-signal selection facilities. A high-speed recorder is connected to the output of the receiver to record the test result. The utilized recording width is 100 mm, corresponding to a range of 20 dB for level measurement and 50 angular degrees for phase-difference measurements. Modem without modulation For the transmission of data signals over switched telephone networks, modems are used to modulate the de signals on the send side and to demodulate them on the receive side. If dedicated circuits are used, however, which is especially favourable over short distances - in con- equipment can be used which operates ‘without modulation and which is there- fore less complex. One device of this kind is the Modem N10 developed by Siemens, which uses direct-current key- ing for transmission. The Modem N10 is suitable for data transmission over metallically coupled two - or four-wire line at bit rates of up to 9600 bit/s. The device has no fixed code or speed and permits all conventional operating modes — simplex, half-duplex and duplex oper- ation. This last mode is also possible over two-wite lines, which substantially reduces line costs. The maximum range is rated - depending on operating mode and speed - at al- most 30 km. Under adverse operating conditions and with a bit rate of 2400 bit/s the error rate is only 10-* ie. out of 100 million characters one may be falsified. The device may also be used for multi- point operation. In this mode, one control station transmits over only one line to a number of outstations. AA station address is previously speci- fied, so that the message is only evalu- ated by the data terminal for which it is intended; if necessary, this can then return an answer to the control station, The operational integrity of the Modem N10 can be checked at any time with the aid of the built-in test facility without further aids. The interface to the data terminal corresponds to the relevant CCITT Recommendations, s0 that the modem can interoperate, without matching problems, with the same data terminals as the modems employed for the speed range up to 9600 bit/s over switched telephoneedwin ampli The Edwin amplifier is unusual in that it embodies two types of output stage in fone amplifier. A class A output stage handles the low level signals and also serves as a driver for a class B stage which handles the larger outputs. The principle of operation is shown in figure 1. T2 and T3 are biased on by the voltage drop across the diodes Di- D3. T2 and T3 function as a class A stage at low signal levels supplying current to the load via resistors R. As the signal is increased the voltage drop across these resistors becomes sufficient to cause T4 and TS to conduct and the class B partof the output stage begins to operate, Crossover distortion is quite low with this type of design The complete circuit As figure 2 shows, the complete amplifier circuit consists of a voltage amplifier, @ class A driver stage and a class B output stage. The input stage consists of TI and T2 in a Darlington configuration, re- sulting in a high input impedance. The signal passes to the base of T3 via the limiting resistor R4. T3 operates as a voltage amplifier and in its collector circuit has T4, which is connected as a simulated zener diode to provide 2 constant d.c. bias voltage of about 2 V across the bases of the driver transis- tors T7 and T8. Feedback is applied be- tween the output and the junction of R11 and R12 to provide a high collector impedance so that true current drive is achieved. This helps to reduce crossover distortion still further so that despite the small amount of overall negative feed- back and the absence of quiescent cur- rent in the output stage the distortion figures are very good 4 sce voltage ample - aaa Gee eS Some Roe features — output power from 10-40 W depending on power supply. — high efficiency. — low crossover distortion. — short circuit proof. — no quiescent current in the output transistors. — output transistors and drivers need not be matched. — unconditionally stable. figures — Sensitivity: = 1V (RMS). — Input impedance: = 45 kQ. — Distortion: 1 kHz, 30 W: 0.1%, 10 kHz, 30 W: 0.3%. — Power bandwidth: 20 Hz — 100 kHz. — SIN ratio: > 90 cB.[TS ee eee 8CI71 gcia7 Bci71 -———-® av Pom, 4x 54,100 Figure 1. Basie circuit of an Edwin-type output stage. Figure 2. Final circuit of the Edwin amplifier for output powers up to 40 W. Figure 3. The power supply. Figure 4. Graph of available output power versus transformer secondary voltage for 4 {2 and 8 loads. % max 4% 80 evloed voltage$12 ~ elektor september 1975 din amplifion The output stage differs from the con- figuration shown in figure 1 because it comprises two NPN transistors of the same type and not a complementary: pair. To maintain symmetrical operation: of the output stage D1 is included across R18. This simulates the base-emitter junction which would be present across R18 if the configuration of figure 1 has been used. The values of R17, R18 and R19 are low (10 2) to reduce cross over distortion, ‘Overall negative feedback is applied from the output to the emitter of T2. The inclusion of C3 means, that 100% dic, feedback is applied, which stabilises the d.c. operating point of the output at around half supply voltage over a wide oe Lemna range of supply voltages without the ee een need for adjustment potentiometers. 6 es Vout alee ! | | | re T | “acon | ee i i lf ce cae ata —~ Frequeneyite} [2 tion (36) 04 ‘Output power 6 dB below os| o2 Frequency (Ha) Bi ee, Le ud Figure 5. Maximum output power versus fre- ‘quency. Figure 6. Frequency response. Figure 7. Distortion versus frequency for ‘output power 6 dB below maximum. Figure 8. Distortion versus output power.| edwin amplifier elektor september 1975 ~ 913 Distortion (6) 1% os on} Freavency the | o5| oa 03 o2| oor oF Output power RW! The a.c.gain of the amplifier is, of course, given by It is worth noting the effect of the com- bination R7, C5 on the operation of the amplifier. Some amplifiers, when used with an’ unstabilised supply, display ripple on the peaks of the waveform when driven to clipping. This is elimin- ated by R7 and C5 as follows. When the amplifier is being driven, current flows through R7 and the voltage on C5 is always below the ripple ‘troughs’ on the supply. The drive voltage available from T3 is limited to the voltage on CS and the output of the amplifier can never swing into the ripple region of the supply voltage. R7 also limits the current through T3 in the event of an overload. Overload protection The protection circuit is designed to prevent excessive current peaks from occurring during signal overloads or short-circuiting of the output. The pro- tection circuit consists of transistors TS and T6. Their base bias is set such that under normal operating conditions the voltage across R20 and R2I is insuf- ficient to turn them on. In the event of excessive output current flowing in R20 or R21, due to a signal overload or a short-circuited output, the voltage across these resistors is sufficient to cause TS or T6 to conduct. This reduces the drive voltage to the output stage and there- fore limits the output current, thus protecting the amplifier. Power supply A stabilised power supply is unnecessary with the Edwin amplifier, as its perform- ance will not be significantly improved. A simple unregulated supply is quite adequate and two suitable circuits are given in figure 3. Figure 32 shows a supply using a normal full-wave bridge rectifier, whilst figure 3b shows a full- ‘wave rectifier with a centre-tapped trans- form: ‘The component values and specification forsupplies suitable for 20, 35 and 40 W versions of the amplifier’ are given in table 1. Of course any suitable trans- former may be used, there is no need to adhere to the exact voltages specified. Figure 4 gives the output power available versus transformer secondary voltage. The only points to watch are that the current rating of the transformer is adequate for the required output power, that the voltage rating of the smoothing capacitor is sufficient and that the RMS secondary voltage of the trans former does not exceed 33 V on load, otherwise the voltage rating of the tran- sistors may be exceeded. Over the range of supply voltages given in figure 4 nothing need be changed in the amplifier as the operating point is self-adjusting. Performance figures ‘The performance figures, as measured on the 35.W prototype of the amplifier are summarised in table II and displayed graphically in figures 5, 6, 7, and 8. As can be seen they are quite exceptional. Among the outstanding features are the large power bandwidth, good signal to noise ratio, immunity to transients, low distortion ‘and absolute stability, even with large capacitive loads. Figure 9 shows the printed circuit board and component layout of the amplifier. Table 1 Po max (W) Ver_| ter max (A) eo Power (RU=4 Ohm) — | AMS| figure 3b figure 3a supply LE working | no-load Mono lottage | voltage Mono | Stereo) Stereo _[Mono|Stereol(v)__| (vd 42 33 jar |22 jas Jo fas. 35 zo |1 [2 [4 [e500 |s000 so | aa. 2 2 jos 15 [3 uo | 34edwin amplifion sloktor soptombor 1975 — 915, @! af Lf eee eee ey becteees Figure 9. Printed circuit board and component, layout. | Components lst for figures 2 and 9 oie €2=340n, sov = G3= 1004 25 i ee Ge= 1opewame @ €5= 250 yu, 50V €6 = 2500 u, 50 V C7 = 3n3 6k8, 3% W AS.R14,RI5 = 100, % W w 680, % Semiconductors: (Cire Ww T6= BC 178, 8 158 8k2, % W 17=80137 RI7,RIE,RI9- 10, % W ‘T9,710 = 2N3055, 8D 130 R20,R21 = 0.12, 2W i= By 127 1916 — oloktor september 1975, 10 “42Wer. 35Wett Wer Hestsink fortwo output Figure 10, Heatsink details for the driver and output transistors. ‘The driver transistors are mounted on the board, with a cooling fin as detailed in figure'10a. The output transistors are mounted on a separate extruded alu- minium heatsink, details of which are given in figure 10b and the associated table. Most manufacturers of heatsinks will have something similar to this in their range. If resistors R20 and R21 are not readily obtainable they may be wound from suitable resistance wire. Alternatively wire eight 1 20.25 Wresistors in parallel, there is plenty of space on the board to mount them vertically (figure 1 1). Concluding remarks Whilst the Edwin amplifier meets an exacting specification this is no reason to recommend its construction by the Hi-Fi enthusiast. There are many other designs with similar performance. What makes the amplifier eminently suitable for the amateur is its problem-free con- struction and virtual (electrical) inde- SS structibility, K Performance figures of 35 W version ‘Maximum output power 35 W (4 9); 20 W (82) 45 W (4.9); 27 W (BQ) f= 1 kHz, THD Efficiency >60% = 1 kHE: Po = 35 W Load impedance 0... 29(Maximum poner into 42) Overload protection _| Proof against long duration short-circuit Maximum capacitive load | > 100 uF (!) Sensitivity =1V RMS kHz, Po = 35 W Input impedance = ab KD Distortion Po = 0. 30.W 01% Tee 02% f= 30 He 03% tokHe Frequency response | 25Hz... 1,2MHz(-348) | Vig = 245 mV 40 Hz .. 1/0 MHz (—1 dB) Power bandwidth >100 kHz (-3 48) Noise rejection 73.48 93.08 input open-circuit input short-circuit Signal to noise ratio 9548 >105 48 input open-circuit input short-circuit Feedback factor Stability 36 d8 unconditional‘The input and driver stages T1 and T2 operate as voltage amplifiers. The out- put stage, T3 and T4, operates in class B to achieve long battery life. D.C. feed- back is provided by means of R3 and AC. feedback by means of R3, R4 and 2. This defines the gain, stabilises the ‘operating point and increases the input impedance. The biassing of TI is critical and the values for R1 and R2 must be adhered to, Should the circuit fail to operate cor- rectly the D.C. conditions may be checked at the base of T3 and T4 and the junction of R6 and R7. If 25 ohm loudspeakers are difficult to obtain, then 8 or 15 ohm types may be used instead. In-that case R6 and R7 should be replaced by wire links. ‘As can be seen from figure 2 the p.c. board is extremely miniature and finding space in the record player cabi- net should be no problem. To improve loudspeaker efficiency the loudspeaker cabinet should be as large as possible. Figure 1. The very simple amplifier circuit. Figure 2. Layout of match-box size printed- circuit boerd. Input impedance 2 2600 k Input sensitivity vj = 200 mV (r.m..) ‘Output powor (with 25 $2) P = 50 mW. Parts list. R6=220 R7=220 Pt =2M2 log Resistors: Capacitors ct=470n c2= 10u/t0v C3 100p/10V Semiconductors: 174 =TUP 12.73 = TUN 1,2 = DUS Sundi Torch battery 4.5 V Loudspeaker 25 ee SRR, O09 © He He -0 vt= slektor september 1975 versatile digital clock. Elektor has previously published designs for ‘one-chip’ digital clocks, with both ‘mains and crystal reference frequencies. Whilst ICs such as the MMS314_ are excellent for simple timekeeping, they have certain disadvantages. Since the output to the display is multiplexed the time output of the clock is not easily accessible in a parallel form, This means that the clock is unsuitable for driving time-controlled devices such as alarms, calendars, central heating programming, automatic recording of radio pro- ‘grammes or other systems. The clock described in this article is based on TIL circuitry and is eminently suitable for control systems. The time is avail- able as a BCD coded output, and clock pulse trains with rates varying from one a second to one a day are obtainable. Many constructors will probably have some of the ICs in their junk box’. The complete circuit of the clock (ex- ‘cluding power supply) is given in fig- ure 1. The basic operation is quite simple, With all the switches in the Positions shown the clock runs nor- mally. The 50 Hz input is rectified by DI, clamped to 4.7 V by D2 and then fed into the NAND Schmitt trigger ST1. ‘ASO Hz square wave suitable for driving TTL appears at the output of STI and is fed to IC1O which is connected as a divide-by-five counter. Asymmetric 10 Hz. pulses are available at the ‘D’ out- put of ICI0, These puilses are fed to IC9, which is connected as a divide-by- 10 counter. A symmetrical 1 Hz square wave is available at output ‘A’ of this Ic. The 1 Hz pulses are fed to IC6, which is connected as a BCD decade counter. This counts seconds from 0 to 10 and the BCD output may be decoded for display using a 7447. The ‘D’ output of 1C6 produces one pulse every ten sec- ‘nds, and this is fed to ICS, which is connected as a divide-by-6 cotinter. This counts tens of seconds from 0 to 6. When the tens of seconds count reaches 6 (ie. the seconds display changes from 59 to 60) the BCD output of ICS is 0110, that is to say the ‘B’ and ‘C” out- puts of the IC are both ‘1’, These out- puts are connected to the Reset 0 in- puts, so that when the count reaches 6 ICS ‘is reset instantaneously, and the 6 display is never seen, One puise per minute is obtained from the °C’ output of ICS, and this is fed through N2 and N1 to IC4, which is again connected as a BCD decade coun- ter. The time-setting circuits around NI and N2 will be discussed later. Like ICS, 1C3 is connected as a divide- by-6 counter, so that it counts tens of minutes. Counting of the hours is slightly more complicated. Since the clock is a 24 hour design, the hours counter (IC2) ‘must count up to 10 twice, then reset at 40on the third count sequence (i.e, when the hours count reaches 24), Since the tens of hours counter only counts to 2 a counter is made up from two JK flip- flops (7473) instead of using a 7490. Resetting is accomplished as follows: During the first 0-10 count of IC2 the Qoutputs of FFI and FF2 are low. When the ‘D’ output of IC2 goes low on. the tenth count the Q output of FFI goes high. At the end of the second count sequence the Qoutput of FFI goes low and the Q output of FF2 goes high. The Qoutput of FF2 and the SC’ output of IC2 are connected to the Reset 0 inputs of 1C2, so that when IC2 reaches 4 in its third count sequence it is reset. However FF2 cannot similarly be reset as it has no gating on the clear input. This difficulty is overcome by feeding the ‘B' output of IC2 to the clear input of FF2 via C1 and R3. On count 4 of IC2 the ‘B’ output goes iow, feeding a momentary reset pulse to FF2. Of course this occurs at count 8 also, and during the first and second count sequences, However, it is only during the third count sequence that the Qoutput of FFI is high anyway, so these earlier reset pulses do not matter, since the flipflop is reset already. The capacitive coupling (C1, R3) is necessary to ensure that only a short ling were used then the clear input would be held low on count 10 during the second count sequence, and the Q output of FF2 could not go high, Provision of ‘tick’ It will be noted that IC9 is connected differently from the other divide-by- 10 counters (IC2, IC4 and IC6). This is because a BCD output is required from the other counters. IC9 is connected to give a symmetrical square-wave output, as a convenient simulated ‘tick’, and this happens to sound better with a 1:1 mark-space ratio, Time-setting Three timesetting switches ate pro~ vided. Two to make the clock advance: at a fast rate, and one to stop the clock This is useful because the clock can be set to a particular time, stopped, then the stop button can be released exactly oon the time signal from radio or tele- phone, It is also handy if the clock is accidentally advanced too far as it saves going all the way ‘round the dial. Gating for the timesetting is provided by a 7400 (IC7) plus the spare half of the 7413 Schmitt trigger (IC) The operation is as follows: when $2 is in the position shown in figure 2a the set-reset flipflop N3/N4 is reset, so the output of N3 is high and the output of N4 is low. This means that the output of ST2 is high. Pulses from output °C’ of Figure 1. Cireuit diagram of the clock. The Drinted circuit bosrd does not include the dis. play ond its essocisted decoder/drivers, Figure 2a. Logic levels st NAND gztes for normal timekeeping. Figure 2b. Logic levels st NAND getes for reset pulse is provided. If direct coup- timesorting.ben a 7 500 text ee “we MS aA i song \ytoHe —1He ‘ od Ns sm ico 27413 ; 7 Na seo txt oF. 2bversatile digital clock: ICS are thus transferred through N2 and NI. When $2 is changed over (figure 2b) the flipflop is set. The output of N3 is low and the output of N4 is high. The output of N2 therefore goes high. 1 Hz or 10 Hz pulses (depending on position of S1) are now transferred through ST2 and NI to the input of IC4. The clock will therefore count at the rate of one minute per second or 10 minutes per second. As an alternative to the 10 Hz rate, 50Hz pulses may be used. This rate is useful only for setting the hours rapidly. The flipflop is necessary to suppress contact bounce on S2. The flipflop is set (or reset) when the switch initially makes contact on being changed over. Subsequent switch bounce will not affect the state of the flipflop. When $3 is changed over the 1 Hz drive is disconnected from C6 so the clock stops. The position of $3 during time- setting with S2 is unimportant, Power Supply The clock requires a supply of about 1A at S V. As transient interference on the mains supply could interfere with the timekeeping of the clock a stable, well-filtered mains supply is essential, ‘The circuit of figure 3 is recommended, as this can deliver up to 2 A and is well stabilised, The 50 H1z drive for the clock can be derived from either side of the transformer secondary winding, Construction The p.c, board and layout for the clock are given in figure 4, and the assembly requires little comment. The BCD out- puts of the counters are brought out to the edge of the board. Display decoding is not provided on the board. Suitable decoder and display boards are the ‘Universal Display’ (Elektor No 2, Page 223). If zero suppression on the tens of hours display is required pin 5 of the 7447 should be grounded. The layout and p.c. board of the power supply are given in figure S. The output voltage of the supply should be set to 5 V before connecting to the clock. Pars list Resistors: R1,R2 = 3k9 R; R7=1k RB,R9,RI0 ~1.5 2 PI= 1k, preset Capecitors: cree caca=in cs. ce 200 1/16 V 220 u/aV 7 = 10m/16V ce. 470 116.3 V Semiconductors: Bridge rectifier, e.g. 820C2200 omitted, 2,03 = DUS pa’ 1 zener 4.7 V, 400 mW TuP 12,73,75 = TUN Tas BD240 or equ. Sundries: FL Te 2 A delay action fuse ransformer, 8 V/2 A elektor september 1975 ~ 921 Clock FFI FF2 [Leo Pulse dis Num-|tn-] Our] in- | outputs |play ber | put | put | pur +-———} ai} at} v2] a2 [a2 aa} a2] | @) |e |con-| BD] on-) Bcd | ted | A | ted) B & | ar Table 1. BCD code. Table 2. Truth table for 1C1 (7473 connected a5 1:3 divider), Figure 3. Circuit of 6.V stabilised power sup- ply. Figure 4. Printed cirouit and component lay- ‘ut of the clock, Figure 5, Printed circuit and component lay- ‘out of the 6-V stabilised power supply. Toble 1 inpur: count | lojelsla 0 |oJo}ojo 1 Joloolr 2 |olo}1/o 3 fololalr 4 ofrlolo 5 |olrlolr 6 ofr|1)o 7 [olalafy 8 |r fololo 9 |1 Jojo):822 — elektor september 1975, phasing G. Knapienski and F. Mitschke er of nur’ ere ore 28 Faust on oust pt ro a oe nse “pn ind nN met is oye Jou ao pretation ec ‘well 1y We «nO and ef ke ee me atest oy ie Se 900 eo eve 0 wet DEAE race on ero “pris 68 ye sit vaio Phasing occurs when a portion of a signal is delayed and then mixed with the original signal, In the middle of the audio spectrum delays of less than about 100 us will produce no noticeable effect, whilst delays greater than about 30 ms will produce a distinct echo. A delay between these limits will give the required ‘phasing’ effect. Of course, a fixed delay time will not hhave the same effect on signals of all frequencies. If, for example, a 1 kHz signal is delayed by exactly 1 ms and mixed at equal amplitude with the original, then the result will be a signal with twice the amplitude of the original, since the delayed signal has in fact been phaseshifted by 360°. For a 500 Hz signal, however, the situation is quite different, Here a 1 ms delay corresponds to a 180° phase shift, so if the delayed signal is mixed with the original signal the two will cancel, resulting in no sig- nal, This cancellation will occur for all frequencies for which the delay time is an odd number of half-periods. For example with a delay time of 1 ms and a 1.5 kHz signal, the (delayed signal is phase shifted by 540°, or 3 half cycles. At 2.5 kHz the delayed signal is phase shifted by 5 half-cycles, ‘As with the 1 KHz signal, all signals for which the delay time is an even number of half periods have their amplitude doubled, This is true for 2 kHz, 3 kHz, 4 kHz etc, The result is a series of peaks and nulls throughout the spectrum, as shown in figure 1. A circuit that pro- duces this type of response is known as a ‘comb filter’, because of the unusual shape of the response curve. Practical Realisation Early attempts at phasing often used tape recorders running slightly out of synchronism, but this entails a number of difficulties, not least of which being that the sound is not ‘live’, and conse- quently the musician cannot adjust the sound during the performance. There are numerous methods of achiev- ing “live” phasing. Electrical delay lines are impractical for the relatively long delay times required. Electromechanical delay lines can be used to give the required delay, but their delay times are fixed by their mechanical dimensions. All-pass LC or RC phase-shift networks may also be used, but these have the disadvantage that the phase shift cannot easily be varied over a wide range, An obvious solution would be to use an analogue shift register such as the TCAS9O, but these devices are rather expensive ‘A cheap alternative is the path filter, the principle of which is shown in figure 2, SI-ST are closed and opened success- ively at a high rate, i. SI is closed, then S2 is closed while $1 is opened, then $3 is closed while $2 is opened and son. This cycle is repeated continu- ously. When a particular switch is closed, the associated capacitor can charge from the input voltage through the input resistor R, The voltage on each capaci tor is dependent on the time con- stant RC (which is fixed if all the capaci- tors have the same value) the time for which each switch remains closed (which is also fixed) and the instantaneous level Of the input signal, It is therefore apparent that after a cycle of the switch sequence the voltages on the capacitors are a sample replica of the input waveform during that period (albeit slightly distorted due to the non- linear charging of the capacitors). If successive cycles of the input wave form and the switching cycle occur in the same phase relationship, then the voltage on each capacitor will eventually become equal to the input voltage at @ particular point along the waveform. No further charging of the capacitors will occur, and the input signal will be avail- able at the output. This is true for the frequency at which one cycle of the input frequency is equal to the switching cycle time, and also for multiples of that frequency. At other frequencies the signal is heavily attenuated, Consider what happens when a half-cycle of the input waveform is equal to the switch cycle time, Im- agine that on the positive half-cycle the peak of the input waveform is stored on C4 in figure 2. During the negative half- cycle $4 will be closed at the trough of the waveform. The net voltage on C4 will be zero. This is true for the other capacitors, so the output signal is zero. This wil ‘also occur at all frequencies a eal r Frequency Tee EL To therest ofthe pathsphasing where an odd number of half-cycles is equal to the switch cycle time. In practice, of course, the switching is accomplished electronically, for example by a ring counter, The result is a comb filter whose rejection fre- quencies can be varied by varying the clock frequency of the ring counter. The Qufactor can be altered by the Single input resistor, R. Distortion of the output signal may be reduced by in- creasing the number of ‘paths’, i. the number of capacitors. A practical realisation of a 40-path filter is shown in figure 3, A 7490 decade counter and a 7474 dual D-flip-flop form a divide-by-40 counter. The out- puts of the 7474 are decoded by ten 7401 packages, each of which switches four capacitors, making 40 in all, The outputs of the 7490 are decoded by a 74141 BCD-to-decimal decoder/driver and used to switch the supplies to the 7401's via PNP transistors. The capaci- tors are thus arranged in a 4x 10 matrix, and ate switched as follows: At the start of a cycle the outputs of the 7474 are all ‘0’ so the capacitors connected to pin 4 of each 7401 are switched in sequence as the 7490 counts from 0 to 9 and the supplies to each 7401 package are switched in tun, When the count reaches 10 output E of the 7474 becomes ‘I’, ‘The capacitors connected to pin 13 of the 7401's are switched as the 7490 counts the second decade, and so on. The Q-control is provided ‘by the 50k potentiometer. The signal source must have 2 low output impedance and the output of the filter must be connected toa high impedance load, Applications This filter has a very narrow bandwidth, with the Q-control at maximum typi cally less than asemitone, Various effects can be obtained with the circuit, If a narrow pulse waveform is fed in, chimes or percussion effects can be produced at the output, depending on the control frequency. ‘Aircraft noises and other engine noises can also be simulated by filtering out harmonics of complex tones, ‘The phasing effect occurs when a clock frequency is used which is higher than the upper limit of the audio spectrum (say 20 to 100kHz). The Q-control ‘must be set in a fairly high position, The path filter may also be used with an electronic organ or synthesizer, to Produce strange effects, A particularly unusual sound can be obtained by feeding the clock input of the filter from the signal outputs of an electronic ‘organ (squarewave outputs from div- ders, before filtering) and by feeding a noise signal into the signal input. The results are, to say the least, unlike any ‘organ in existence, ‘The circuit as described does have its limitations. It will not operate effec- tively below the frequency whose half- cycle is the same length as the counter cycle. Lowering the clock frequency to compensate for this introduces problems lektor september 1975 ~ 923 3 ql sv : BV a ae Ei lock ‘generator eat ara op amo al sc? e Esch 10 107401 10 to 8 other 680.0 ‘nd 7401 IC's me » with noise due to the clock frequency and the switching of the supplies to the 7401's. This is aggravated by differences in the characteristics of the 7401's and of the transistors, Increasing the number of stages so that a higher clock fre- quency may be used will overcome many of these problems, K Figure 1. Frequency response of a comb filter. Frequencies phasoshifted by odd multiples of 180” are almost completely rejected, Figure 2. Principle of a path filter. All capaci- tors have the same value and the number of ‘capacitors may be optionally increasod almost indefinitely. Figure 3. Circuit for a practical path fitter, ‘The 7401 and the associated supply switching transistor are duplicated 10 times. Each 7401 fs connected to the outputs of the 7474 es shown. Figure 4. Showing the intemal circuitry of ‘one gate in a 7401 package, and how each ‘capacitor is connected, ‘To Decoder maa i 1 e740. ' 1824 — elektor september 1975 Various kinds of psychedelic flashing light display can be generated easily using logic shift registers, Several cir cuits are discussed here of varying complexity. The shift register chosen for this appli cation is the 7495. This is a four-bit par- allel/serial load, parallel/serial out, shift- left, shift right register, and was chosen because of its versatility. The circuit for generating a type of dis play commonly used is given in figure 1. Four lamps light in sequence until all are lit, then all are extinguished simulta- neously. The circuit operates as Follows ‘The outputs A to D of the shift register are initially at ‘0° so the mode control input (pin 6) is low. In this mode serial data is entered at pin 1 and is shifted one place right on each clock pulse. Since the serial input is held high by the 1k resistor, outputs A to D successively go high until all are high. When output D goes high the mode control goes high with it and the shift register is now in the parallel load, shift left mode. The parallel inputs A’to D are grounded so that ‘0's are entered and subsequently appear on the outputs. The cycle then repeats. A truth table for the sequence is given in Table 1 Figure 2 is a variation on the circuit of figure 1. Instead of changing the mode when all the outputs have become ‘1° the D output is connected to the serial input via an inverter. When the D out- put goes high this input is held low so that ‘0's are entered, The outputs A to D go low in turn. When all are low the serial input goes high and the sequence repeats. The visual effect is that lamp A lights, then lamp B and so on until all are lit. The lamps then extinguish starting with lamp A until all are ex- tinguished, Table 2 is the truth table for this se- quence. A reset button is provided to clear the register of unwanted states that may occur at switch-on. It is also possible to operate the shift segister in the serial in, shift left mode. To do this it is necessary to connect wvausunao c ° ° ° 1 1 1 ° ° ° ecco-cccclodisco lights Figures 1.5. These five basic cireuts show the versatility of the 7485. Each circuit gives @ different output sequence. Tobles 1:5. These tables show the output sequences of the corresponding circuits Figure 6. One way of isolating the contro! circuitry from the ‘tise is to use trans formers. The input to this circuit is TTL compatible. each output to the preceding input (D to C, C to B, B to A). Serial data is then entered at the D input and is shifted left on each clock pulse This may be used to make a display where the lamps light in the order A to D, then extinguish in the order D to A, as in figure 3. This circuit operates as follows: the flip-flop NI, N2 is ine itially set so that the serial input is high and the mode control is low. ‘1's are thus entered at the serial input and appear successively at the outputs A to D. When output D becomes high 1 is turned on and the flip-flop is reset. The mode control input becomes ‘1’, reversing the shift direction. Since the D input is grounded the data entered is 0”, so the outputs go low starting with the D output, When all the outputs are low the flip-flop is set and the sequence repeats. The truth table is given in table a The circuits so far discussed are similar in that after four clock pulses all lamps are lit. It is, however a simple matter to devise a circuit where the lamps light in sequence but only one at a time, by cir culating a single ‘I’ through the shift register. In the circuit of figure 4 the lamps light in the sequence A to D, with only one lamp at a time being lit. When lamp D extinguishes lamp A lights and the se- quence repeats (table 4). The circuit op- erates 38 follows: on switching on the A to D outputs will set randomly so that more than one output may be high. This would mean that a number of ‘I’s would be circulating, whereas only a single ‘1° is required. For this reason a reset button is provided, When SI is de- pressed the flip-flop comprising NS and NG is set, taking the serial input high. ‘At the same time the mode control is taken high by the other half of $1, so that on the next clock pulse the resister shifts the ‘0°s from the grounded A to D inputs to the corresponding outputs, clearing the register. When the switch is released the register is in the serialin, shift right mode, so on the next clock pulse the ‘I’ present on lektor september 1975 ~ 925 wa O5¥ 5 clock cen, Table 5 clock|A BCD 0 joooo 1 |1000 2 lo100 3 joo1o0 4 |oo01 5 |oo10 6 joroo 7 |1000 a lo1oo Ss mm © “yl iene ‘iae 400v-6A 16 JU po TUN the serial input will appear at output A. ‘The output of N2 will go low, resetting the flip-flop N5, N6 so that the serial in- put is low. On each successive clock pulse the ‘I’ on the A output is shifted ‘one place to the right until it appears at output D, When this occurs the output of NI goes low, setting the flip-flop. A ‘1 now appears at the serial input, and the process repeats. Figure 5 shows a variation on this cir- cuit, in which a ‘I’ travels back and forth from one end of the register to the other (table 5), The circuit is initially reset by pressing S1. This sets the flip- flop NI, N2 so that a ‘I’ appears at the serial input, It also puts the register in the shiftleft mode, so that the ‘0” on the grounded D input is shifted left, clearing the register. The flip-flop com-{825 — elektor september 1975 7498 7895 prising N3, N4 is reset, so that the out- put of N3 is low. ‘On releasing $1 the register mode con- trol is connected to the (low) output of N3, so the register is in the shift-right mode. On the first clock pulse the “I” applied to the serial input appears at the ‘A output. It is inverted and resets the flip-flop NI, N2 so that no more ‘I's are entered. The ‘1’ which is in the regis- ter is shifted right on each successive clock pulse until it reaches the D out- put. Flip-flop N3, N4 is then set, revers- ing the shift direction. When the ‘I’ again reaches the A output flip-flop N3, N4 is reset and the shift direction again Teverses, and so on. ‘The number of variations on these cir- cuits is, of course, limited only by the ingenuity of the constructor. Actual switching of the lamps is ac- complished using triacs. Unless the entire circuitry is housed in an insulated box the logic circuitry must be isolated from the mains. This may be ac complished by either transformer or op- tical isolation of the control circuitry from the triac, and suitable circuits are ‘given in figures 6 and 7 In figure 6 the secondary voltage of Trl is full-wave rectified and applied to the collector of the transistor via the primary of Tr2. When the transistor is turned on by one of the outputs of the disco tia Figure 7. A more elegant solution is to opto-solation. Figure 8. Two NAND gates (or inverters! ‘can be connected as an astable multivibeator. register, current flows through the primary of Tr2 and the current induce im the secondary triggers the tria choice of transformers is not critic: whilst T:2 should have 2 ratio of about 1: 6 and can be a miniature type sin only a small current flows through it. Of course Trl and the bridge rectifie are required only once in the circuit but the transistor, Tr2 and the tri must be repeated for every output fror the shift register. With this circuit trig: gering of the triac is not optimum. isolation, as in the circuit of figure 7. This is simply a conventional triac di mer with the main potentiometes replaced by a light-dependent resist housed in a light-tight enclosure with LED. When the transistor is turned the resistance of the LDR to fall. triggering point of the triac is thus ad. vanced and the lamp lights. ‘The base current of the transistor in both these circuits is about 3 mA, pends on the maximum load to switched. ‘A simple clock generator circuit is give in figure 8. This is simply two NAN gates or inverters connected as astable_multivibrator. light pattern could be altered. Figure 9 and 10 show how the discolights ma be extended with several shift registe Figure 9 is intended for extension o| figures 1, 2, and 4, and figure 10 for the extension of figures 3 and 5.\ Trane operation, is ae" oT OP partie tier esr ona TY REA cont of OP agus or of OF i i s otteee ge : Tgieento so mare sete Ges SON A cat BE Oo cate vet ag tat 200 co, THES? Ag DY IN ata nol grier OOo S00 est" gait 1 piano axione! are sven SA EOSE ION Cott ect eos = va (ne ole s\ 28 Nye OF TNE possible coral PE |ektOr Pie usel™ Dey te or TBE GB DY On SHE spol PI gtte ral nc ace THEO vinci gine? F xing P wor ‘As the gain in an OTA can be controlled by the current from an external source (the bias current 14 BC), possibilities are ‘opened up for new applications which have up to now been difficult to perform satisfactorily with discrete components. A simple application of the OTA, for example, is amplitude modulation, Al though it is basically possible to effect, this with one or more discrete transis- tors, closer inspection shows that dis- crete circuits do not achieve all forms of amplitude modulation really satisfac- torily. Tremolo (amplitude modulation of a signal which is to be reproduced acoustically) is not easy to achieve electronically without relatively high distortion or interference, Other appli- cations of the OTA such as multiplexing or sampling of signals are more success- ful than with other methods because of the OT A’s high slew rate of 50 V/usec. Automatic volume control is also an obviously attractive application for OTAs. More applications of two types of OTA, the CA3080 and the CA 3094 AT, will be given in future issues. These are the most interesting of the large range of OTAs which have been developed by RCA. The CA 3094 AT has in fact been developed from the CA 3080, and the only basic difference concerns the output circuit, Linear transconductance (forward slope) Before using the OTA in practical cir- cuits, it is important to understand the meaning of the term ‘forward slope” for which the abbreviation ‘gm’ is used, The term gm is expressed in mho (1/2) imho (71). The amplificatio: oF millimho (1 G3)- The amplification factor of a normal operational amplifier (known as a voltage amplifier) corre- sponds to the gym of a voltage-driven current source Ue. an OTA). The relationship between the output current and the corresponding input voltage of an OTA Alout = 8m x AVin. ‘The output signal of an OTA is thus a current which is proportional to its gm. The output voltage (AVout) appearing asa result of the output current, Alout, in an OTA is the product of this current and the load resistor. CA 3080 Figure 1 shows a simplified circuit of the CA 3080. Ty and Tz in figure 1 form a differential amplifier, which is also found in most normal operational amplifiers. W, X, Y and Z are known as current mirrors, A current mirror consists in principle of two transistors, ‘one of which is connected as a diode, Figure 2 gives the circuit of a current mirror of this type. As the transistors Tq and Tp are supposedly identical, a current I’ into T results in a second current I into Ty with the following relationship to 1’ In this formula @’ is the current emplifi- cation of transistors T, and Tp. A current mirror can be regarded in prac- tice as a current source in which the out- put current (I) is almost identical to the control current (I'), and in which the two currents I and I’ can in fact be regarded as isolated from one another, A disadvantage of the current mirror as shown in figure 2 is that it is sensitive to small differences in the current ampli- fications of transistors Ty and Tp, these differences resulting in the currents I’ and I not being precisely equal. This effect can be greatly reduced by the inclusion of a third transistor (Ty in figure 3), Current mirror W in figure 1 has the circuit shown in figure 2, while current ‘Simplified circuit of the CA 3080. rs Ty and T2 form the differential amplifier. W, X, Y and 2 are socal current mirrors. Figure 2. A current mirror can be simply made up with two transistors (Tq and Tp). The drive current 1” gives rise to a current | which is proportional to I’. Figure 3. The current mirror of figure 2 is tensitive to differences between the current eins of the two transistors (Ty and Tp). Addition of T; reduces this sensitivity ean tidorably.erential arpthe! mirrors X, Y and Z. are as shown in figure 3. It should also be noted that Y and Z have PNP transistors. The complete circuit diagram of the CA 3080 is given in figure 4, The circled points indicate the connection numbers in the TO-S housing. This housing, as seen from the upper side, is shown diagrammatically in figure 5. Inone of the RCA data sheets a drawing corresponding to figure 5 shows the reference tip between connections 1 and 8. This can lead to confusion: the drawing in figure 5 is correct, In the circuit shown in figure 4, and Tp are the differential input amplifier. Transistor T; is the common emitter impedance of this differential amplifier. The most significant difference from the input stage of a normal opamp is that Ty is part of a current mirror, so that its collector current is equal to the bias current (Iagc), The value of the collector current of 3 determines the emitter current of the differential amplifier Ty/T2, and this provides an effective means of controlling the overall transconductance. The gm of an OTA in normal ambient temperatures (16°C . ..27°C) is given by: (/2x 10) and Iagc in mA. In figure 4 the output signal of the OTA is taken from the collectors of T> and Tyo (connection 6) which form part of the current mirrors Z and X respectively in figure 1. As Ipc is varied, the gm of the OTA changes and therefore the output current does likewise; hence: Alout=8m * AVin = 19.2 x IABC * Vin (at normal temperatures!) ‘The OTA can easily be made to operate as a voltage amplifier by connecting a load resistor Ry, between the output and circuit earth. The output voltage then becomes AVout = RL x 19.2 x IABC x AVin in which IARC is in mA, Ry is in kQ, Vout and Vin are in volts, Characteristics of the CA 3080 Table 1 gives various important limiting values for the CA3080 and the CA3080A. The difference between these two types is related only to their working temperature ranges. In addition to these characteristics, which are for the specified supply voltages and an IABc of 500 WA, it can be said that the limit of the working frequency range is about 2MHz. The quoted input Figure 4. Complete circuit of the CA 3080 IC. ‘The circled numbers correspond to the coding of the connecting leads. Figure 5. Connections for the CA 3080 IC fare the same as on the 1JA709 except for Pins Tand 5. ‘The drawing shows the top view of the IC. Figure 6. These curves show changes in out: put current (Aloyt) plotted as functions of ‘changos in the input voltage (AVin). Figure 7. Input resistance (Rjq) a8 8 function of the so-called bias current (Tage) Figure 8. As with the input resistance, the output resistance of the CA 3080 is depen- dent on the bias current Iagc. As figure 7 ‘and this graph show, both relationships are ‘completely linear. Figure 9. The CA 3080 connected as 2 D.C.- coupled differential amplifier. Gain can be varied by potmeter Pi from about 30 to ‘about 100 times. 5 ‘Maximum ratings: DC supply voltage between +Vp and Vp: 36V Differential input voltage: tv ‘Common mode input voltage: Vp t0 Vb Input signal current: 1maA Bias current (gc) 2ma Output short-circuit duration: no limitation Device dissipstion: 125 mi Operating tempersture range: cae CA 3080 0° to 70° CA 3080 A 85° to +128°C. ‘Table 1. Characteristics and maximum rating of the CA 3080 and CA 3080 A ICs. Characteristics: (Vp 2415 Vi Vp = 15 Vv: age = 500 UA) Input capacitance: 36 pF Input resistance: 262 Input offeet current: 02 uA Input bies eurrent 2uA ‘Slew rate with unity asin: 50 VIS ‘Trensconductance (Gqq): 9600 imho Output resistance: 15 MQ Peak output current: 1500 WA Peak output voltage: positive 138 negative —14av ‘Amplifier supply current: mA Device dissipation: 30 mW| For the sake of completeness, it should elektor soptember 1975 — 929 tape Wa) Ro (Ma) — [+ Vo= 15 V,—Ve=15V (Tal = 25°C] resistance of 26k is dependent on the value of IABC. If a value of 1 MQ is chosen for the output toad resistor, the voltage gain is easy to work out from the character- istics given in table 1: AVout _ vet = LX om = 108 x 9.6 x 10° ~ 80 dB It can be deduced from this last formula that the voltage gain can also be varied by changing the load resistance. ‘The curves in figure 6 show that the overall characteristic of the CA 3080 is completely linear for small inputs to the differential amplifier. The curves show the relative values of Aloyt and the deviations from linearity as functions of the relative values of AVin. Figure 7 shows the input impedance of the CA 3080 as a function of the bias current 1agc. The maximum impedance attainable in this OTA is about 40 M2 with a bias current of 0.1 A. The output impedance is also, of course, dependent on the value of IABc. Figure 8 shows that this relationship is linear. be said that the characteristic of figure 7 also holds good for the CA 3094 AT. Figure 8 also holds good for the CA 3094 AT, but only for its current ‘output. This IC has other outputs. OTA - opamp Figure 9 shows a practical circuit for the CA 3080 from which a comparison can be made with normal opamps. ‘The power supply is symmetrical at 6 V. Both inputs are D.C-coupled and are connected to chassis earth through R; and Rp respectively. Resistor Ry of figure 9 is introduced in order to obtain. voltage amplification, as in an opamp, ‘The usual feedback from the output to the inverting input of the IC is miss- ‘ing, because the gain can be controlled by the bias current Tap at pin S. Tage is easy to calculate, While re- calling that the emitter of T; is connec- ted to —Vb (pin 4), assume that IABC is drawn via a resistor Ry from chassis earth, which is 6 V positive in relation to —Vp. The relationship then becomes: Vp - 0,7 a laBc~ In this formula, Lape is in mA, while Vb is in volts ‘and Ry is in k®. The quantity 0.7 is the base-emitter poten- tal of Ty in figure 4. To find the value Of Rx for a desired value of Tac, the formula can, of course, be rewntten: ww ¥b-07 TABC. In figure 9 Rx is replaced by the combi- nation Ra, Rg and Py. When the slider Of Py is at the positive end of its travel (ie. at OV), the voltage across. the series connection of Rx and the base- emitter junction of Ts is Vp, so that Rx tapc= Y8=97 20.53 ma. It therefore follows that the voltage gain A=RaxgmxlABC~10x19.2x0,53%100. When, however, the slider of Py is at the negative end of its travel (the junction of Rs _and Py), the voltage zelative to =Vp at the slider of Py is: RafPs x (Vp 40.7 RafP: + Rs 5x67 ite ea ‘The effective voltage across Re is there- fore: Vx 2.2V-O.7V=1LSV so the bias current is given by Tape ® 15 ma = 0.15 ma. ‘The voltage gain is therefore: A®R3xemalABC¥I0x19.2x0.15=29X If the gain of this IC is allowed to drop substantially, considerable distortion may result unless special attention is given to the design of the differential inputs. Should the input transistors Ty and T2 (figure 4) not be exactly Matched, their emitter currents will differ when Iapc is low, and this will cause distortion. In this connection, the following rules of thumb apply: a. If the OTA gain is fixed, resistors Ry and Rz (figure 9) must have values which are lower, by a factor of at least 2 : 1, than the value of input impedance read from figure 7 for the relevant value of LABC. Ry and Rz must have the same value when IBC is less than about 0.5 4A. For fixed gain with values of TBC between I WA and 10 UA, the values of resistors Ry and Rj may differ by, a factor of 2. When [ABC is over 10A, Ry and Ra may differ in value by a factor of 4. . If the gain is to be variable over a range greater than I : 5, resistors Ry and R must have values lower, by a factor of at least 2, than the1930 — elektor september 1975, 10 1609 10 1 value of input impedance, indicated by figure 7, for the maximum IARC. Negative feedback Negative voltage feedback can be used with an OTA as it can with a normal opamp. Figure 10 gives an example of a circuit for a CA 3080. The bias current IaBc is determined by Ry. The poten- tial across this resistor is the negative power supply (15V), less the 0.7V base-emitter voltage which was discussed in relation to T3 of figure 4. In this case ‘the value of bias current is given by: 15 -0.7 4 108 “he TaBC so that gm works out at 19.2 x [apc = 2.74 mmho. ‘The voltage gain given by a CA 3080 in the crreuit shown in figure 10 is not determined solely by the value of the load resistor Rs, but also by Ry and Rs. In the first place, the effective output load resistance is Rs and (Rs + Ry) in parallel. In the second place, the voltage developed at pin 6 across this effective ‘output load is fed back to the inverting input (pin 2) with a step-down ratio Ri/(Ri+Rs). The effective voltage gain between the input and pin 6 is thus: m-RL am + [Ri + Rs )/Rs] 1 +m [(Ri + RaWRs] + pe 8m + Rs Veen Reo ay Ry Ret It can be seen from figure 10 that if Rs is omitted there will be no voltage fed- back from the output. This is equivalent to making R; infinitely large in the foregoing calculations, and results in in the voltage gain being increased by a factor of about 8. (fis the feedback factor If a comparison is now made between the circuit of figure 10 and a normal ‘opamp, such as the HA 741, a number of similarities become evident. Both the OTA and the opamp can be operated as voltage amplifiers, and voltage nega- tive feedback can be used with either. Both can have either symmetrical or asymmetrical inputs, inverting or non- inverting. The OTA, however, becomes a pure current source when ‘it has no load resistance; a feature which can be advantageous for some applications. Besides this, the OTA has the feature that, as the transconductance is varied by varying the bias current, the input and output impedances also vary over Toble 2. Cheractoristies and maximum ratings of the CA 3094 AT. ‘Maximum ratings: DC supply voltage be- ‘ween +Vp and —Viy: 36V Differentia ingut voltage: 46v ‘Common mode input voltage: ++Vpt0 Vp Input signal current: 1mA Bias current (1AgC) ama ‘Output current: peak: 300 ma ‘averag 100 ma without heatsink: 630 mw with heat sink: 18 Peak dissipation (1 mS): iow Operating temperature rome: 88° 10 +125°C Charactristes: (+¥p = 15: Vp = -18V: Tage = 100 HA). input cpecitance: 2.6 pF input reistance: (ase = 208) 1M Input offaet current 0.02 ua Input bias curront: 0.2uA Deviee dsipation 10 mw Bandwidth (Unity goin): 90 MHz Amplitir bias voltage: 0.68 V ety 12 - @ 8 ° 9 On ° ® ° Ole Q omeote Figure 10. This circuit incorporates negative feedback from the output through Rj to the inverting input, Figure 11. A CA 3080 connected as an A.C.- coupled “asymmetrical amplifier. Negative feedback is taken from the emitter of Ti ‘through R to the inverting input of the IC. Figure 12, Detailed circuit of the output section of a CA 3994 AT OTA. The difference ‘from the CA 3080 consists of the addition of resistors Ry/R2 and transistors Tia/Ti3- Figure 13. Functional diagram of the CA 3094 AT. The corresponding output of ‘the CA 3080 (Pin 6) is connected in this caso wo Pin 1. a wide range. If it is important for a particular application that one of these parameters (but not the other two) should have a specific value, it can be adjusted to this value by controlling the bias current. Yet another feature possessed only by the OTA is that the gain can be con- trolled as may be required by D.C, or A.C. potentials, thus making amplitude modulation, sampling or switching functions possible, Output buffer stage : Table 1 shows that the peak output current of the CA 3080 is only 500 uA, and this can be a drawback in a number of applications; moreover a ‘power’ OTA such as the CA 3094 AT costs twice as much. A simple solution is given in figure 11, which shows a buffer transistor follow: ing the OTA. By this means the output current (Aloyt) of the OTA is multiplied in the same ratio as the current amplifi- cation of Ty. Another advantage accru- ing to the addition of Ty is that, being an emitter follower, its output im pedance is low. The load impedance which the OTA ‘sees’ at its output is equal to the values of Rs, Re and the input impedance of T;, all in parallel. The fact that the CA 3094 AT has been developed shows that RCA themselves have, indeed, given thought to the need for higher output currents. This OTA is equivalent to the CA 3080 except for the addition of two resistors and two transistors. Figure 12 shows in detail the output circuit of a CA 3094 AT. ‘A comparison with figure 4 shows that R,/Ro and T;/T have been added in figure 12. Some of the characteristics of the CA 3094 AT are given in table 2, and the connections are given in figure 13. Pins 8 and 6 become power ‘output points for ‘sink’ or ‘drive’ cur rents respectively. The low-power out- put, which is at Pin 6 in the CA 3080, 4s ‘brought. out at’ Pin 1 in. the CA 3094 AT. “ {ish feeder F. Sax lektor september 1875 — 931 The store of dried feed is held in a trough with V-shaped sides (figure 1). At the bottom of the trough is a cylin- drical container which runs the length ‘of the trough, and which has one side ‘cut away. This cylinder is driven, via a reduction gear, from a small model motor. As the container rotates it will fill when the open side is uppermost, and empty into the aquarium as it ro- tates. The number of revolutions made at each feeding session, and hence the amount of food delivered, is controlled by the electronic circuitry. A cowl at the bottom of the dispenser prevents splashing caused by the fish or the aer- ator from making the feed sticky and thus clogging the dispenser. The circuit In figure 2, TI is an emitter-follower whose base potential is controlled by a light dependent resistor R1 and a potentiometer PI. This is followed by a ‘Schmitt trigger, T2 and T3, which has a large degree of hysteresis. This drives T4 via R9 and zener diode D2. During dark- weg e ness the resistance of the LDR is high. The emitter potential of TI is therefore high, T2 is turned on and T3 is turned off. Hence T4 is also turned on, When daybreak comes the resistance of the LDR drops, the emitter potential of T1 drops, and when the switch off threshold of the Schmitt trigger is reached T2 turns off and T3 turns on, T4 therefore tums off. Point A goes up to supply potential. The switch-on threshold at daybreak can be adjusted with P1. The hysteresis of the Schmitt trigger is so great that even large brightness variations during the day will not cause spurious trig- gering. However, care must be taken to ensure that the LDR is screened from room lighting so that spurious trig- gering does not occur in the evening. ‘The motor control circuit is shown in figure 3. When T4 switches off at day- break, TS tums on. This shorts out the base of T6 through C2, and T6 turns off until C2 has charged sufficiently through R15 and P2 for Té to turn on again. During this time T7 and T8 are tumed on and the motor runs, The charging rate of C2, and hence the motor running time, can be adjusted by P2. In the evening when T4 turns on, TS tums off but this does not affect the state of the following stage, so no feeding occurs. If the fish feeder is to be used other than at holiday times the triac switch of figure 4 may be used to control the aquarium lighting, It is important to in- clude C3 across the choke of the fluor- escent tube to avoid high voltages being applied to the triac. If the lighting cir- cuit is connected to the automatic feeder it is imperative to ensure that the finished construction is adequately in- sulated as the ground connection of the fish feeder is connected to the mains neutral. No part of the circuit should be accessible, and in particular the motor should be insulated, including the drive shaft, Potentiometer P2 should have a plastic shaft, and the whole assembly should be mounted in a plastic box, ‘with no metal protrusions, a|| $32 — sloktor soptomber 1975 fish feeder Partition Dimensions, b and c depend on the daily quantity of feed Rotating container Construction of the dispenser 2 The dispenser is probably best con- structed of clear acrylic sheet, so that the level of food may easily ‘be seen, This may be glued together with acrylic cement. Motors with suitable reduction gearboxes can be obtained from most model shops 0125 delay fuse 0s | ‘ waetristoble elektor september 1975 ~ 933) J. Koper Fj n —# Jar peer 13, @ \aa gp 13; ugy —2 |ar {ewe It is possible to use two NAND or | [> NOR gates to make up a flipflop a circuit with two stable con- ditions. The process can be extended to obtain circuits with three or even more stable states. 126 a 12g & ae rig——J ay n ra r3g— KS r3g— ae ‘The arrangements shown in figure 1 both have 3 stable states. The state taken up by the outputs will depend on | [ 4 1 the input conditions applied. These cit- cuits have the objection that correct | | 1 operation is only guaranteed when | | drive is applied to two of the inputs 5 at once (see table 1). It is however possible to modify the cir- | |_NAND gates NAND gatet cuits so that a single input drive will | [7], 15 produce the desired output state. The | | circuit as a whole becomes more exten- | | 0 sive; but it becomes easier to use. | | 0 Figure 2 shows the modified arrange. | | 1 ment. The operation of this circuit can be followed from table 2. Tobie Table? Figure 3 shows a master-slave shift regis- ter. If C is at logic ‘1’, then the OR-gate 3 outputs will also be ‘I’, so that the state of Qr02Q5 does not change. If C gels de becomes ‘0’, the AND-gate outputs ra will also be ‘0’, so that the state of N-It Q+-Q5-Q6 is held Suppose for example that C is logic ‘0’, ole with [1 = ‘0, I ‘0" and [3 = ‘I’. We find that Qy = ‘1', Qa= “I and Qa= ‘0’. 2-12 Since C is ‘0” the state of Q¢-Qs-Q¢ is maintained. If C now goes to ‘I’, the outputs Q1, Q2 and Qs will not change. = “ata wars 08 This state is also the state at the inputs a3-z3-4 Ia, Is and Is. Qs therefore becomes ‘0", Qs also ‘0’ and Q¢ ‘1’. This shows that ce the input information ‘0'-0'-1" appears, after one clock pulse, at the output. e+00x9 eo+oxg en+0xp eno-x9 oo--xg eoo=Kg soonos cece a x-o+-9 xaro-g xoon+9 xo-o-9 x-co-p Bos$34 ~ elektor september 1975 L. Wiechers roll out the bandit This oscillator was designed for driving electronic games of chance, such as the ‘three-eyed bandit’ (Elektor No. 2 page 238) ‘or an electronic ‘roulette wheel’. The disadvantage of the ‘three- eyed bandit’ is that a stop button must be pressed to stop the three oscillators and obtain the final display. Thus the tension and anticipation obtained with a real one-armed bandit as the number drums slowly grind to a halt is missing. The voltage-controlled oscillator overcomes this by pro- viding an output whose frequency slowly reduces until it finally stops. This can also be used to simulate the ‘rolling out’ of the ball ina roulette wheel. The circuit is based on the well-known astable multivibrator (figure 1), The fre- quency of oscillation of this circuit is determined by the charging current into Cl and C2 through Rp2 and Rp) when either of the transistors is in the cutoff state. The multivibrator can be turned into a simple VCO by connecting RB} and Rg? to a seperate supply instead of to +Vj. Increasing the voltage applied to the base resistors will increase the charging current through them into Cl and C2, and will hence increase the fre- quency of oscillation. Reducing the voltage has the opposite effect. There are however, certain limitations to this simple approach. The base res- sistors perform two functions. When (for example) TI is turned on RB) sup- plies its base current. The minimum cur- rent must be such that the transistor remains in saturation. This places a restriction on the minimum voltage that may be applied for a given base resistor. When TI is turned off and 72 is turned con, Rp1 supplies the discharge current for C2 whilst RI supplies the charging current for Cl. For correct operation CI must have charged up to almost +Vip before TI tums on again. This means that C2 must discharge more slowly than Cl charges, and this limits the ‘maximum control voltage that may be applied to the base resistors. In practice frequency changes of between 10 : 1 and 50 :1 can be achieved, depending om the gain of the transistors. This is in- sufficient for this application, Base current feed through zener diodes ‘The limited frequency range can be ex- tended by the circuit of figure 2. Nor- mally the coupling capacitors supply a portion of the base current whilst they are charging from the collector resistors, but this ceases as soon as the capacitors are charged. The zener diodes perform two functions: 1. they limit the voltage to which the coupling capacitors charge, and hence the time required for charging. 2. they provide a D.C. path for the transistor base current, even when the capacitors have charged. This means that the base resistors only provide the discharge current for the capacitors with the transistors in a cutoff condition. They can thus be much larger. Also, since one transistor is always cut off, only one base resistor is required (Ghown dotted in figure 2) provided the transistor bases are isolated by diodes. When this circuit is used as a VCO by connecting Rp to a control voltage @‘oll out the bandit lektor soptember 1975 — 935, Figure 1. A basic astable multivibrator. This may be voltage-controlled by connecting the base resistors to 8 variable voltage instead of +Vb. Figure 2. Addition of zener diodes makes transistor bate current almost independent of base resistors. The two hase resistors can bbe replaced by 8 single resistor and two diodes. Figure 3. Base resistor replaced by a voltago- ccontrolied current source. Figure 4. Addition of emitter followers im- proves risetime without sacrificing loop gain. ‘T6 further improves rsatime and drives TTL. frequency range of between 200 : 1 and $00 : | is obtainable. Decaying frequency characteristic ‘The next step is to achieve the gradual | decay of frequency required. This is | accomplished in the circuit of figure 3 When the pushbutton is pressed C3 is charged rapidly. The voltage on C3 turns on T3 which causes the oscil- lator to start. As C3 slowly discharges the collector current of T3 decreases and the oscillator frequency reduces until the voltage across C3 is less than about 0.6 V, when T3 cuts off and the oscillator stops. The final circuit Figure 4 shows the final circuit. Emitter followers T4 and TS are incorporated to provide a low impedance charging path for the coupling capacitors, thus improving the rise time of the waveform without reducing RI and R2, which would reduce the loop gain. T6 converts the output to a level suitable for driving TTL circuits. ‘The discharge rate of C3, and hence the ‘rolling out’ time, is adjusted by P1. The initial frequency of oscillation is ad- justed by P2. Note that Cl and C2 should be non-electrolytic types. With the component values shown the results obtained were as follows Starting frequency 100 to 300 Hz, Final frequency about 0.3 Hz. ‘Rolling out’ time 25s maximum. x This simple triac dimmer can be used to control incandescent filament lamps up to 1500 W. The circuit operates on the phase-control principle. The main con- trol is provided by P2. This determines the rate at which C2 charges and hence the point along the mains waveform at which the voltage on C2 reaches the breakdown voltage of the diac, which is when the triac is triggered. PI, in con- Junction with R1 and C1 determines the ‘minimum brightness level, or alterna tively may be used as a fine brightness control. Interference suppression is provided by R2 and C3. Construction The printed circuit board is very com- Pact and can easily be accommodated inside the modern, square type of flush- mounting switch ‘panel, or in a small box for portable applicatigns. The following safety points should be noted. No part of the circuit should be access. ible from the outside. The case should preferably be made of plastic or other insulating material, and fixing screws for the board should be nylon. If a metal case is used the board must be ad- equately insulated from it and the case should be earthed. The potentiometer should have a plastic spindle, TH = 40669 1RCA\| $38 — elektor september 1975 HL. Krielen . fe] © Oo char Q) | © tock errator © 1 © The dual-slope technique is one of the simplest and most reliable DVM sys- tems, The voltage to be measured is fed to an integrator for a fixed period of time. The current into the integrator, and therefore the charge on the inte- gator capacitor at the end of this period, is proportional to the input volt- age. The capacitor is then discharged at a (known) constant current and the time taken for the capacitor to com- pletely discharge is measured. Since the ischarge current is constant the time faken to discharge is proportional to the original charge, which in turn is pro- portional to the input voltage. The dis- charge time is measured by feeding clock pulses from an oscillator to a digi- tal counter until the voltage on the ca- pacitor reaches zero. The same oscillator is used to determine the original charge time. ‘This means that any long-term variations in the oscillator frequency are unimportant since they will affect both the charge time and the measured dis- charge time equally. The long-term stab- ility and absolute frequency of the oscil- ator are thus unimportant. The only reference standard in the DVM is the constant discharge current, which must be stable. Looking at the system mathematically: Current into integrator I = V/R, where V is voltage to be measured and R is in- Figure 2. The input voltage applied, igure 3. The time axis. The time ty-t2 is the charging time, £23 isthe discharge time, and {atta it the display time, Figure 4. Output voltage of the reset pulse (onerator. Figure 5, The input voltage to the integrator. Figure 6. The discharge current of the inte- ‘rator. Figure 7. The output voltage of the intogrator. Figure 8 The output voltage of the zero: crossing detector. Figure 9. The clock pulses applied to the ‘counter, They first consist of 100 reference: pulses, which determine the charging time, followed by 2 number of pulses proportional 10 the voltage of the input signal. The fre- ‘quency of these pulses can be assumed t0 be: ‘constant during the time ty-19- Figure 10. The clock generator. Via line A the pulses go to the count gate, whilst via line B ‘the reset pulse generator is driven, Figure 11. The reset pulse gonerator. Thie supplies reset pulses to the counter (via C and D), and a start pulse to the control logic (via e). tegrator input resistor. Charge on integrator capacitor at end of charge period At, v R Time taken for eapacitor to discharge at constant current Is =0_Vaty a HR Since At, and At, are derived from the same oscillator it can be seen that a vari ation in oscillator frequency will affect both At, and At, equally, and the final result will remain the same, provided Ty does not change and R is fixed. Q=hAn = 2+ At: Atedual stope dv The block diagram of the DVM is given in figure 1 and an operational timing diagram in figures 2.9. The timing dia- gram is drawn for both a positive and a negative input voltage ‘The sequence of operation is as follows: the input chopper applies the input volt- age (figure 2) to the integrator for a fixed time ty ta (figure 3). The chopped input to the integrator is shown in fig- ure 5. During this time the integrator output rises linearly as the capacitor charges (figure 7). The step in the inte- grator output waveform at the beginning and end of the charge period is ex- plained in the detailed description of the integrator later in the text. At the end of the charge period the inte~ grator is disconnected from the input voltage and is connected to the dis- charge circuit. The integrator capacitor ischarges linearly during the period t2- ts (figure 6). During the whole period ty-ts the output of the zero- crossing detector (figure 8) is positive When the voltage on the integrator ca- Pacitor reaches zero the output of the zero-crossing detector falls to zero. This is used to control the clock pulses to the counter (figure 9). The operation is ef- fected by the control logic in the block diagram. Before each measuring period the counter and control logic are reset by a pulse from the reset oscillator (figure 4), Measurement of a negative voltage is performed in a similar manner. The only differences are that the output of the zero-crossing detector is negative. This is detected by the polarity detector and is used to reverse the polarity of the con- stant current from the discharge circuit. (Otherwise the integrator output, being already negative, would simply become more negative and would never cross zero.) ‘A refinement is incorporated in the form of a drift compensator circuit, which nulls out the effect of zero drift in the integrator and zero-crossing detec- tor. Circuits in the DVM Clock Generator The clock generator, which provides drive pulses for the counter, is shown in figure 10 and consists simply of a two- transistor astable multivibrator with a slektor september ‘st measurement 100pulses ‘and measurement frequency of approximately 15 kHz. As stated earlier, the long-term stability of this oscillator is unimportant. The Reset Pulse Generator This circuit is shown in figure 11 and is based on a programmable unijunction transistor, TI4. The gate of this device receives a D.C. bias from R42 and R43. Pulses from the clock generator are applied to point (B) and charge up C7 through D1O and R41. When the uni- Junction fires C7 discharges through the unjjunction and R44, and the voltage across R44 causes TIO to tum on. This causes TI1 to tum off. A negative- going pulse is therefore available at the collector of TIO and a positive-going pulse is available at the collector of TH ‘The time between reset pulses is deter: mined by the time constant R41 x C7, and in this case is one second. The inter- val between reset pulses determines the measurement repetition rate and also the time for which each reading is dis- played. It may be altered to suit per- sonal taste, provided it is longer than the measuring period ty -ts C7 should be a low-leakage type, prefer- ably tantalum. The Counter ‘The counter circuit (figure 12) consists of two 7490 decade counters and a JK flipilop (half of a 7473). The 7490's count the two least significant decades and drive 7447 seven segment decoders and LED or Minitron displays. The JK flipflop counts the ‘hundreds’. Since the maximum display is 199 only a one need be displayed by the hundreds dis- 101938 — elektor soptember 1975 dual slope 12 FFI Ha 7473 ooo play. For economy, a seven segment dis- play is not used but simply two LED's in series, The other half of the 7473 is ‘used in the control logic. Clock pulses to the counter are gated by a two-input NAND-gate (quarter of a 7401). Input Chopper The input chopper (figure 13) connects the input voltage to the non-inverting input of the integrator during the charge period, ty-t2. For the rest of the ‘measurement cycle it grounds this input. The circuit functions as follows: during the interval ty-tz point His at logic ‘I’ (+5 V) so T! is turned off. T2 is also tumed off, The gate of F1 is held at about —10 V so FI is cut off. The gate of F2 is held at about ~1.2 V, so F2 is tumed on. The input voltage therefore appears at point! via the FET F2, and is thus fed to the input of the integrator. At time t; point H becomes low, so TI and T2 are both tured on. The gate voltage of F1 becomes about —2 V, so it conducts and grounds the input of the integrator. The gate voltage of F2 be- comes about —10 V, so it is cut off and the input voltage is disconnected from the integrator. The Integrator The integrator of figure 14 serves to establish a voltage-time relationship, ic. the number of clock pulses counted must be proportional to the input volt- ‘age. The circuit operates in the follow- ing manner: to achieve a reasonably high input im- pedance without additional buffer amplifiers a non-inverting integrator configuration is used. When the input voltage is applied to the input I by the input chopper, the output of the 741 will swing positive. Since C2 appears as a short circuit to this step input there is 100% negative feedback through C2. ‘The output voltage of the 741 must therefore assume the same value as the voltage on the inverting input (pin 4), which by definition is the same as the input voltage on pin 5. The input volt age thus appears at the output as a positive-going step. Since there is now a voltage across R12 a (constant) current flows through it which is proportional to the input voltage. Since no current can flow into the inverting input of the 741 this current must flow into C2. Since the current is constant the charge ‘on the capacitor, and therefore the volt- age across it, increases linearly. The capacitor is allowed to charge for a period of 100 clock pulses. The voltage across C2 is then I, + At_ Vin = At G Ra-G where At represents the time interval ty- tb At time ty the input chopper discon- nects the input voltage and grounds the non-inverting input of the integrator. This causes a negative-going step, which cancels out the earlier positive-going step. The voltage on the inverting input of the amplifier is now zero, so the volt- age across C2 is the same as the output voltage. When the discharge circuit is connected to point J (the inverting input) the inte~ grator begins to function in the in- verting mode. The discharge circuit sup- plies a constant current Iz into the ca- pacitor of opposite polarity to the charging current. The capacitor thus discharges linearly. The voltage on the inverting input is, by definition, zero, so as the voltage across C2 falls so does the v Figure 12. The counter. C and D are reset i puts, Ais the count input, F the driver for ‘count gote. Output G supplies « pulse to control logic at the one hundredth pulse. Figure 13. The input chopper. It is driven respectively. C2 is the integration capacitor. PI serves for zero-adjustment: with input | arth and the lines J and K interrupted, ‘output L must be adjusted to O with thi potentiometer. Figure 15. The zero-crossing detector. amplifies the output voltage of thei Wine L), and drives the drift compensator the polarity detactor (via M and N). Figure 16. The polarity detector. This is fact a three-position switch: for input volt higher than +600 mV output 0 is low’ and ‘high'; for voltages between +600 mV 600 mV both outputs are ‘high’; whilst ‘voltages below —600.mV 0 is high and P i Tow. Figure 17. The polarity indicator. It drives jot lamps, depending on the polarity of Input signal during the measuring peri Figure 18. The discharge circuit. This ‘switched on vie line $ or T from the con logic, and ensures that the integration ppacitor Is discharged via Tine J. The DVM calibrated with adjustment potentiometer fer positive, and with P3 for negative i voltages. Both are adjusted until the cou indieates one unit per milfivolt.741 output voltage (which During this time the counter is counting clock pulses, until the zero-crossing detector monitors zero_volts_ on the output of the 741. The discharge time Atx is given by: voltage on C2, y+ At_Iy* Aty G G therefore but Vin 1, = Yin Rv therefore Vin + At “eae Ruk Since At, R12 and I; are all fixed Aty is proportional to Vin The Zero-Crossing Detector This circuit also uses an op-amp (fig- ure 15), but in this case a 709 is used which has a greater slew-rate than the 741. The circuit has a high gain, about 70 x, so a small swing of the input volt- age positive or negative will make the output swing hard over to plus or minus 10 V. D5 and D6 provide input protec- tion by limiting the voltage on pin 5 of the IC to about +0.2 V maximum, and R23 limits the current through the diodes. C3 and C4 are included to keep the 709 stable. ‘The output of the zero-erossing detector is connected to the input of the polarity detector (figure 16). For outputs from the zero-crossing detector greater than +0.6 V T6 is turned on and T7 is turned off, while for outputs more negative than -0.6 V T7 is turned on and TO is tured off, For voltages between —0.6 V and +0.6 V both transistors are turned off, thus providing a zero indication. Polarity Indicator This consists of two high power NAND- gates (7440), connected as a set-reset flipflop (figure 17). During —_ the measuring period this flipflop is either set or reset by the polarity detector de- pending on the polarity of the measured voltage and the appropriate LED is lit. The flipflop is necessary to store the polarity indication during the display period, when the output of the inte grator (and hence of the zero-crossing detector) is ze. The Discharge Circuit There are in fact two discharge circuits, one of which is used depending on the polarity of the input signal. The circuit is shown in figure 18, When the input signal is positive, the output of the 741 in figure 14 is positive, and C2 charges so that the ‘right-hand’ end is more positive than the ‘left-hand’ end. This means that to discharge the capacitor the output of the integrator must be negative-going during the dischargepee — sickstor coptoentbor 1875 period. Current must therefore flow into point J. For a negative input signal the ‘output of the integrator is negative, so the output must be positivegoing during the discharge period. Current must therefore flow out of point J. The discharge circuits operate as fol- lows: when the input signal is positive point T is grounded by a control signal from the polarity indicator via the con- trol logic while point S remains high, T4 and TS are therefore turned off, whilst 13 is turned on. +5.6 V therefore ap- pears across ZI. The voltage applied to R13, and therefore the current through R13 into the integrator, can be adjusted by P2. When the measured voltage is negative, point T is ‘high’ and point $ is ‘low’. T3 is tumed off and T4 is tumed on. =5.6 V appears across Z2 and the cur- rent through R14 can be adjusted by P3. The Drift Compensator To prevent zero drift in the integrator and zero-crossing detector from causing It ie switched on via line U, and provides @ feod- bback from the output of the zero-crossing detector to the inverting input of the inte- srator. igure 20. The control logic. It receives signals from the reset pulse generator (E), the counter (G) and the polarity detector (0 and P). It drives the input chopper (H), the count ‘gato (FI, the discharge circuit (S and T), the polarity’ indicator (Q and R) and the drift ‘compensator (U). Figure 21. The overall diagram. P1 serves for ‘zero adjustment of the intogrator (see fig- ture 14). P2 and P3 serve to calibrate the DVM. {see figure 18). 13 20 @ ® on a ics io bus wm: O—Ts. ait z (sams 7401 @ rae 2 eas : Let—e at E Lies Lily —<@ A ‘a L@® ¢ inaccuracies feedback is applied round these circuits during the display period. During this time point U is low, so T13 is turned on and hence F3 is conducting (figure 19), Points M and K are connec ted to the output of the zero-crossing detector and the inverting input of the integrator respectively, so any voltage offset on the output of the zero-crossi detector will be integrated, which will tend to null out the offset. During the measuring period point U is ‘high’, and the drift compensator is switched off so that the integrator and zero-crossing detector can function normally. Control logic The measurement sequence timing i performed by the control logic, the ci cuit of which is given in figure 20. The measurement sequence starts with positive pulse from the reset pulse gen erator, which resets the counter vi point D (figure 12). This pulse is applied to point E of the control I and on the trailing edge of the pulse th JK flipflop (4IC3) is set. The Q output connected to line H switches on the input chopper, whilst the 0 output “low” and sets the set-reset flipflop con- sisting of two NAND-ates. Output F thus goes ‘high’, opening the gate to the counter, so that it begins to count clocl pulses. The drift compensator is also switched off via line U. A negative going pulse presets FFI in figure 12 via line C. ‘The Qoutput of the 7473 holds the inputs of gates A, and Az low via D8, which holds both inputs to the dis: charge circuit (S and T) high. The dis- charge circuit is therefore inoperative. When 100clock pulses have been counted (time t,) output G in figure 12 goes ‘low’, resetting the JK flipflop. (41C3). The input chopper is now switched off via line H. In the meantime the flipflop comprising As and A, has been either set or reset by the polarity: detector. Since the Qoutput of the 7473 is now ‘high’ the outputs of Ay and Ag can be gated through Ay and. ‘Az to set the polarity indicator vie lines Q and R, and also to enable the appropriate part of the discharge cir- cuit. The counter continues to count clock pulses. Note that it is not necess- ary to reset the counter at time ty, as at the hundredth pulse it has reached zero! When the output of the integrator reaches zero the output of the zero- crossing detector is also zero. Both out- puts of the polarity detector go ‘high’, so the output of gateBs goes low, resetting the flipflop (B; and Bz), which disables the discharge circuit vis D7 and Ay, Az. The counter gate is closed via line F so the count ceases. ‘The display now indicates the measured value of the input voltage until the next reset pulse.lektor soptembor 1975 ~ 941942 — elektor september 1975 A. Schulz The principle of a LEP can be compared to a light-activated switch. The light source, however, has been replaced by the light pistol. By pulling the trigger of the LEP a short flash of light is emitted. This flash of light is produced by a lamp built into the barrel of the pistol. A microswitch operated by the trigger, connects a previously charged capacitor across the lamp. Via this low- resistance load the capacitor discharges quickly. As a result the lamp lights momentarily. To increase the intensity the lamp is briefly loaded with three to four times its nominal voltage. Moreover, the light flash is focussed by a biconvex lens into a parallel beam. Figure 1 shows the circuit disgram of the flash circuit. One possible practical realisation is shown in figure 2. The. target has a light sensitive bull's eye. ‘The circuits for hit and time indication are situated behind the target. Block diagram Figure 3 shows the block diagram of the hit and time indicator. A start-stop-oscillator (block 1) supplies the counting pulses for the timer. This timer (block 2) indicates the time be- tween start and hit with a resolution of 1/100 second, If the bull's eye is not hit wi 9 seconds, the change-over from 8 to 9 is used by means of an AND-gate (block 3) to start 2 monostable multi- vibrator via block 4. The latter blocks the start-stop-oscillator for 2 seconds. After these 2 seconds the monostable resets, As a result the oscillator is started simultaneously with a second mono- stable (block 6). The second monostable resets the counter. By means of this second monostable the counter is maintained at zero for one more second, so that the total inter- val time is about 3 seconds. Figure 4 shows the diagrams of the pulse trains a, b and c, After this interval time the counter starts all over again, Figure 1. Circuit diagram of the flash circuit. Figure 2. The practical construction can be secommadated inside the pistol. Figure 3. Block diagram of the LEP. Figure 4, Timing diagram of a: the output ‘of MMI, b: the output of MMZ and c: the ‘the oscillator output. Figure 5. Circuit of the “LEP complete with timer and traffic light’.ee This counting cycle can be interrupted only by a direct hit on the light sensitive resistor (block 7). Then a pulse is pro- duced which via the OR-gate (block 4) triggers the monostable multivibrator (block 5) so that the oscillator is stopped. Diagram Figure 5 gives the overall diagram of the hit and time indication. The indi- cation unit uses three cascaded 7490 decade counters with 7447 decoders driving Minitrons and needs little further discussion. The minitron has been chosen for the display. Any other seven-segment dis- play could, however, be used instead ‘The counter input is connected to an astable multivibrator, This multivibrator is formed by two nand gates (NS and N6) with frequency-determining el- ements. ‘A control unit starts and stops the oscillator, also resets the counter, An important part of the control unit is the hit detector built around TI Transistor TI is adjusted so that it is normally conducting. The voltage div- ider consisting of Ri and R2 coupled by a capacitor can, however, briefly influence the bias of T1. When a light flash hits R2, the base of T1 will be briefly grounded. As a result TI blocks and by means of the trigger circuit consisting of NI and N2, a short pulse going from ‘I’ to ‘0’ is generated. This pulse is fed to a NAND (N4) which triggers the monostable multivibrator MM1 connected behind. The sensitivity of the trigger circuit can to some extent be adjusted with P1. NAND Né is used as an OR-gate here! ‘As long as the output of the Schmitt lektor september 1975 — 943 iS ees R | coe | stop [ exitator OID OOO sin ce} we vet trigger produces no pulse, it remains at the logic ‘I” level. The same applies to the output of N3. This output remains ‘I’ until the last counter has reached position 1001. This happens after 9 seconds. As soon as the position 1001 is reached, the output of N3 changes to ‘0’, so that MM1 is triggered via N4. From then on it is impossible to trigger ‘MMI from the hit indicator, ‘The triggering of the monostable multi- vibrator. MMI causes the oscillator formed by the gates NS and N6 to be blocked, because one input of N6 is connected to the Q-output of MMI. At the same time the set-reset flipflop (N7 plus N8) is reset by the Q-signal on the input of NB. As a result the output of N8 becomes ‘1” so that the red lamp, connected in the collector circuit of T2, ser=ni tisdda | waded | a= 7400 raves aha ef re oh as es eves eo ek ab 72 Tas 2N2218,aNIBIS‘944 — sloktor september 1975 lights up. The output of N7 goes to ‘0’, so that the green lamp in the collector circuit of T4 extinguishes. The cireuit uses a kind of ‘traffic light” as an indicator, which gives the start signal (green) and also indicates either that a hit has been made or that the playing time is over (red). ‘The cycle time of MMI is about 2 seconds. MM2 is started on the trailing edge of the output signal of Q of MMI (that is after two seconds). The Q-output of MM2 then becomes ‘I*. This causes the amber lamp of the traffic lights to light up. The set-resex flipflop is reset ‘on the trailing edge > the Q-signal. The red and amber lamp 2

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