350 IEEE MICROWAVE AND WIRELESS COMPONENTS LETTERS, VOL. 19, NO.

6, JUNE 2009

Dual-Band Bandpass Filter Design

Using a Novel Feed Scheme
Xiu Yin Zhang, Student Member, IEEE, Jin Shi, Jian-Xin Chen, and Quan Xue, Senior Member, IEEE

Abstract—A planar dual-band bandpass filter based on a novel

feed scheme is presented in this letter. The proposed filter em-
ploys two sets of resonators operating at diverse frequencies to gen-
erate two passbands. A novel scheme is introduced to feed the res-
onators. One set of resonators is utilized to not only generate the
lower passband but also feed other resonators. Source-load cou-
pling for upper passband is inherently realized. This scheme pro-
vides sufficient degrees of freedom to control the center frequen-
cies and bandwidth of the two passbands. Four transmission zeros
can be created close to passband edges, resulting in high skirt se-
lectivity. To validate the proposed idea, a demonstration filter is
implemented. The design methodology, filter sensitivity and exper-
imental results are presented.
Index Terms—Bandpass filter (BPF), dual-band, feed scheme,
source-load coupling.

I. INTRODUCTION
Fig. 1. Filter configuration.

W ITH the ever-increasing demand for dual-band wire-

less communication systems, dual-band bandpass fil-
ters (BPFs) are highly desired. To meet the need, much research
yields the lower passband but also feeds other resonators with
source-loading coupling inherently realized. Using this scheme,
work has been performed and various design approaches have it is convenient to obtain desired external quality factors and
been proposed [1]–[8]. Among them, two categories of methods coupling coefficients at two passbands. Consequently, passband
are very popular. The first category is to utilize a resonator’s fun- performance requirements can be easily fulfilled. This scheme
damental frequency and second harmonic to generate two pass- uses planar structure, facilitating the design and reducing
bands [1]–[4]. A typical example is to use stepped-impedance fabrication cost. In addition, four transmission zeros can be
resonators (SIRs). By controlling impedance and length ratios of realized. They are close to passband edges and greatly improve
SIRs, desired operating frequencies can be obtained. However, skirt selectivity. Based on the proposed idea, an experimental
it is not convenient to meet specific bandwidth requirement. filter is implemented operating at 1.8 and 2.4 GHz for DCS and
The second category is to combine two sets of resonators with WLAN applications. Predicted responses are well confirmed
common input and output ports [5]–[8]. Using proper configura- by the measured results.
tions, the required external quality factors and coupling coeffi-
cients can be obtained at two passbands. Although the structure
is relatively complex, it is easy to meet performance require- II. TOPOLOGY AND MECHANISM
ments of two passbands, i.e., center frequencies and bandwidth.
However, the combination circuit design remains challenging. A. Filter Topology
In this letter, a novel approach is introduced for designing Fig. 1 shows the topology of the proposed microstrip dual-
dual-band BPFs. Two sets of resonators operating at different band BPF. This is a two-order filter with a symmetrical struc-
frequencies are utilized to generate two passbands. A special ture. It consists of two sets of half-wavelength resonators. With
feed scheme is employed. One set of resonators not only longer length, resonators 1 and 4 are used to generate the lower
passband with center frequency . Open-circuited couple lines
are employed to realize electrical coupling. Embedded between
Manuscript received December 15, 2008; revised January 08, 2009. First pub-
lished May 26, 2009; current version published June 05, 2009. This work was
resonators 1 and 4, resonators 2 and 3 are utilized for yielding
supported by the Research Grants Council of Hong Kong Special Administra- the upper passband at . An interdigital capacitor is utilized to
tive Region, China, under Grant CityU122407. couple RF signals. Two 50 lines are connected to resonators
The authors are with the State Key Laboratory of Millimeter Waves, Depart- 1 and 4, acting as input and output ports. The feed and coupling
ment of Electronic Engineering, City University of Hong Kong, Hong Kong,
China (e-mail: zhangxiuyin@hotmail.com). schemes vary with frequency and thus should be addressed at
Digital Object Identifier 10.1109/LMWC.2009.2020009 respective resonant frequencies.
1531-1309/$25.00 © 2009 IEEE
ZHANG et al.: DUAL-BAND BPF DESIGN 351

At , resonators 1 and 4 do not resonate and thus are non-

resonating nodes [10]. In this case, they function as feed cir-
cuits for resonators 2 and 3, like that in [11]. Since the non-res-
onating nodes 1 and 4 are part of source/load and they are cou-
pled, source-load coupling is inherently realized. The source-
load coupling could help create a pair of transmission zeros on
both sides of upper passband [12], [13].

III. FILTER IMPLEMENTATION

The design procedures of this kind of filters are as follows.
We can determine the initial values of the parameters from the
design specifications. Then, fine tuning is performed to achieve
the desired response. The first step is to obtain desired passband
Fig. 2. Feed and coupling schemes. (a) At lower resonant frequency. (b) At
upper resonant frequency. frequencies. The lower passband frequency is affected by the
length of resonators 1 and 4 as well as the loading effect of the
other two resonators. Therefore, the total length of resonators 1
and 4 is chosen to be slightly shorter than half guided wave-
length at . The upper passband frequency is mainly deter-
mined by the length of resonators 2 and 3. Thus, the total length
of these two resonators is around half guided wavelength at .
The second step is to determine the dimensions of tapping and
coupling structures to satisfy the requirement of external quality
factors and coupling coefficients . The desired values
Fig. 3. Interdigital capacitor and its equivalent circuit. of and at the two passbands can be obtained using the syn-
thesis methods presented in [11]–[13]. At lower passband, the
and between resonators 1 and 4 are determined by the tap
B. Feed and Coupling Schemes for Lower Passband position , gap and length , respectively. At upper pass-
band, the is determined by the coupling between resonators
Fig. 2(a) illustrates the feed and coupling schemes at lower
1 and 2, i.e., the gap and line width and . The be-
resonant frequency , where the black and hollow nodes repre-
tween resonators 2 and 3, , depends on the dimensions of the
sent resonators and source/load, respectively. At , resonators
interdigital capacitor. Thus, the finger number and dimensions
1 and 4 resonate and thus the coupling between them is strong.
of the interdigital capacitor can be determined based on the re-
In contrast, the coupling between resonators 2 and 3 is relatively
quired . The source-load coupling or the coupling between
weak because of their non-resonance at . In this case, res-
resonators 1 and 4, , relies on the gap and length . It is
onators 2 and 3 function as loading elements of resonators 1
worth mentioning that the coupling between resonators 1 and 4
and 4, which shifts the resonant frequency downward and re-
should meet the demand of both passbands and this can be real-
duces circuit size. RF signals at are mainly coupled between
ized by tuning two variables, i.e., the gap and length . So
resonators 1 and 4. The coupling coefficient is determined by
far, the initial dimensions of the filter can be obtained and thus
the length and gap , as shown in Fig. 1.
fine tuning can be carried out to meet the specifications.
As for feed circuit at , 50 microstrip lines are directly
For demonstration purpose, a dual-band BPF is implemented
tapped at the resonators 1 and 4, like that of open-loop filters [4],
operating at 1.8 and 2.4 GHz for DCS and WLAN applications.
[9]. The external quality factor is determined by the tap position
The experimental filter is fabricated on the substrate with a rela-
or the length [9].
tive dielectric constant of 2.94 and thickness of 0.762 mm. Fol-
lowing the preceding design process, the dimensions are ob-
C. Feed and Coupling Schemes for Upper Passband
tained as follows: , ,
The feed and coupling schemes at upper resonant frequency , , , ,
is illustrated in Fig. 2(b). At , resonators 2 and 3 resonate , , , ,
and the coupling is much stronger than that between resonators , and . The dimensions of the 14
1 and 4. In order to provide sufficient degrees of freedom for finger interdigital capacitor are: , ,
achieving both wide and narrow upper passband, interdigital ca- . Due to the unavoidable presence of fabrication
pacitor is employed to control the coupling strength within a tolerance, it is necessary to analyze the sensitivity. In this filter,
wide range. Fig. 3 shows the interdigital capacitor and its equiv- the performance of both passbands is affected by the coupling
alent circuit. By changing the finger number, the length , gap between resonators 1 and 4. Furthermore, the coupling gap
and width , the coupling coefficient can be conveniently is narrow and thus it is more sensitive to fabrication errors than
tuned to the desired value within a wide range. It is noted that other parameters. Therefore, the gap is chosen to perform
coupled lines can be used instead of interdigitial capacitors in the sensitivity analysis. It undergoes random errors of 16.7%,
case of a narrow upper passband. or 0.05 mm, which is usually the maximum fabrication error.
352 IEEE MICROWAVE AND WIRELESS COMPONENTS LETTERS, VOL. 19, NO. 6, JUNE 2009

of 2.4 GHz WLAN system (2.4–2.485 GHz). The minimum in-

sertion loss is measured to be 1.1 dB. The return loss within
the passband is greater than 16 dB. Two transmission zeros are
generated at 2.32 and 2.70 GHz. They are due to coupling and
routing scheme with source-load coupling. The mechanism is
detailed in [12] and [13].

IV. CONCLUSION
This letter has presented a planar dual-band BPF based on
a novel feed scheme. This scheme makes full use of one set
of resonators. They perform dual functions, acting as both res-
onators and feed circuits at different frequencies. Meanwhile,
source-load coupling is inherently realized and help create a
pair of transmission zeros. Sufficient degrees of freedom are
provided to achieve various specifications including center fre-
quency and bandwidth. The design methodology and filter sen-
Fig. 4. Filter responses when the gap g undergoes random errors of 16.7%.
sitivity have been presented and a demonstration filter has been
implemented for DCS and WLAN applications. Four transmis-
sion zeros are realized, resulting in high selectivity. The high
performance, planar structure and compact size make it attrac-
tive for wireless communications.

REFERENCES
[1] J.-T. Kuo, T.-H. Yeh, and C.-C. Yeh, “Design of microstrip bandpass
filter with a dual-passband response,” IEEE Trans. Microw. Theory
Tech., vol. 53, no. 4, pp. 1331–1337, Apr. 2005.
[2] S. Sun and L. Zhu, “Compact dual-band microstrip bandpass filter
without external feeds,” IEEE Microw. Wireless Compon. Lett., vol.
15, no. 10, pp. 644–646, Oct. 2005.
[3] Y. P. Zhang and M. Sun, “Dual-band microstrip bandpass filter using
stepped-impedance resonators with new coupling schemes,” IEEE
Trans. Microw. Theory Tech., vol. 54, no. 10, pp. 3779–3885, Oct.
2006.
[4] X. Y. Zhang and Q. Xue, “Novel centrally loaded resonators and their
applications to bandpass filters,” IEEE Trans. Microw. Theory Tech.,
vol. 56, no. 4, pp. 913–921, Apr. 2008.
[5] H. Miyake, S. Kitazawa, T. Ishizaki, T. Yamanda, and Y. Nagatomi,
Fig. 5. Simulated and measured results. “A miniaturized monolithic dual-band filter using ceramic lamination
technique for dual-mode portable telephones,” in IEEE MTT-S Int.
Dig., Denver, CO, Jun. 1997, pp. 789–792.
[6] C.-Y. Chen and C.-Y. Hsu, “A simple and effective method for
Fig. 4 illustrates the simulated responses. As can be observed, microstrip dual-band filters design,” IEEE Microw. Wireless Compon.
Lett., vol. 16, no. 3, pp. 246–248, May 2006.
the performance in all situations is acceptable, indicating that [7] J.-X. Chen, T. Y. Yum, J.-L. Li, and Q. Xue, “Dual-mode dual-band
this is a low-sensitivity design. bandpass filter using stacked-loop structure,” IEEE Microw. Wireless
The simulation and measurement are accomplished using Compon. Lett., vol. 16, no. 9, pp. 502–504, Sep. 2006.
[8] X. Y. Zhang and Q. Xue, “Novel dual-mode dual-band bandpass filters
IE3D and 8753ES network analyzer, respectively. Fig. 5 de- using coplanar-waveguide-fed ring resonators,” IEEE Trans. Microw.
picts the simulated and measured results, which show good Theory Tech., vol. 55, no. 10, pp. 2183–2190, Oct. 2007.
agreement. Centered at 1.84 GHz, one passband has the 1 [9] J. S. Hong and M. J. Lancaster, Microwave Filter for RF/Microwave
Application. New York: Wiley, 2001.
dB bandwidth of 86 MHz or 4.9%, which covers the DCS [10] S. Amari, U. Rosenberg, and J. Bornemann, “Singlets, cascaded sin-
downlink frequency from 1.805 to 1.880 GHz. The insertion glets, and the nonresonating node model for advanced modular design
loss, including the loss from SMA connectors, is measured of elliptic filters,” IEEE Microw. Wireless Compon. Lett., vol. 14, no.
5, pp. 237–239, May 2004.
to be 0.8 dB. Two transmission poles within the passband [11] C. S. Cho, J. W. Lee, and J. Kim, “Dual- and triple-mode branch-line
are observed and the return loss is greater than 17 dB. Two ring resonators and harmonic suppressed half-ring resonators,” IEEE
transmission zeros are realized close to the passband edges at Trans. Microw. Theory Tech., vol. 54, no. 11, pp. 3968–3974, Nov.
2006.
1.64 and 2.01 GHz. These two transmission zeros are due to [12] J. R. Montejo-Garai, “Synthesis of N-even order symmetric filters with
the tap connection of the ports which leads to quarter-wave N transmission zeros by means of source-load cross coupling,” Elec-
resonance [4]. tron. Lett., vol. 36, no. 3, pp. 232–233, Feb. 2000.
[13] S. Amari, “Direct synthsis of folded symmetric resonator filters with
The other passband is located at 2.45 GHz. The 1 dB band- source-loading coupling,” IEEE Microw. Wireless Compon. Lett., vol.
width is 95 MHz or 3.8%, which covers the frequency range 11, no. 6, pp. 264–266, Jun. 2001.