0% found this document useful (0 votes)

33 views37 pages

FN 6453

High-Efficiency, Quad-Output, Main Power Supply Controllers for Notebook Computers

Uploaded by

mileny

We take content rights seriously. If you suspect this is your content, claim it here.

Available Formats

Download as PDF, TXT or read online on Scribd

0% found this document useful (0 votes)

33 views37 pages

FN 6453

High-Efficiency, Quad-Output, Main Power Supply Controllers for Notebook Computers

Uploaded by

mileny

We take content rights seriously. If you suspect this is your content, claim it here.

Available Formats

Download as PDF, TXT or read online on Scribd

You are on page 1/ 37

®

ISL6236A

Data Sheet March 18, 2008 FN6453.3

High-Efficiency, Quad-Output, Main Power Features

Supply Controllers for Notebook • Wide Input Voltage Range 4.5V to 25V
Computers
• Dual Fixed 1.05V/3.3V and 1.5V/5.0V Outputs or
The ISL6236A dual step-down, switch-mode power-supply Adjustable 0.7V to 5.5V (SMPS1) and 0V to 2.5V
(SMPS) controller generates logic-supply voltages in (SMPS2), ±1.5% Accuracy
battery-powered systems. The ISL6236A includes two
• Secondary Feedback Input (Maintains Charge Pump
pulse-width modulation (PWM) controllers, 5V/3.3V and
Voltage)
1.5V/1.05V. The output of SMPS1 can also be adjusted from
0.7V to 5.5V. The SMPS2 output can be adjusted from 0V to • 1.7ms Digital Soft-Start and Independent Shutdown
2.5V by setting REFIN2 voltage. An optional external charge • Fixed 3.3V/5.0V, or Adjustable Output 0.7V to 4.5V,
pump can be monitored through SECFB. This device features ±1.5% (LDO): 200mA
a linear regulator providing 3.3V/5V, or adjustable from 0.7V to
4.5V output via LDOREFIN. The linear regulator provides up • 3.3V Reference Voltage ±2.0%: 5mA
to 100mA output current with automatic linear-regulator • 2.0V Reference Voltage ±1.0%: 50µA
bootstrapping to the BYP input. When in switchover, the LDO
• Constant ON-TIME Control with 100ns Load-Step
output can source up to 200mA. The ISL6236A includes
Response
on-board power-up sequencing, the power-good (POK)
outputs, digital soft-start, and internal soft-stop output • Frequency Selectable
discharge that prevents negative voltages on shutdown. • rDS(ON) Current Sensing
A constant on-time PWM control scheme operates without • Programmable Current Limit with Foldback Capability
sense resistors and provides 100ns response to load
• Selectable PWM, Skip or Ultrasonic Mode
transients while maintaining a relatively constant switching
frequency. The unique ultrasonic pulse-skipping mode • BOOT Voltage Monitor with Automatic Refresh
maintains the switching frequency above 25kHz, which
• Independent POK1 and POK2 Comparators
eliminates noise in audio applications. Other features include
pulse skipping, which maximizes efficiency in light-load • Soft-Start with Pre-Biased Output and Soft-Stop
applications, and fixed-frequency PWM mode, which reduces • Independent ENABLE
RF interference in sensitive applications.
• High Efficiency - up to 97%
Ordering Information
• Very High Light Load Efficiency (Skip Mode)
PART TEMP
NUMBER PART RANGE PACKAGE PKG. • 5mW Quiescent Power Dissipation
(Note) MARKING (°C) (Pb-free) DWG. #
• Thermal Shutdown
ISL6236AIRZ ISL6236 AIRZ -40 to +100 32 Ld 5x5 QFN L32.5x5B
• Extremely Low Component Count
ISL6236AIRZ-T* ISL6236 AIRZ -40 to +100 32 Ld 5x5 QFN L32.5x5B
(Tape and Reel) • Pb-Free (RoHS Compliant)

*Please refer to TB347 for details on reel specifications. Applications

NOTE: These Intersil Pb-free plastic packaged products employ special
Pb-free material sets; molding compounds/die attach materials and 100% • Notebook and Sub-Notebook Computers
matte tin plate PLUS ANNEAL - e3 termination finish, which is RoHS
compliant and compatible with both SnPb and Pb-free soldering operations. • PDAs and Mobile Communication Devices
Intersil Pb-free products are MSL classified at Pb-free peak reflow
• 3-Cell and 4-Cell Li+ Battery-Powered Devices
temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC
J STD-020. • DDR1, DDR2, and DDR3 Power Supplies
• Graphic Cards
• Game Consoles
• Telecommunications

1 CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright Intersil Americas Inc. 2007, 2008. All Rights Reserved
All other trademarks mentioned are the property of their respective owners.
ISL6236A

Pinout
ISL6236A
(32 LD 5x5 QFN)
TOP VIEW

PHASE2
UGATE2
REFIN2

POK2
OUT2
ILIM2

SKIP

EN2
32 31 30 29 28 27 26 25

REF 1 24 BOOT2

TON 2 23 LGATE2

VCC 3 22 PGND

EN LDO 4 21 GND

VREF3 5 20 SECFB

VIN 6 19 PVCC

LDO 7 18 LGATE1

LDOREFIN 8 17 BOOT1

9 10 11 12 13 14 15 16
OUT1

ILIM1
BYP

FB1

POK1

EN1

UGATE1

PHASE1

2 FN6453.3
March 18, 2008
ISL6236A

Absolute Voltage Ratings Thermal Information

VIN, EN LDO to GND. . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +27V Thermal Resistance (Typical) θJA (°C/W) θJC (°CW)
BOOT to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +33V 32 Ld QFN (Notes 1, 2) . . . . . . . . . . . . 32 3.0
BOOT to PHASE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +6V Operating Temperature Range . . . . . . . . . . . . . . . .-40°C to +100°C
VCC, EN, SKIP, TON, PVCC, POK to GND . . . . . . . . . -0.3V to +6V Junction Temperature . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +150°C
LDO, FB1, REFIN2, LDOREFIN to GND . . . -0.3V to (VCC + 0.3V) Storage Temperature Range . . . . . . . . . . . . . . . . . .-65°C to +150°C
OUT, SECFB, VREF3, REF to GND . . . . . . . . -0.3V to (VCC + 0.3V Pb-free reflow profile . . . . . . . . . . . . . . . . . . . . . . . . . .see link below
UGATE to PHASE . . . . . . . . . . . . . . . . . . . . -0.3V to (PVCC + 0.3V) http://www.intersil.com/pbfree/Pb-FreeReflow.asp
ILIM to GND . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to (VCC + 0.3V)
LGATE, BYP to GND . . . . . . . . . . . . . . . . . . -0.3V to (PVCC + 0.3V)
PGND to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +0.3V
LDO, REF, VREF3 Short Circuit to GND . . . . . . . . . . . . Continuous
VCC Short Circuit to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1s
LDO Current (Internal Regulator) Continuous . . . . . . . . . . . . 100mA
LDO Current (Switched Over to OUT1) Continuous . . . . . . +200mA

CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product reliability and
result in failures not covered by warranty.

NOTES:
1. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See
Tech Brief TB379.
2. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside.
3. Limits established by characterization and are not production tested.
4. Parts are 100% tested at +25°C. Temperature limits established by characterization and are not production tested.

Electrical Specifications No load on LDO, OUT1, OUT2, VREF3, and REF, VIN = 12V, EN2 = EN1 = VCC, VBYP = 5V, PVCC = 5V,
VEN_LDO = 5V, TA = -40°C to +100°C, unless otherwise noted. Typical values are at TA = +25°C.

MIN MAX
PARAMETER CONDITIONS (Note 4) TYP (Note 4) UNITS

MAIN SMPS CONTROLLERS

VIN Input Voltage Range LDO in regulation 5.5 25 V

VIN = LDO, VOUT1 <4.43V 4.5 5.5 V

3.3V Output Voltage in Fixed Mode VIN = 4.5V to 25V, REFIN2 > (VCC - 1V), SKIP = 5V 3.285 3.330 3.375 V
1.05V Output Voltage in Fixed Mode VIN = 4.5V to 25V, 3.0 < REFIN2 < (VCC - 1.1V), 1.038 1.05 1.062 V
SKIP = 5V

1.5V Output Voltage in Fixed Mode VIN = 4.5V to 25V, FB1 = VCC, SKIP = 5V 1.482 1.500 1.518 V
5V Output Voltage in Fixed Mode VIN = 5.5V to 25V, FB1 = GND, SKIP = 5V 4.975 5.050 5.125 V

FB1 in Output Adjustable Mode VIN = 4.5V to 25V 0.693 0.700 0.707 V

REFIN2 in Output Adjustable Mode VIN = 4.5V to 25V 0.7 2.50 V

SECFB Voltage VIN = 4.5V to 25V 1.920 2.00 2.080 V

SMPS1 Output Voltage Adjust Range SMPS1 0.70 5.50 V

SMPS2 Output Voltage Adjust Range SMPS2 0.50 2.50 V

SMPS2 Output Voltage Accuracy REFIN2 = 0.7V to 2.5V, SKIP = VCC -1.0 1.0 %
(Referred for REFIN2)

DC Load Regulation Either SMPS, SKIP = VCC, 0A to 5A -0.1 %

Either SMPS, SKIP = REF, 0A to 5A -1.7 %

Either SMPS, SKIP = GND, 0A to 5A -1.5 %

Line Regulation Either SMPS, 6V < VIN < 24V 0.005 %/V
Current-Limit Current Source Temperature = +25°C 4.75 5 5.25 µA

ILIM Adjustment Range 0.2 2 V

3 FN6453.3
March 18, 2008
ISL6236A

MIN MAX
PARAMETER CONDITIONS (Note 4) TYP (Note 4) UNITS
Current-Limit Threshold (Positive, Default) ILIM = VCC, GND - PHASE 93 100 107 mV
(No temperature compensation)

Current-Limit Threshold GND - PHASE VILIM = 0.5V 40 50 60 mV

(Positive, Adjustable)
VILIM = 1V 93 100 107 mV

VILIM = 2V 185 200 215 mV

Zero-Current Threshold SKIP = GND, REF, or OPEN, GND - PHASE 3 mV

Current-Limit Threshold (Negative, Default) SKIP = VCC, GND - PHASE -120 mV

Soft-Start Ramp Time Zero to full limit 1.7 ms

Operating Frequency (VtON = GND), SKIP = VCC SMPS 1 400 kHz

SMPS 2 500 kHz

(VtON = REF or OPEN), SMPS 1 400 kHz

SKIP = VCC
SMPS 2 300 kHz

(VtON = VCC), SKIP = VCC SMPS 1 200 kHz

SMPS 2 300 kHz

On-Time Pulse Width VtON = GND (400kHz/500kHz) VOUT1 = 5.00V 0.895 1.052 1.209 µs

VOUT2 = 3.33V 0.475 0.555 0.635 µs

VtON = REF or OPEN VOUT1 = 5.05V 0.895 1.052 1.209 µs
(400kHz/300kHz)
VOUT2 = 3.33V 0.833 0.925 1.017 µs

VtON = VCC (200kHz/300kHz) VOUT1 = 5.05V 1.895 2.105 2.315 µs

VOUT2 = 3.33V 0.833 0.925 1.017 µs

Minimum Off-Time TA = -40°C to +100°C 200 300 425 ns

TA = -40°C to +85°C 200 300 410 ns

Maximum Duty Cycle VtON = GND VOUT1 = 5.05V 88 %

VOUT2 = 3.33V 85 %

VtON = REF or OPEN VOUT1 = 5.05V 88 %

VOUT2 = 3.33V 91 %

VtON = VCC VOUT1 = 5.05V 94 %

VOUT2 = 3.33V 91 %

Ultrasonic SKIP Operating Frequency SKIP = REF or OPEN 25 37 kHz

INTERNAL REGULATOR AND REFERENCE

LDO Output Voltage BYP = GND, 5.5V < VIN < 25V, LDOREFIN < 0.3V, 4.925 5.000 5.075 V
0 < ILDO < 100mA

BYP = GND, 4.5V < VIN < 25V, LDOREFIN > (VCC - 1V), 3.250 3.300 3.350 V
0 < ILDO < 100mA

BYP = GND, 4.5V < VIN < 25V, LDOREFIN = 2V, 3.94 4.00 4.06 V
0 < ILDO < 100mA
LDO Output Accuracy in Adjustable Mode VIN = 4.5V to 25V, VLDOREFIN = 0.35V to 0.5V ±2.5 %

VIN = 4.5V to 25V, VLDOREFIN = 0.5V to 2V ±1.5 %

VIN = 5.5V to 25V, VLDOREFIN = 2V to 2.25V ±1.5 %

LDOREFIN Input Range VLDO = 2 x VLDOREFIN 0.35 2.25 V

4 FN6453.3
March 18, 2008
ISL6236A

MIN MAX
PARAMETER CONDITIONS (Note 4) TYP (Note 4) UNITS
LDO Output Current BYP = GND, VIN = 4.5V to 25V (Note 3) 100 mA

LDO Output Current During Switchover to 5V BYP = 5V, VIN = 5.5V to 25V, LDOREFIN < 0.3V 200 mA

LDO Output Current During Switchover BYP = 3.3V, VIN = 4.5V to 25V, LDOREFIN > (VCC - 1V) 100 mA
to 3.3V

LDO Short-Circuit Current LDO = GND, BYP = GND 200 400 mA

Undervoltage-Lockout Fault Threshold Rising edge of PVCC 4.35 4.5 V

Falling edge of PVCC 3.9 4.05 V

LDO 5V Bootstrap Switch Threshold to BYP Rising edge at BYP regulation point 4.53 4.68 4.83 V
LDOREFIN = GND

LDO 3.3V Bootstrap Switch Threshold to Rising edge at BYP regulation point 3.0 3.1 3.2 V
BYP LDOREFIN = VCC

LDO 5V Bootstrap Switch Equivalent LDO to BYP, BYP = 5V, LDOREFIN > (VCC - 1V) 0.7 1.5 Ω
Resistance (Note 3)

LDO 3.3V Bootstrap Switch Equivalent LDO to BYP, BYP = 3.3V, LDOREFIN < 0.3V (Note 3) 1.5 3.0 Ω
Resistance

VREF3 Output Voltage No external load, VCC > 4.5V 3.235 3.300 3.365 V

No external load, VCC < 4.0V 3.220 3.300 3.380 V

VREF3 Load Regulation 0 < ILOAD < 5mA 10 mV

VREF3 Current Limit VREF3 = GND 10 17 mA

REF Output Voltage No external load 1.980 2.000 2.020 V

REF Load Regulation 0 < ILOAD < 50µA 10 mV

REF Sink Current REF in regulation 10 µA

VIN Operating Supply Current Both SMPSs on, FB1 = SKIP = GND, REFIN2 = VCC 25 50 µA
VOUT1 = BYP = 5.3V, VOUT2 = 3.5V

VIN Standby Supply Current VIN = 5.5V to 25V, both SMPSs off, EN LDO = VCC 180 250 µA
VIN Shutdown Supply Current VIN = 4.5V to 25V, EN1 = EN2 = EN LDO = 0V 20 30 µA

Quiescent Power Consumption Both SMPSs on, FB1 = SKIP = GND, REFIN2 = VCC, 5 7 mW
VOUT1 = BYP = 5.3V, VOUT2 = 3.5V

FAULT DETECTION

Overvoltage Trip Threshold FB1 with respect to nominal regulation point +8 +11 +14 %

REFIN2 with respect to nominal regulation point +12 +16 +20 %

Overvoltage Fault Propagation Delay FB1 or REFIN2 delay with 50mV overdrive 10 µs

POK_ Threshold FB1 or REFIN2 with respect to nominal output, falling -12 -9 -6 %
edge, typical hysteresis = 1%

POK_ Propagation Delay Falling edge, 50mV overdrive 10 µs

POK_ Output Low Voltage ISINK = 4mA 0.2 V

POK_ Leakage Current High state, forced to 5.5V 1 µA

Thermal-Shutdown Threshold +150 °C

Out-Of-Bound Threshold FB1 or REFIN2 with respect to nominal output voltage 5 %

Output Undervoltage Shutdown Threshold FB1 or REFIN2 with respect to nominal output voltage 65 70 75 %

Output Undervoltage Shutdown Blanking From EN signal 10 20 30 ms

Time

5 FN6453.3
March 18, 2008
ISL6236A

MIN MAX
PARAMETER CONDITIONS (Note 4) TYP (Note 4) UNITS
INPUTS AND OUTPUTS

FB1 Input Voltage Low level 0.3 V

High level VCC - 1.0 V

REFIN2 Input Voltage OUT2 Dynamic Range, VOUT2 = VREFIN2 0.5 2.50 V

Fixed OUT2 = 1.05V 3.0 VCC - 1.1 V

Fixed OUT2 = 3.3V VCC - 1.0 V

LDOREFIN Input Voltage Fixed LDO = 5V 0.30 V

LDO Dynamic Range, VLDO = 2 x VLDOREFIN 0.35 2.25 V

Fixed LDO = 3.3V VCC - 1.0 V

SKIP Input Voltage Low level (SKIP) 0.8 V

Float level (ULTRASONIC SKIP) 1.7 2.3 V

High level (PWM) 2.4 V

TON Input Voltage Low level 0.8 V

Float level 1.7 2.3 V

High level 2.4 V

EN1, EN2 Input Voltage Clear fault level/SMPS off level 0.8 V
Delay start level 1.7 2.3 V

SMPS on level 2.4 V

EN LDO Input Voltage Rising edge 1.2 1.6 2.0 V

Falling edge 0.94 1.00 1.06 V

Input Leakage Current VtON = 0V or 5V -1 +1 µA

VEN = VEN LDO = 0V or 5V -0.1 +0.1 µA

VSKIP = 0V or 5V -1 +1 µA

VFB1 = VSECFB = 0V or 5V -0.2 +0.2 µA

VREFIN = 0V or 2.5V -0.2 +0.2 µA

VLDOREFIN = 0V or 2.75V -0.2 +0.2 µA

INTERNAL BOOT DIODE

VD Forward Voltage PVCC - VBOOT, IF = 10mA 0.65 0.8 V

IBOOT LEAKAGE Leakage Current VBOOT = 30V, PHASE = 25V, PVCC = 5V 500 nA

MOSFET DRIVERS

UGATE Gate-Driver Sink/Source Current UGATE1, UGATE2 forced to 2V 2 A

LGATE Gate-Driver Source Current LGATE1 (source), LGATE2 (source), forced to 2V 1.7 A

LGATE Gate-Driver Sink Current LGATE1 (sink), LGATE2 (sink), forced to 2V 3.3 A

UGATE Gate-Driver ON-resistance BOOT_ - PHASE_ forced to 5V (Note 3) 1.5 4.0 Ω

LGATE Gate-Driver ON-resistance LGATE, high state (pull-up) (Note 3) 2.2 5.0 Ω

LGATE, low state (pull-down) (Note 3) 0.6 1.5 Ω

Dead Time LGATE Rising 15 20 35 ns

UGATE Rising 20 30 50 ns

OUT1, OUT2 Discharge ON-resistance 25 40 Ω

6 FN6453.3
March 18, 2008
ISL6236A

Pin Descriptions
PIN NAME FUNCTION

1 REF 2V Reference Output. Bypass to GND with a 0.1µF (min) capacitor. REF can source up to 50µA for external loads.
Loading REF degrades FB and output accuracy according to the REF load-regulation error.

2 TON Frequency Select Input. Connect to GND for 400kHz/500kHz operation. Connect to REF (or leave OPEN) for
400kHz/300kHz operation. Connect to VCC for 200kHz/300kHz operation (5V/3.3V SMPS switching frequencies,
respectively).

3 VCC Analog Supply Voltage Input for PWM Core. Bypass to GND with a 1µF ceramic capacitor.

4 EN LDO LDO Enable Input. The LDO is enabled if EN LDO is within logic high level and disabled if EN LDO is less than the logic
low level.

5 VREF3 3.3V Reference Output. VREF3 can source up to 5mA for external loads. Bypass to GND with a 0.01µF capacitor if
loaded. Leave open if there is no load.

6 VIN Power-Supply Input. VIN is used for the constant-on-time PWM on-time one-shot circuits. VIN is also used to power the
linear regulators. The linear regulators are powered by SMPS1 if OUT1 is set greater than 4.78V and BYP is tied to
OUT1. Connect VIN to the battery input and bypass with a 1µF capacitor.

7 LDO Linear-Regulator Output. LDO can provide a total of 100mA external loads. The LDO regulates at 5V If LDOREFIN is
connected to GND. When the LDO is set at 5V and BYP is within 5V switchover threshold, the internal regulator shuts
down and the LDO output pin connects to BYP through a 0.7Ω switch. The LDO regulates at 3.3V if LDOREFIN is
connected to VCC. When the LDO is set at 3.3V and BYP is within the 3.3V switchover threshold, the internal regulator
shuts down and the LDO output pin connects to BYP through a 1.5Ω switch. Bypass LDO output with a minimum of
4.7µF ceramic.

8 LDOREFIN LDO Reference Input. Connect LDOREFIN to GND for fixed 5V operation. Connect LDOREFIN to VCC for fixed 3.3V
operation. LDOREFIN can be used to program LDO output voltage from 0.7V to 4.5V. LDO output is two times the
voltage of LDOREFIN. There is no switchover in adjustable mode.

9 BYP BYP is the switchover source voltage for the LDO when LDOREFIN is connected to GND or VCC. Connect BYP to 5V
if LDOREFIN is tied to GND. Connect BYP to 3.3V if LDOREFIN is tied to VCC.

10 OUT1 SMPS1 Output Voltage-Sense Input. Connect to the SMPS1 output. OUT1 is an input to the Constant on-time-PWM
on-time one-shot circuit. It also serves as the SMPS1 feedback input in fixed-voltage mode.

11 FB1 SMPS1 Feedback Input. Connect FB1 to GND for fixed 5V operation. Connect FB1 to VCC for fixed 1.5V operation.
Connect FB1 to a resistive voltage-divider from OUT1 to GND to adjust the output from 0.7V to 5.5V.

12 ILIM1 SMPS1 Current-Limit Adjustment. The GND-PHASE1 current-limit threshold is 1/10th the voltage seen at ILIM1 over a
0.2V to 2V range. There is an internal 5µA current source from VCC to ILIM1. Connect ILIM1 to REF for a fixed 200mV
threshold. The logic current limit threshold is default to 100mV value if ILIM1 is higher than VCC - 1V.

13 POK1 SMPS1 Power-Good Open-Drain Output. POK1 is low when the SMPS1 output voltage is more than 10% below the
normal regulation point or during soft-start. POK1 is high impedance when the output is in regulation and the soft-start
circuit has terminated. POK1 is low in shutdown.

14 EN1 SMPS1 Enable Input. The SMPS1 is enabled if EN1 is greater than the logic high level and disabled if EN1 is less than
the logic low level. If EN1 is connected to REF, the SMPS1 starts after the SMPS2 reaches regulation (delay start). Drive
EN1 below 0.8V to clear fault level and reset the fault latches.

15 UGATE1 High-Side MOSFET Floating Gate-Driver Output for SMPS1. UGATE1 swings between PHASE1 and BOOT1.

16 PHASE1 Inductor Connection for SMPS1. PHASE1 is the internal lower supply rail for the UGATE1 high-side gate driver.
PHASE1 is the current-sense input for the SMPS1.

17 BOOT1 Boost Flying Capacitor Connection for SMPS1. Connect to an external capacitor according to the “Typical Application
Circuits” on page 19 (Figures 66, 67 and 68). See “MOSFET Gate Drivers (UGATE, LGATE)” on page 29.

18 LGATE1 SMPS1 Synchronous-Rectifier Gate-Drive Output. LGATE1 swings between GND and PVCC.

19 PVCC PVCC is the supply voltage for the low-side MOSFET driver LGATE. Connect a 5V power source to the PVCC pin and
bypass to PGND with a 1µF MLCC ceramic capacitor. Refer to Figure 70 - A switch connects PVCC to VCC with 10Ω
when in normal operation and is disconnected when in shutdown mode. An external 10Ω resistor from PVCC to VCC
is prohibited as it will create a leakage path from VIN to GND in shutdown mode.

20 SECFB The SECFB is used to monitor the optional external 14V charge pump. Connect a resistive voltage-divider from 14V
charge pump output to GND to detect the output. If SECFB drops below the threshold voltage, LGATE1 turns on for
300ns. This will refresh the external charge pump driven by LGATE1 without over-discharging the output voltage.

7 FN6453.3
March 18, 2008
ISL6236A

Pin Descriptions (Continued)

PIN NAME FUNCTION

21 GND Analog Ground for both SMPS and LDO. Connect externally to the underside of the exposed pad.

22 PGND Power Ground for SMPS controller. Connect PGND externally to the underside of the exposed pad.

23 LGATE2 SMPS2 Synchronous-Rectifier Gate-Drive Output. LGATE2 swings between GND and PVCC.

24 BOOT2 Boost Flying Capacitor Connection for SMPS2. Connect to an external capacitor according to the “Typical Application
Circuits” on page 19 (Figures 66, 67 and 68). See “MOSFET Gate Drivers (UGATE, LGATE)” on page 29.

25 PHASE2 Inductor Connection for SMPS2. PHASE2 is the internal lower supply rail for the UGATE2 high-side gate driver.
PHASE2 is the current-sense input for the SMPS2.

26 UGATE2 High-Side MOSFET Floating Gate-Driver Output for SMPS2. UGATE1 swings between PHASE2 and BOOT2.

27 EN2 SMPS2 Enable Input. The SMPS2 is enabled if EN2 is greater than the logic high level and disabled if EN2 is less than
the logic low level. If EN2 is connected to REF, the SMPS2 starts after the SMPS1 reaches regulation (delay start). Drive
EN2 below 0.8V to clear fault level and reset the fault latches.

28 POK2 SMP2 Power-Good Open-Drain Output. POK2 is low when the SMPS2 output voltage is more than 10% below the
normal regulation point or during soft-start. POK2 is high impedance when the output is in regulation and the soft-start
circuit has terminated. POK2 is low in shutdown.

29 SKIP Low-Noise Mode Control. Connect SKIP to GND for normal Idle-Mode (pulse-skipping) operation or to VCC for PWM
mode (fixed frequency). Connect to REF or leave floating for ultrasonic skip mode operation.

30 OUT2 SMPS2 Output Voltage-Sense Input. Connect to the SMPS2 output. OUT2 is an input to the Constant on-time-PWM
on-time one-shot circuit. It also serves as the SMPS2 feedback input in fixed-voltage mode.

31 ILIM2 SMPS2 Current-Limit Adjustment. The GND-PHASE1 current-limit threshold is 1/10th the voltage seen at ILIM2 over a
0.2V to 2V range. There is an internal 5µA current source from VCC to ILIM2. Connect ILIM2 to REF for a fixed 200mV.
The logic current limit threshold is default to 100mV value if ILIM2 is higher than VCC - 1V.

32 REFIN2 Output voltage control for SMPS2. Connect REFIN2 to VCC for fixed 3.3V. Connect REFIN2 to VREF3 for fixed 1.05V.
REFIN2 can be used to program SMPS2 output voltage from 0.5V to 2.50V. SMPS2 output voltage is 0V if
REFIN2 < 0.5V.

Typical Performance Curves Circuit of Figures 66, 67 and 68, no load on LDO, OUT1, OUT2, VREF3, and REF, VIN = 12V,
EN2 = EN1 = VCC, VBYP = 5V, PVCC = 5V, VEN_LDO = 5V, TA = -40°C to +100°C, unless
otherwise noted. Typical values are at TA = +25°C.

7VIN SKIP MODE 12VIN ULTRA SKIP MODE

7VIN SKIP MODE 12VIN ULTRA SKIP MODE
7VIN PWM MODE 25VIN SKIP MODE
7VIN PWM MODE 25VIN SKIP MODE
7VIN ULTRA SKIP MODE 25VIN PWM MODE
7VIN ULTRA SKIP MODE 25VIN PWM MODE
12VIN SKIP MODE 25VIN ULTRA SKIP MODE
12VIN SKIP MODE 25VIN ULTRA SKIP MODE
12VIN PWM MODE
12VIN PWM MODE
100 100
90 90
80 80
EFFICIENCY (%)

70 70
EFFICIENCY (%)

60 60
50 50
40 40
30 30
20 20
10 10
0 0
0.001 0.010 0.100 1.000 10.000 0.001 0.010 0.100 1.000 10.000
OUTPUT LOAD (A) OUTPUT LOAD (A)
FIGURE 1. VOUT2 = 1.05V EFFICIENCY vs LOAD (300kHz) FIGURE 2. VOUT1 = 1.5V EFFICIENCY vs LOAD (200kHz)

8 FN6453.3
March 18, 2008
ISL6236A

7VIN SKIP MODE 12VIN ULTRA SKIP MODE 7VIN SKIP MODE 12VIN ULTRA SKIP MODE
7VIN PWM MODE 25VIN SKIP MODE 7VIN PWM MODE 25VIN SKIP MODE
7VIN ULTRA SKIP MODE 25VIN PWM MODE 7VIN ULTRA SKIP MODE 25VIN PWM MODE
12VIN SKIP MODE 25VIN ULTRA SKIP MODE 12VIN SKIP MODE 25VIN ULTRA SKIP MODE
12VIN PWM MODE 12VIN PWM MODE
100 100
90 90
80 80
EFFICIENCY (%)

EFFICIENCY (%)
70 70
60 60
50 50
40 40
30 30
20 20
10 10
0 0
0.001 0.010 0.100 1.000 10.000 0.001 0.010 0.100 1.000 10.000
OUTPUT LOAD (A) OUTPUT LOAD (A)
FIGURE 3. VOUT2 = 3.3V EFFICIENCY vs LOAD (500kHz) FIGURE 4. VOUT1 = 5V EFFICIENCY vs LOAD (400kHz)

OUTPUT VOLTAGE (V)

1.530
1.064
1.062 1.525

1.060 1.520
1.058 1.515
1.056
1.510
1.054
1.052 1.505

1.050 1.500
0.001 0.010 0.100 1.000 10.000 0.001 0.010 0.100 1.000 10.000
OUTPUT LOAD (A) OUTPUT LOAD (A)
FIGURE 5. VOUT2 = 1.05V REGULATION vs LOAD (300kHz) FIGURE 6. VOUT1 = 1.5V REGULATION vs LOAD (200kHz)

3.37 5.14
OUTPUT VOLTAGE (V)

OUTPUT VOLTAGE (V)

5.12
3.36
5.10
3.35
5.08
3.34
5.06
3.33
5.04
3.32 5.02

3.31 5.00
0.001 0.010 0.100 1.000 10.000 0.001 0.010 0.100 1.000 10.000
OUTPUT LOAD (A) OUTPUT LOAD (A)

FIGURE 7. VOUT2 = 3.3V REGULATION vs LOAD (500kHz) FIGURE 8. VOUT1 = 5V REGULATION vs LOAD (400kHz)

9 FN6453.3
March 18, 2008
ISL6236A

POWER DISSIPATION (W)

2.0 2.0

1.5 1.5

1.0 1.0

0.5 0.5

0.0 0.0
0.001 0.010 0.100 1.000 10.000 0.001 0.010 0.100 1.000 10.000
OUTPUT LOAD (A) OUTPUT LOAD (A)
FIGURE 9. VOUT2 = 1.05V POWER DISSIPATION vs LOAD FIGURE 10. VOUT1 = 1.5V POWER DISSIPATION vs LOAD
(300kHz) (200kHz)

3.0 3.0
POWER DISSIPATION (W)
POWER DISSIPATION (W)

2.5 2.5

2.0 2.0

1.5 1.5

1.0 1.0

0.5 0.5

0.0 0.0
0.001 0.010 0.100 1.000 10.000 0.001 0.010 0.100 1.000 10.000
OUTPUT LOAD (A) OUTPUT LOAD (A)
FIGURE 11. VOUT2 = 3.3V POWER DISSIPATION vs LOAD FIGURE 12. VOUT1 = 5V POWER DISSIPATION vs LOAD
(500kHz) (400kHz)

1.064 1.068
NO LOAD PWM
1.062 1.066
1.064
OUTPUT VOLTAGE (V)

OUTPUT VOLTAGE (V)

1.060
1.062 NO LOAD PWM
1.058
1.060
1.056 1.058

1.054 MID LOAD PWM 1.056

1.054
1.052 MID LOAD PWM
MAX LOAD PWM 1.052
1.050
1.050 MAX
MAX LOAD
LOAD PWM
PWM
1.048 1.048
5 7 9 11 13 15 17 19 21 23 25 5 7 9 11 13 15 17 19 21 23 25
INPUT VOLTAGE (V) INPUT VOLTAGE (V)
FIGURE 13. VOUT2 = 1.05V OUTPUT VOLTAGE REGULATION FIGURE 14. VOUT2 = 1.05V OUTPUT VOLTAGE REGULATION
vs VIN (PWM MODE) vs VIN (SKIP MODE)

10 FN6453.3
March 18, 2008
ISL6236A

1.518 1.530

1.516 1.525
OUTPUT VOLTAGE (V)

OUTPUT VOLTAGE (V)

NO LOAD PWM
1.514 NO LOAD PWM
1.520
1.512 MID LOAD PWM
MID LOAD PWM 1.515
1.510
1.510
1.508 MAX LOAD PWM
MAX LOAD PWM

1.506 1.505

1.504 1.500
5 7 9 11 13 15 17 19 21 23 25 5 7 9 11 13 15 17 19 21 23 25
INPUT VOLTAGE (V) INPUT VOLTAGE (V)
FIGURE 15. VOUT1 = 1.5V OUTPUT VOLTAGE REGULATION FIGURE 16. VOUT1 = 1.5V OUTPUT VOLTAGE REGULATION
vs VIN (PWM MODE) vs VIN (SKIP MODE)

3.340 3.380

3.370
3.335 NO LOAD PWM
OUTPUT VOLTAGE (V)

OUTPUT VOLTAGE (V)

3.360
NO LOAD PWM
3.330
3.350

3.325 3.340 MAX LOAD PWM

3.330
3.320
3.320
MID LOAD PWM
3.315
3.310 MID LOAD PWM
MAX LOAD PWM
3.310 3.300
7 9 11 13 15 17 19 21 23 25 7 9 11 13 15 17 19 21 23 25
INPUT VOLTAGE (V) INPUT VOLTAGE (V)
FIGURE 17. VOUT2 = 3.3V OUTPUT VOLTAGE REGULATION FIGURE 18. VOUT2 = 3.3V OUTPUT VOLTAGE REGULATION
vs VIN (PWM MODE) vs VIN (SKIP MODE)

5.065
NO LOAD PWM 5.14

5.060
OUTPUT VOLTAGE (V)

OUTPUT VOLTAGE (V)

5.12

MAX LOAD PWM 5.10

5.055 NO LOAD PWM
MID LOAD PWM
5.08
5.050 MID LOAD PWM
5.06
5.045
5.04
MAX LOAD PWM
5.040 5.02
7 9 11 13 15 17 19 21 23 25 7 9 11 13 15 17 19 21 23 25
INPUT VOLTAGE (V) INPUT VOLTAGE (V)
FIGURE 19. VOUT1 = 5V OUTPUT VOLTAGE REGULATION vs FIGURE 20. VOUT1 = 5V OUTPUT VOLTAGE REGULATION vs
VIN (PWM MODE) VIN (SKIP MODE)

11 FN6453.3
March 18, 2008
ISL6236A

35
200

RIPPLE (mV)
PWM 30
150 25 PWM
ULTRA-SKIP
20
100
ULTRA-SKIP 15
10
50
5 SKIP
SKIP
0 0
0.001 0.010 0.100 1.000 10.000 0.001 0.010 0.100 1.000 10.000
OUTPUT LOAD (A) OUTPUT LOAD (A)
FIGURE 21. VOUT2 = 1.05V FREQUENCY vs LOAD FIGURE 22. VOUT2 = 1.05V RIPPLE vs LOAD

250 50

PWM 45
200 40
FREQUENCY (kHz)

PWM
35
RIPPLE (mV)

150 30
25
100 20 ULTRA-SKIP SKIP
ULTRA-SKIP
15
50 10
SKIP 5
0 0
0.001 0.010 0.100 1.000 10.000 0.001 0.010 0.100 1.000 10.000
OUTPUT LOAD (A) OUTPUT LOAD (A)
FIGURE 23. VOUT1 = 1.5V FREQUENCY vs LOAD FIGURE 24. VOUT1 = 1.5V RIPPLE vs LOAD

600 14
PWM PWM
500 12
FREQUENCY (kHz)

10
400
RIPPLE (mV)

8 ULTRA-SKIP
300 SKIP
6
200
ULTRA-SKIP 4

100 2
SKIP

0 0
0.001 0.010 0.100 1.000 10.000 0.001 0.010 0.100 1.000 10.000
OUTPUT LOAD (A) OUTPUT LOAD (A)
FIGURE 25. VOUT2 = 3.3V FREQUENCY vs LOAD FIGURE 26. VOUT2 = 3.3V RIPPLE vs LOAD

12 FN6453.3
March 18, 2008
ISL6236A

450 40
PWM
400 35
350 30
FREQUENCY (kHz)

PWM ULTRA-SKIP
300

RIPPLE (mV)
25
250
20
200 SKIP
15
150
100 ULTRA-SKIP 10

50 5
SKIP
0 0
0.001 0.010 0.100 1.000 10.000 0.001 0.010 0.100 1.000 10.000
OUTPUT LOAD (A) OUTPUT LOAD (A)

FIGURE 27. VOUT1 = 5V FREQUENCY vs LOAD FIGURE 28. VOUT1 = 5V RIPPLE vs LOAD

5.04 3.35
5.02
BYP = 0V 3.30
5.00
OUTPUT VOLTAGE (V)

OUTPUT VOLTAGE (V)

4.98 3.25
BYP = 0V
4.96
3.20
4.94
4.92 3.15

4.90 BYP = 3.3V

3.10
4.88 BYP = 5V
3.05
4.86
4.84 3.00
0 50 100 150 200 0 50 100 150 200
OUTPUT LOAD (mA) OUTPUT LOAD (mA)

FIGURE 29. LDO OUTPUT 5V vs LOAD FIGURE 30. LDO OUTPUT 3.3V vs LOAD

3.5 15.5

3.0
15.0
OUTPUT VOLTAGE (V)

OUTPUT VOLTAGE (V)

2.5
14.5
2.0
14.0
1.5
13.5
1.0

0.5 13.0

0 12.5
0 2 4 6 8 10 0 2 4 5 8 10
OUTPUT LOAD (mA) OUTPUT LOAD (mA)
FIGURE 31. VREF3 vs LOAD FIGURE 32. CHARGE PUMP vs LOAD (PWM)

13 FN6453.3
March 18, 2008
ISL6236A

50 1400

45 1200
INPUT CURRENT (mA)

INPUT CURRENT (µA)

1000
40
800
35
600
30
400
25
200

20 0.0
7 9 11 13 15 17 19 21 23 25 7 9 11 13 15 17 19 21 23 25
INPUT VOLTAGE (V) INPUT VOLTAGE (V)

FIGURE 33. PWM NO LOAD INPUT CURRENT vs VIN FIGURE 34. SKIP NO LOAD INPUT CURRENT vs VIN
(EN = EN2 = EN LDO = VCC) (EN1 = EN2 = EN LDO = VCC)

177.5 26.5
177.0 26.0
176.5 25.5
INPUT CURRENT (µA)

INPUT CURRENT (µA)

176.0 25.0
175.5 24.5
175.0 24.0
174.5 23.5
174.0 23.0
173.5 22.5
173.0 22.0
7 9 11 13 15 17 19 21 23 25 7 9 11 13 15 17 19 21 23 25
INPUT VOLTAGE (V) INPUT VOLTAGE (V)

FIGURE 35. STANDBY INPUT CURRENT vs VIN FIGURE 36. SHUTDOWN INPUT CURRENT vs VIN
(EN = EN2 = 0, EN LDO = VCC) (EN = EN2 = EN LDO = 0)

VREF3 500mV/DIV
EN1 5V/DIV
VOUT1 2V/DIV
LDO 1V/DIV
CP 5V/DIV

REF 1V/DIV IL1 2A/DIV

POK1 2V/DIV

FIGURE 37. REF, VREF3, LDO = 5V, CP, NO LOAD FIGURE 38. START-UP VOUT1 = 5V (NO LOAD, SKIP MODE)

14 FN6453.3
March 18, 2008
ISL6236A

EN1 5V/DIV EN1 5V/DIV

VOUT1 2V/DIV VOUT1 2V/DIV

IL1 2A/DIV IL1 5A/DIV

POK1 2V/DIV POK1 2V/DIV

FIGURE 39. START-UP VOUT1 = 5V (NO LOAD, PWM MODE) FIGURE 40. START-UP VOUT1 = 5V (FULL LOAD, PWM MODE)

EN2 5V/DIV EN2 5V/DIV

VOUT2 2V/DIV VOUT2 2V/DIV

IL2 2A/DIV
IL2 2A/DIV
POK2 2V/DIV POK2 2V/DIV

FIGURE 41. START-UP VOUT2 = 3.3V (NO LOAD, SKIP MODE) FIGURE 42. START-UP VOUT1 = 3.3V (NO LOAD, PWM MODE)

EN2 5V/DIV
EN2 5V/DIV

VOUT2 2V/DIV

VOUT2 2V/DIV VOUT1 2V/DIV

IL2 5A/DIV POK2 5V/DIV

POK2 2V/DIV POK1 5V/DIV

FIGURE 43. START-UP VOUT1 = 3.3V (FULL LOAD, FIGURE 44. DELAYED START-UP (VOUT1 = 5V, VOUT2 = 3.3V,
PWM MODE) EN1 = REF)

15 FN6453.3
March 18, 2008
ISL6236A

EN1 5V/DIV
EN1 5V/DIV VOUT2 2V/DIV

VOUT1 2V/DIV VOUT2 2V/DIV

POK1 5V/DIV VOUT1 2V/DIV

POK2 5V/DIV POK1 OR POK2 5V/DIV

FIGURE 45. DELAYED START-UP (VOUT1 = 5V, VOUT2 = 3.3V, FIGURE 46. SHUTDOWN (VOUT1 = 5V, VOUT2 = 3.3V,
EN2 = REF) EN2 = REF)

LGATE1 5V/DIV LGATE1 5V/DIV

VOUT1 RIPPLE 50mV/DIV

VOUT1 RIPPLE 100mV/DIV

IL1 5A/DIV IL1 5A/DIV

VOUT2 RIPPLE 50mV/DIV VOUT2 RIPPLE 50mV/DIV

FIGURE 47. LOAD TRANSIENT VOUT1 = 5V FIGURE 48. LOAD TRANSIENT VOUT1 = 5V (SKIP)

LGATE1 5V/DIV LGATE2 5V/DIV

VOUT1 RIPPLE 20mV/DIV

IL2 5A/DIV

VOUT1 RIPPLE 20mV/DIV

IL2 5A/DIV VOUT2 RIPPLE 50mV/DIV

VOUT2 RIPPLE 50mV/DIV

FIGURE 49. LOAD TRANSIENT VOUT1 = 3.3V (PWM) FIGURE 50. LOAD TRANSIENT VOUT1 = 3.3V (SKIP)

16 FN6453.3
March 18, 2008
ISL6236A

VOUT RIPPLE 20mV/DIV

LDO 1V/DIV
VOUT2 0.5V/DIV

REFIN2 0.5V/DIV LDOREFIN 0.5V/DIV

LDO RIPPLE 50mV/DIV VOUT2 RIPPLE 50mV/DIV

FIGURE 51. VOUT2 TRACKING TO REFIN2 FIGURE 52. LDO TRACKING TO LDOREFIN

EN1 5V/DIV EN1 5V/DIV

VOUT1 0.5V/DIV VOUT1 0.5V/DIV

IL1 2A/DIV
IL1 2A/DIV

POK1 2V/DIV POK1 2V/DIV

FIGURE 53. START-UP VOUT1 = 1.5V (NO LOAD, SKIP MODE) FIGURE 54. START-UP VOUT1 = 1.5V (NO LOAD, PWM MODE)

EN1 5V/DIV EN2 5V/DIV

VOUT1 0.5V/DIV

VOUT2 0.5V/DIV

IL1 5A/DIV IL2 2A/DIV

POK1 2V/DIV POK2 2V/DIV

FIGURE 55. START-UP VOUT1 = 1.5V (FULL LOAD, FIGURE 56. START-UP VOUT2 = 1.05V (NO LOAD,
PWM MODE) SKIP MODE)

17 FN6453.3
March 18, 2008
ISL6236A

EN2 5V/DIV
EN2 5V/DIV VOUT2 0.5V/DIV

VOUT2 0.5V/DIV

IL2 2A/DIV
IL2 2A/DIV

POK2 2V/DIV POK2 2V/DIV

FIGURE 57. START-UP VOUT1 = 1.05V (NO LOAD, FIGURE 58. START-UP VOUT1 = 1.05V (FULL LOAD,
PWM MODE) PWM MODE)

EN2 5V/DIV VOUT1 2V/DIV

EN1 500mV/DIV
VOUT2 0.5V/DIV

VOUT1 2V/DIV VOUT2 500mV/DIV

POK2 5V/DIV POK1 5V/DIV

POK1 5V/DIV POK2 5V/DIV

FIGURE 59. DELAYED START-UP (VOUT1 = 1.5V, FIGURE 60. DELAYED START-UP (VOUT1 = 1.5V,
VOUT2 = 1.05V, EN1 = REF) VOUT2 = 1.05V, EN2 = REF)

LGATE1 5V/DIV

EN1 5V/DIV

VOUT1 RIPPLE 50mV/DIV

VOUT2 2V/DIV

VOUT1 2V/DIV
IL1 5A/DIV

POK1 OR POK2 5V/DIV

VOUT2 RIPPLE 20mV/DIV

FIGURE 61. SHUTDOWN (VOUT1 = 1.5V, VOUT2 = 1.05V, FIGURE 62. LOAD TRANSIENT VOUT1 = 1.5V (PWM)
EN2 = REF)

18 FN6453.3
March 18, 2008
ISL6236A

LGATE1 5V/DIV LGATE2 5V/DIV

VOUT1 RIPPLE 20mV/DIV

VOUT1 RIPPLE 50mV/DIV

IL1 5A/DIV IL1 5A/DIV

VOUT2 RIPPLE 20mV/DIV VOUT2 RIPPLE 20mV/DIV

FIGURE 63. LOAD TRANSIENT VOUT1 = 1.5V (SKIP) FIGURE 64. LOAD TRANSIENT VOUT1 = 1.05V (PWM)

LGATE2 5V/DIV

VOUT1 RIPPLE 20mV/DIV

IL2 5A/DIV
VOUT2 RIPPLE 20mV/DIV

FIGURE 65. LOAD TRANSIENT VOUT1 = 1.05V (SKIP)

Typical Application Circuits a pin-selectable switching frequency, allowing operation for

200kHz/300kHz, 400kHz/300kHz, or 400kHz/500kHz on the
The typical application circuits (Figures 66, 67 and 68) generate
SMPSs.
the 5V/7A, 3.3V/11A, 1.25V/5A, dynamic voltage/10A, 1.5V/5A,
1.05V/5A and external 14V charge pump main supplies in a Light-load efficiency is enhanced by automatic Idle-Mode
notebook computer. The ISL6236A is also equipped with a operation, a variable-frequency pulse-skipping mode that
secondary feedback, SECFB, used to monitor the output of the reduces transition and gate-charge losses. Each step-down,
14V charge pump. In an event when the 14V drops below its power-switching circuit consists of 2 N-Channel MOSFETs, a
threshold voltage, SECFB comparator will turn on LGATE1 for rectifier, and an LC output filter. The output voltage is the
300ns. This will refresh an external 14V charge pump without average AC voltage at the switching node, which is
overcharging the output voltage. The input supply range is regulated by changing the duty cycle of the MOSFET
5.5V to 25V. switches. The gate-drive signal to the N-Channel high-side
MOSFET must exceed the battery voltage, and is provided
Detailed Description
by a flying-capacitor boost circuit that uses a 100nF
The ISL6236A dual-buck, BiCMOS, switch-mode power capacitor connected to BOOT.
supply controller generates logic supply voltages for
notebook computers. The ISL6236A is designed primarily for Both SMPS1 and SMPS2 PWM controllers consist of a triple
battery-powered applications where high efficiency and low mode feedback network and multiplexer, a multi-input PWM
quiescent supply current are critical. The ISL6236A provides comparator, high-side and low-side gate drivers and logic. In

19 FN6453.3
March 18, 2008
ISL6236A

addition, SMPS2 can also use REFIN2 to track its output from as measured by the VIN input and proportional to the output
0.5V to 2.50V. The ISL6236A contains fault-protection circuits voltage. This algorithm results in a nearly constant switching
that monitor the main PWM outputs for undervoltage and frequency despite the lack of a fixed-frequency clock
overvoltage conditions. A power-on sequence block controls generator. The benefit of a constant switching frequency is
the power-up timing of the main PWMs and monitors the that the frequency can be selected to avoid noise-sensitive
outputs for undervoltage faults. The ISL6236A includes an frequency regions, as shown in Equation 1:
adjustable low drop-out linear regulator. The bias generator K ( V OUT + I LOAD ⋅ r DS ( ON ) ( LOWERQ ) )
t ON = ---------------------------------------------------------------------------------------------------------- (EQ. 1)
blocks include the linear regulator, 3.3V precision reference, 2V V IN
precision reference and automatic bootstrap switchover circuit.
See Table 2 for approximate K- factors. Switching frequency
The synchronous-switch gate drivers are directly powered
increases as a function of load current due to the increasing
from PVCC, while the high-side switch gate drivers are
drop across the synchronous rectifier, which causes a faster
indirectly powered from PVCC through an external capacitor
inductor-current discharge ramp. On-times translate only
and an internal Schottky diode boost circuit.
roughly to switching frequencies. The on-times established
An automatic bootstrap circuit turns off the LDO linear in the “Electrical Specifications” table on page 4 are
regulator and powers the device from BYP if LDOREFIN is influenced by switching delays in the external high-side
set to GND or VCC. See Table 1. power MOSFET. Also, the dead-time effect increases the
TABLE 1. LDO OUTPUT VOLTAGE TABLE
effective on-time, reducing the switching frequency. It occurs
only in PWM mode (SKIP = VCC) and during dynamic output
LDO VOLTAGE CONDITIONS COMMENT voltage transitions when the inductor current reverses at
VOLTAGE at BYP LDOREFIN < 0.3V, Internal LDO is light or negative load currents. With reversed inductor
BYP > 4.63V disabled. current, the inductor's EMF causes PHASE to go high earlier
VOLTAGE at BYP LDOREFIN > VCC - 1V, Internal LDO is than normal, extending the on-time by a period equal to the
BYP > 3V disabled. UGATE-rising dead time.
5V LDOREFIN < 0.3V, Internal LDO is TABLE 2. APPROXIMATE K-FACTOR ERRORS
BYP < 4.63V active.
SWITCHING APPROXIMATE
3.3V LDOREFIN > VCC - 1V, Internal LDO is FREQUENCY K-FACTOR K-FACTOR
BYP < 3V active. SMPS (kHz) (µs) ERROR (%)
2 x LDOREFIN 0.35V < LDOREFIN < 2.25V Internal LDO is (tON = GND, REF, 400 2.5 ±10
active. or OPEN), VOUT1

(tON = GND), 500 2.0 ±10

FREE-RUNNING, CONSTANT ON-TIME PWM VOUT2
CONTROLLER WITH INPUT FEED-FORWARD (tON = VCC), 200 5.0 ±10
The constant on-time PWM control architecture is a VOUT1
pseudo-fixed-frequency, constant on-time, current-mode (tON = VCC, REF, 300 3.3 ±10
type with voltage feed forward. The constant on-time PWM or OPEN), VOUT2
control architecture relies on the output ripple voltage to
provide the PWM ramp signal; thus the output filter For loads above the critical conduction point, the actual
capacitor's ESR acts as a current-feedback resistor. The switching frequency is:
high-side switch on-time is determined by a one-shot whose
V OUT + V DROP1
period is inversely proportional to input voltage and directly f = ------------------------------------------------------- (EQ. 2)
t ON ( V IN + V DROP2 )
proportional to output voltage. Another one-shot sets a
minimum off-time (300ns typ). The on-time one-shot triggers
where:
when the following conditions are met: the error comparator's
output is high, the synchronous rectifier current is below the • VDROP1 is the sum of the parasitic voltage drops in the
current-limit threshold, and the minimum off time one-shot inductor discharge path, including synchronous rectifier,
has timed out. The controller utilizes the valley point of the inductor and PC board resistances
output ripple to regulate and determine the off-time. • VDROP2 is the sum of the parasitic voltage drops in the
ON-TIME ONE-SHOT (tON) charging path, including high-side switch, inductor and PC
board resistances
Each PWM core includes a one-shot that sets the high-side
• tON is the on-time calculated by the ISL6236A
switch on-time for each controller. Each fast, low-jitter,
adjustable one-shot includes circuitry that varies the on-time
in response to battery and output voltage. The high-side
switch on-time is inversely proportional to the battery voltage

20 FN6453.3
March 18, 2008
ISL6236A

VIN: 5.5V TO 25V

5V
C5
1µF
C8
1µF
PVCC VCC LDO NC

VIN LDOREFIN GND

C10 C1
10µF BOOT1 BOOT2 10
10µF

Q3a Q1
SI4816BDY UGATE1 UGATE2 IRF7821
OUT1 – PCI-e C9 C4 OUT2-GFX
L1: 3.3µH L2: 2.2µH TRACK REFIN2/10A
1.25V/5A 0.1µF 0.22µF
PHASE1 PHASE2

Q3b Q2
C11 LGATE2 C2
LGATE1
330µF IRF7832
2 x 330µF
9mΩ 4mΩ
6.3V OUT1 PGND 6.3V
VCC
R1 OUT2
EN1
7.87kΩ
5V BYP VCC 2 BITS
ISL6236A EN2
DAC
REFIN2: DYNAMIC 0V TO 2.5V
FB1 REFIN2 TIED TO VREF3 = 1.05V
FB1 TIED TO GND = 5V
FB1 TIED TO VCC = 1.5V REFIN2 TIED TO VCC = 3.3V + -
AGND REFIN2 + DROOP
R2 R3 R5
200kΩ +
10kΩ 200kΩ
ILIM1 ILIM2
C3 VCC VCC
OPEN
SKIP VREF3
C7
R4 R6
GND EN LDO 0.1µF
REF 200kΩ 200kΩ

VCC SECFB POK1

VCC TON POK2

PAD

FREQUENCY-DEPENDENT COMPONENTS

1.25V/1.05V SMPS tON = VCC

SWITCHING
FREQUENCY 200kHz/300kHz

L1 3.3µH

L2 2.7µH

C2 2 x 330µF

C11 330µF

FIGURE 66. ISL6236A TYPICAL DYNAMIC GFX APPLICATION CIRCUIT

21 FN6453.3
March 18, 2008
ISL6236A

VIN: 5.5V TO 25V

5V
C5
1µF
LDOREFIN TIED TO GND = 5V
C8 LDOREFIN TIED TO VCC = 3.3V
1µF LDO
PVCC VCC LDO
VCC C6
VIN LDOREFIN F
4.7µF
C10 C1
10µF SI4816BDY BOOT1 BOOT2 10
10µF
Q3a
UGATE2 Q1a
UGATE1
OUT1 C9 C4 OUT2
L1: 3.3µH L2: 2.2µF 1.05V/5A
1.5V/5A 0.1µF 0.22µF
PHASE1 PHASE2

Q3b
C11 LGATE2 Q1b SI4816BDY C2
LGATE1
330µF 330µF
9mΩ 4mΩ
6.3V OUT1 PGND 6.3V
VCC
EN1 OUT2

3.3V BYP ISL6236A VCC

EN2
VCC
FB1 TIED TO GND = 5V FB1 REFIN2: DYNAMIC 0V TO 2.5V
FB1 TIED TO VCC = 1.5V REFIN2 TIED TO VREF3 = 1.05V
REFIN2 VREF3 REFIN2 TIED TO VCC = 3.3V
R3 AGND
R5
200kΩ 200kΩ
ILIM1 ILIM2
C3 VCC VCC
0.01µF
SKIP VREF3
C7
ON R4 R6
EN LDO 0.1µF 200kΩ
REF 200kΩ
OFF

VCC SECFB POK1

VCC TON POK2

PAD

FREQUENCY-DEPENDENT COMPONENTS

1.5V/1.05V SMPS tON = VCC

SWITCHING
FREQUENCY 200kHz/300kHz

L1 3.3µH

L2 2.7µH

C2 330µF

C11 330µF

FIGURE 67. ISL6236A TYPICAL SYSTEM REGULATOR APPLICATION CIRCUIT WITHOUT CHARGE PUMP

22 FN6453.3
March 18, 2008
ISL6236A

VIN: 4.5V TO 5.5V*

R7 C5
10Ω 1µF

LDOREFIN TIED TO GND = 5V

C8 LDOREFIN TIED TO VCC = 3.3V
1µF
LDO
PVCC VCC LDO
VCC C6
VIN LDOREFIN F
4.7µF
C10 C1
10µF SI4816BDY BOOT1 BOOT2 10
10µF
Q3a
UGATE2 Q1a
UGATE1
OUT1 C9 C4 OUT2
L1: 3.3µH L2: 2.2µF 1.05V/5A
1.5V/5A 0.1µF 0.22µF
PHASE1 PHASE2

Q3b
C11 LGATE2 Q1b SI4816BDY C2
LGATE1
330µF 330µF
9mΩ 4mΩ
6.3V OUT1 PGND 6.3V
VCC
EN1 OUT2

3.3V BYP ISL6236A VCC

EN2
VCC
FB1 TIED TO GND = 5V FB1 REFIN2: DYNAMIC 0 TO 2.5V
FB1 TIED TO VCC = 1.5V REFIN2 TIED TO VREF3 = 1.05V
REFIN2 VREF3 REFIN2 TIED TO VCC = 3.3V
AGND
R3 R5
200kΩ 200kΩ
ILIM1 ILIM2
C3 VCC VCC
0.01µF
SKIP VREF3
ON C7
R4 R6
EN LDO 0.1µF 200kΩ
REF 200kΩ
OFF

VCC SECFB POK1

VCC TON POK2

PAD

*NOTE: To prevent noise in high loading conditions, place a 10Ω resistor between VIN and PVCC. Additional
electrolytic bulk decoupling can also be used for this purpose

FREQUENCY-DEPENDENT COMPONENTS

1.5V/1.05V SMPS tON = VCC

SWITCHING
FREQUENCY 200kHz/300kHz

L1 3.3µH

L2 2.7µH

C2 330µF

C11 330µF

FIGURE 68. ISL6236A TYPICAL LOW INPUT VOLTAGE SYSTEM REGULATOR APPLICATION CIRCUIT WITHOUT CHARGE PUMP

23 FN6453.3
March 18, 2008
ISL6236A

VIN: 5.5V TO 25V

C5
1µF
LDOREFIN TIED TO GND = 5V
LDOREFIN TIED TO VCC = 3.3V
VCC LDO
PVCC LDO
C6
VIN LDOREFIN
4.7µF
C10 C1
10µF BOOT1 BOOT2 10µF
10

Q3 Q1
IRF7807V UGATE1 UGATE2 IRF7821
OUT1 C9 C4 OUT2
L1: 4.7µH 0.1µF L2: 4.7µH 3.3V/11A
5V/7A 0.1µF
PHASE1 PHASE2

Q4 Q2
C11 LGATE1 LGATE2 C2
IRF7811AV IRF7832
330µF 330µF
9mΩ 9mΩ
6.3V OUT1 PGND 4V
VCC
D3 EN1 ISL6236A OUT2

D1 BYP EN2 VCC

REFIN2: DYNAMIC 0V TO 2V
C8 FB1 REFIN2 TIED TO VREF3 = 1.05V
C12 0.1µF FB1 TIED TO GND = 5V
D1a FB1 TIED TO VCC = 1.5V REFIN2 TIED TO VCC = 3.3V
0.1µF AGND REFIN2 VCC
D1b R3 R5
200kΩ 150kΩ
ILIM1 ILIM2
C3 VCC VCC
C14
OPEN
0.1µF SKIP VREF3
CP D2a
ON C7
14V/10mA D2b R4 R6
EN LDO 0.1µF 200kΩ
REF 200kΩ
D2 OFF
C15
0.1µF SECFB POK1
R1 GND
200kΩ R2 TON POK2
39.2kΩ
PAD

FREQUENCY-DEPENDENT COMPONENTS

tON = REF
5V/3.3V SMPS
tON = VCC (OR OPEN) tON = GND
SWITCHING
FREQUENCY 200kHz/300kHz 400kHz/300kHz 400kHz/500kHz

L1 6.8µH 6.8µH 4.7µH

L2 7.6µH 4.7µH 4.7µH

C2 2x470µF 2x330µF 2x330µF

C11 330µF 330µF 330µF

FIGURE 69. ISL6236A TYPICAL SYSTEM REGULATOR APPLICATION CIRCUIT WITH 14V CHARGE PUMP

24 FN6453.3
March 18, 2008
ISL6236A

TON SKIP

BOOT1 BOOT2

UGATE1 UGATE2

PHASE1 PHASE2
PVCC PVCC

LGATE1 SMPS1 SMPS2

SYNCHRONOUS LGATE2
SYNCHRONOUS
PWM BUCK PWM BUCK
GND CONTROLLER CONTROLLER
PGND

ILIM1 ILIM2
EN1 EN2
SECFB
FB1 POK1 POK2 REFIN2

OUT1 OUT2
OUT1 OUT2

BYP POK2
SW THRESHOLD
-

POK1

LDO
LDO
VCC
LDOREFIN INTERNAL
LOGIC
10Ω

VIN

PVCC
EN LDO
VREF3
POWER-ON VREF3
EN1 SEQUENCE
SQUENCE
CLEAR FAULT
CLEAR FAULT
LATCH
EN2 LATCH THERMAL
THERMAL REF
SHUTDOWN
SHUTDOWN REF

FIGURE 70. DETAILED FUNCTIONAL DIAGRAM ISL6236A

25 FN6453.3
March 18, 2008
ISL6236A

tON
MIN. tOFF
VIN Q TRIG
ONE SHOT
+ TO UGATE DRIVER
OUT Q
R Q
S Q
Q
REFIN2 (SMPS2)+
+
VREF +
ILIM +
COMP
+ SLOPE COMP +
BOOT BOOT
UV
5µA DETECT

VCC +
TO LGATE DRIVER
Σ
Â
+
PHASE
OUT
+ Q
S Q
R Q
Q

SKIP ONE-SHOT

SECFB
+
2V
FB + SMSP1 ONLY
PGOOD
DECODER 0.9VREF

FB +
OV LATCH
1.1VREF FAULT
FAULT
LATCH
UV LATCH LATCH
LOGIC
+
20ms
0.7VREF
BLANKING

FIGURE 71. PWM CONTROLLER (ONE SIDE ONLY)

26 FN6453.3
March 18, 2008
ISL6236A

Automatic Pulse-Skipping Switchover conduction, the output voltage has a DC regulation higher
(Idle Mode) than the trip level by 50% of the ripple. In discontinuous
conduction (SKIP = GND, light load), the output voltage has
In Idle Mode (SKIP = GND), an inherent automatic
a DC regulation higher than the trip level by approximately
switchover to PFM takes place at light loads. This switchover
1.0% due to slope compensation.
is affected by a comparator that truncates the low-side
switch on-time at the inductor current's zero crossing. This Forced-PWM Mode
mechanism causes the threshold between pulse-skipping
The low-noise, forced-PWM (SKIP = VCC) mode disables
PFM and nonskipping PWM operation to coincide with the
the zero-crossing comparator, which controls the low-side
boundary between continuous and discontinuous
switch on-time. Disabling the zero-crossing detector causes
inductor-current operation (also known as the critical
the low-side, gate-drive waveform to become the
conduction point):
complement of the high-side, gate-drive waveform. The
K ⋅ V OUT V IN – V OUT
I LOAD ( SKIP ) = ------------------------ -------------------------------- (EQ. 3) inductor current reverses at light loads as the PWM loop
2⋅L V IN
strives to maintain a duty ratio of VOUT/VIN. The benefit of
forced-PWM mode is to keep the switching frequency fairly
where K is the on-time scale factor (see “ON-TIME ONE-
constant, but it comes at a cost: the no-load battery current
SHOT (tON)” on page 20). The load-current level at which
can be 10mA to 50mA, depending on switching frequency
PFM/PWM crossover occurs, ILOAD(SKIP), is equal to half
and the external MOSFETs.
the peak-to-peak ripple current, which is a function of the
inductor value (Figure 72). For example, in the ISL6236A Forced-PWM mode is most useful for reducing
typical application circuit with VOUT1 = 5V, VIN = 12V, audio-frequency noise, improving load-transient response,
L = 7.6µH, and K = 5µs, switchover to pulse-skipping providing sink-current capability for dynamic output voltage
operation occurs at ILOAD = 0.96A or about on-fifth full load. adjustment, and improving the cross-regulation of
The crossover point occurs at an even lower value if a multiple-output applications that use a flyback transformer or
swinging (soft-saturation) inductor is used. coupled inductor.

ΔI V IN -V OUT Enhanced Ultrasonic Mode

=
t L (25kHz (min) Pulse Skipping)
I PEAK
Leaving SKIP unconnected or connecting SKIP to REF
INDUCTOR CURRENT

activates a unique pulse-skipping mode with a minimum

switching frequency of 25kHz. This ultrasonic pulse-skipping
mode eliminates audio-frequency modulation that would
ILOAD = IPEAK/2
otherwise be present when a lightly loaded controller
automatically skips pulses. In ultrasonic mode, the controller
automatically transitions to fixed-frequency PWM operation
when the load reaches the same critical conduction point
(ILOAD(SKIP)).
An ultrasonic pulse occurs when the controller detects that
0 ON-TIME TIME
no switching has occurred within the last 20µs. Once
FIGURE 72. ULTRASONIC CURRENT WAVEFORMS triggered, the ultrasonic controller pulls LGATE high, turning
on the low-side MOSFET to induce a negative inductor
The switching waveforms may appear noisy and current. After FB drops below the regulation point, the
asynchronous when light loading causes pulse-skipping controller turns off the low-side MOSFET (LGATE pulled low)
operation, but this is a normal operating condition that and triggers a constant on-time (UGATE driven high). When
results in high light-load efficiency. Trade-offs in PFM noise the on-time has expired, the controller re-enables the
vs light-load efficiency are made by varying the inductor low-side MOSFET until the controller detects that the
value. Generally, low inductor values produce a broader inductor current dropped below the zero-crossing threshold.
efficiency vs load curve, while higher values result in higher Starting with a LGATE pulse greatly reduces the peak output
full-load efficiency (assuming that the coil resistance remains voltage when compared to starting with a UGATE pulse, as
fixed) and less output voltage ripple. Penalties for using long as VFB < VREF, LGATE is off and UGATE is on, similar
higher inductor values include larger physical size and to pure SKIP mode.
degraded load-transient response (especially at low
input-voltage levels).

DC output accuracy specifications refer to the trip level of the

error comparator. When the inductor is in continuous

27 FN6453.3
March 18, 2008
ISL6236A

D2a. Finally, C14 charges C15 thru D2b when LGATE1

40µs (MAX)
switched to high. CP output voltage is shown in Equation 4:
INDUCTOR CP = V OUT1 + 2 ⋅ V LGATE1 – 4 ⋅ V D (EQ. 4)
CURRENT

where:

• VLGATE1 is the peak voltage of the LGATE1 driver

ZERO-CROSSING
Zero-Crossing
DETECTION • VD is the forward diode dropped across the Schottkys
Detection

0A SECFB is used to monitor the charge pump through the

resistive divider. In an event when SECFB dropped below
FB<REG.POINT
FB<Reg.Point 2V, the detection circuit force the highside MOSFET
(SMPS1) off and the lowside MOSFET (SMPS1) on for
) )
ON-TIME (tON)ON 300ns to allow CP to recharge and SECFB rise above 2V. In
the event of an overload on CP where SECFB cannot reach
FIGURE 73. ULTRASONIC CURRENT WAVEFORMS more than 2V, the monitor will be deactivated. Special care
should be taken to ensure enough normal voltage ripple on
Reference and Linear Regulators (VREF3, each cycle as to prevent CP shut-down. The SECFB pin has
REF, LDO and 14V Charge Pump) ~17mV of hysteresis, so the ripple should be enough to bring
The 3.3V reference (VREF3) is accurate to ±1.5% the SECFB voltage above the threshold by ~3x the
over-temperature, making VREF3 useful as a precision hysteresis, or (2V + 3*17mV) = 2.051V. Reducing the CP
system reference. VREF3 can supply up to 5mA for external decoupling capacitor and placing a small ceramic capacitor
loads. Bypass VREF3 to GND with a 0.01µF capacitor. (10pF to 47pF) in parallel with the upper leg of the SECFB
Leave it open if there is no load. resistor feedback network (R1 of Figure 68), will also
increase the robustness of the charge pump.
The 2V reference (REF) is accurate to ±1% over- temperature,
also making REF useful as a precision system reference. Current-Limit Circuit (ILIM ) with rDS(ON)
Bypass REF to GND with a 0.1µF (min) capacitor. REF can
Temperature Compensation
supply up to 50µA for external loads.
The current-limit circuit employs a "valley" current-sensing
An internal regulator produces a fixed 5V algorithm. The ISL6236A uses the ON-resistance of the
(LDOREFIN < 0.2V) or 3.3V (LDOREFIN > VCC - 1V). In an synchronous rectifier as a current-sensing element. If the
adjustable mode, the LDO output can be set from 0.7V to magnitude of the current-sense signal at PHASE is above
4.5V. The LDO output voltage is equal to two times the the current-limit threshold, the PWM is not allowed to initiate
LDOREFIN voltage. The LDO regulator can supply up to a new cycle. The actual peak current is greater than the
100mA for external loads. Bypass LDO with a minimum current-limit threshold by an amount equal to the inductor
4.7µF ceramic capacitor. When the LDOREFIN < 0.2V and ripple current. Therefore, the exact current-limit
BYP voltage is 5V, the LDO bootstrap-switchover to an characteristic and maximum load capability are a function of
internal 0.7Ω P-Channel MOSFET switch connects BYP to the current-limit threshold, inductor value and input and
LDO pin while simultaneously shutting down the internal output voltage.
linear regulator. These actions bootstrap the device,
powering the loads from the BYP input voltages, rather than
through internal linear regulators from the battery. Similarly,
when the BYP = 3.3V and LDOREFIN = VCC, the LDO I PEAK
bootstrap-switchover to an internal 1.5Ω P-Channel
MOSFET switch connects BYP to LDO pin while
INDUCTOR CURRENT

I LOAD
simultaneously shutting down the internal linear regulator. ΔI
No switchover action in adjustable mode.
I LIMIT
In Figure 68, the external 14V charge pump is driven by I LOAD(MAX)
LGATE1. When LGATE1 is low, D1a charged C8 sourced
from OUT1. C8 voltage is equal to OUT1 minus a diode ΔI
ILIM (VAL) = I
drop. When LGATE1 transitions to high, the charges from C8 LOAD - 2
will transfer to C12 through D1b and charge it to VLGATE1
plus VC8. As LGATE1 transitions low on the next cycle, C12 TIME
will charge C14 to its voltage minus a diode drop through
FIGURE 74. “VALLEY” CURRENT LIMIT THRESHOLD POINT

28 FN6453.3
March 18, 2008
ISL6236A

For lower power dissipation, the ISL6236A uses the

ON-resistance of the synchronous rectifier as the 5V
5V
current-sense element. Use the worst-case maximum value
BOOT
for rDS(ON) from the MOSFET data sheet. Add some margin 10Ω
10
VIN
for the rise in rDS(ON) with temperature. A good general rule
is to allow 0.5% additional resistance for each °C of UGATE
Q1
temperature rise. The ISL6236A controller has a built-in 5µA
current source as shown in Figure 75. Place the hottest C BOOT
OUT
power MOSEFTs as close to the IC as possible for best
thermal coupling. The current limit varies with the PHASE
ON-resistance of the synchronous rectifier. When combined ISL88732
ISL88733
with the undervoltage-protection circuit, this current-limit ISL6236A
ISL6236A
ISL88734
method is effective in almost every circumstance.
FIGURE 76. REDUCING THE SWITCHING-NODE RISE TIME

The internal pull-down transistors that drive LGATE low have

ILIM a 0.6Ω typical ON-resistance. These low ON-resistance
+
pull-down transistors prevent LGATE from being pulled up
during the fast rise time of the inductor nodes due to
5µA capacitive coupling from the drain to the gate of the low-side
+
synchronous-rectifier MOSFETs. However, for high-current
RILIM VILIM 9R TO CURRENT applications, some combinations of high- and low-side
VCC LIMIT LOGIC
MOSFETs may cause excessive gate-drain coupling, which
leads to poor efficiency and EMI-producing shoot-through
R
currents. Adding a 1Ω resistor in series with BOOT
increases the turn-on time of the high-side MOSFETs at the
expense of efficiency, without degrading the turn-off time
(see Figure 76).

Adaptive dead-time circuits monitor the LGATE and UGATE

FIGURE 75. CURRENT LIMIT BLOCK DIAGRAM
drivers and prevent either FET from turning on until the other
A negative current limit prevents excessive reverse inductor is fully off. This algorithm allows operation without shoot-
currents when VOUT sinks current. The negative through with a wide range of MOSFETs, minimizing delays
current-limit threshold is set to approximately 120% of the and maintaining efficiency. There must be low-resistance,
positive current limit and therefore tracks the positive current low-inductance paths from the gate drivers to the MOSFET
limit when ILIM is adjusted. The current-limit threshold is gates for the adaptive dead-time circuit to work properly.
adjusted with an external resistor for ISL6236A at ILIM. The Otherwise, the sense circuitry interprets the MOSFET gate
current-limit threshold adjustment range is from 20mV to as "off" when there is actually charge left on the gate. Use
200mV. In the adjustable mode, the current-limit threshold very short, wide traces measuring 10 to 20 squares (50 mils
voltage is 1/10th the voltage at ILIM. The voltage at ILIM pin to 100 mils wide if the MOSFET is 1” from the device).
is the product of 5µA*RILIM. The threshold defaults to
100mV when ILIM is connected to VCC. The logic threshold Boost-Supply Capacitor Selection (Buck)
for switch-over to the 100mV default value is approximately The boost capacitor should be 0.1µF to 4.7µF, depending on
VCC -1V. the input and output voltages, external components, and PC
The PC board layout guidelines should be carefully board layout. The boost capacitance should be as large as
observed to ensure that noise and DC errors do not corrupt possible to prevent it from charging to excessive voltage, but
the current-sense signals at PHASE. small enough to adequately charge during the minimum
low-side MOSFET conduction time, which happens at
MOSFET Gate Drivers (UGATE, LGATE) maximum operating duty cycle (this occurs at minimum input
voltage). The minimum gate to source voltage (VGS(MIN)) is
The UGATE and LGATE gate drivers sink 2.0A and 3.3A
determined by:
respectively of gate drive, ensuring robust gate drive for
C BOOT
high-current applications. The UGATE floating high-side V GS ( MIN ) = PVCC ⋅ --------------------------------------- (EQ. 5)
C BOOT + C GS
MOSFET drivers are powered by diode-capacitor charge
pumps at BOOT. The LGATE synchronous-rectifier drivers
are powered by PVCC.

29 FN6453.3
March 18, 2008
ISL6236A

where: OVERVOLTAGE PROTECTION

• PVCC is 5V When the output voltage of VOUT1 is 11% (16% for VOUT2)
above the set voltage, the overvoltage fault protection
• CGS is the gate capacitance of the high-side MOSFET activates. This latches on the synchronous rectifier MOSFET
with 100% duty cycle, rapidly discharging the output
Boost-Supply Refresh Monitor capacitor until the negative current limit is achieved. Once
In pure skip mode, the converter frequency can be very low negative current limit is met, UGATE is turned on for a
with little to no output loading. This produces very long off minimum on-time, followed by another LGATE pulse until
times, where leakage can bleed down the BOOT capacitor negative current limit. This effectively regulates the
voltage. If the voltage falls too low, the converter may not be discharge current at the negative current limit in an effort to
able to turn on UGATE when the output voltage falls to the prevent excessively large negative currents that cause
reference. To prevent this, the ISL6236A monitors the BOOT potentially damaging negative voltages on the load. Once an
capacitor voltage, and if it falls below 3V, it initiates an overvoltage fault condition is set, it can only be reset by
LGATE pulse, which will refresh the BOOT voltage. toggling SHDN, EN, or cycling VIN (POR).

POR, UVLO and Internal Digital Soft-Start UNDERVOLTAGE PROTECTION

Power-on reset (POR) occurs when VIN rises above When the output voltage drops below 70% of its regulation
approximately 3V, resetting the undervoltage, overvoltage, voltage for at least 100µs, the controller sets the fault latch
and thermal-shutdown fault latches. PVCC undervoltage and begins the discharge mode (see “Shutdown Mode” and
lockout (UVLO) circuitry inhibits switching when PVCC is “Discharge Mode” on page 30). UVP is ignored for at least
below 4V. LGATE is low during UVLO. The output voltages 20ms (typical), after start-up or after a rising edge on EN.
begin to ramp up once PVCC exceeds its 4V UVLO and REF Toggle EN or cycle VIN (POR) to clear the undervoltage fault
is in regulation. The internal digital soft-start timer begins to latch and restart the controller. UVP only applies to the buck
ramp up the maximum-allowed current limit during start-up. outputs.
The 1.7ms ramp occurs in five steps. The step size are 20%, THERMAL PROTECTION
40%, 60%, 80% and 100% of the positive current limit value.
The ISL6236A has thermal shutdown to protect the devices
Power-Good Output (POK) from overheating. Thermal shutdown occurs when the die
temperature exceeds +150°C. All internal circuitry shuts
The POK comparator continuously monitors both output
down during thermal shutdown. The ISL6236A may trigger
voltages for undervoltage conditions. POK is actively held
thermal shutdown if LDO is not bootstrapped from OUT
low in shutdown, standby, and soft-start. POK1 releases and
while applying a high input voltage on VIN and drawing the
digital soft-start terminates when VOUT1 outputs reach the
maximum current (including short circuit) from LDO. Even if
error-comparator threshold. POK1 goes low if VOUT1 output
LDO is bootstrapped from OUT, overloading the LDO causes
turns off or is 10% below its nominal regulation point. POK1
large power dissipation on the bootstrap switches, which
is a true open-drain output. Likewise, POK2 is used to
may result in thermal shutdown. Cycling EN, EN LDO, or
monitor VOUT2.
VIN (POR) ends the thermal-shutdown state.
Fault Protection Discharge Mode (Soft-Stop)
The ISL6236A provides overvoltage/undervoltage fault
When a transition to standby or shutdown mode occurs, or
protection in the buck controllers. Once activated, the
the output undervoltage fault latch is set, the outputs
controller continuously monitors the output for undervoltage
discharge to GND through an internal 25Ω switch. The
and overvoltage fault conditions.
reference remains active to provide an accurate threshold
OUT-OF-BOUND CONDITION and to provide overvoltage protection.
When the output voltage is 5% above the set voltage, the Shutdown Mode
out-of-bound condition activates. LGATE turns on until
The ISL6236A SMPS1, SMPS2 and LDO have independent
output reaches within regulation. Once the output is within
enabling control. Drive EN1, EN2 and EN LDO below the
regulation, the controller will operate as normal. It is the "first
precise input falling-edge trip level to place the ISL6236A in
line of defense" before OVP. The output voltage ripple must
its low-power shutdown state. The ISL6236A consumes only
be sized low enough as to not nuisance trip the OOB
20µA of quiescent current while in shutdown. When
threshold. The equations in “Output Capacitor Selection” on
shutdown mode activates, the 3.3V VREF3 remain on. Both
page 33 should be used to size the output voltage ripple
SMPS outputs are discharged to 0V through a 25Ω switch.
below 3% of the nominal output voltage set point.

30 FN6453.3
March 18, 2008
ISL6236A

Power-Up Sequencing and On/Off Controls (EN ) second SMPS remains on until the first SMPS turns off, the
EN1 and EN2 control SMPS power-up sequencing. EN1 or device shuts down, a fault occurs or PVCC goes into
EN2 rising above 2.4V enables the respective outputs. EN1 undervoltage lockout. Both supplies begin their power-down
or EN2 falling below 1.6V disables the respective outputs. sequence immediately when the first supply turns off. Driving
EN below 0.8V clears the overvoltage, undervoltage and
Connecting EN1 or EN2 to REF will force its outputs off while thermal fault latches.
the other output is below regulation. The sequenced SMPS
will start once the other SMPS reaches regulation. The

TABLE 3. OPERATING-MODE TRUTH TABLE

MODE CONDITION COMMENT

Power-Up PVCC < UVLO threshold. Transitions to discharge mode after a VIN POR and after REF becomes valid. LDO,
VREF3 and REF remain active.

Run EN LDO = high, EN1 or EN2 Normal operation

enabled.

Overvoltage Either output > 111% (VOUT1) or LGATE is forced high. LDO, VREF3 and REF active. Exited by a VIN POR, or by
Protection 116% (VOUT2) of nominal level. toggling EN1 or EN2.

Undervoltage Either output < 70% of nominal after The internal 25Ω switch turns on. LDO, VREF3 and REF are active. Exited by a VIN
Protection 20ms time-out expires and output is POR or by toggling EN1 or EN2.
enabled.

Discharge Either SMPS output is still high in Discharge switch (25Ω) connects OUT to GND. One output may still run while the
either standby mode or shutdown other is in discharge mode. Activates when PVCC is in UVLO, or transition to UVLO,
mode standby, or shutdown has begun. LDO, VREF3 and REF active.

Standby EN1, EN2 < start-up threshold, EN LDO, VREF3 and REF active.
LDO = High
Shutdown EN1, EN2, EN LDO = low Discharge switch (25Ω) connects OUT to PGND. All circuitry off except VREF3.

Thermal Shutdown TJ > +150°C All circuitry off. Exited by VIN POR or cycling EN. VREF3 remain active.

TABLE 4. SHUTDOWN AND STANDBY CONTROL LOGIS

VEN LDO VEN1 (V) VEN2 (V) LDO SMPS1 SMPS2

Low Low Low Off Off Off

“>2.5” → High Low Low On Off Off

“>2.5” → High High High On On On

“>2.5” → High High Low On On Off

“>2.5” → High Low High On Off On

“>2.5” → High High REF On On On (after SMPS1 is up)

“>2.5” → High REF High On On (after SMPS2 is up) On

31 FN6453.3
March 18, 2008
ISL6236A

Adjustable-Output Feedback (Dual-Mode FB) condition. The minimum practical inductor value is one
Connect FB1 to GND to enable the fixed 5V or tie FB1 to that causes the circuit to operate at critical conduction
(where the inductor current just touches zero with every
VCC to set the fixed 1.5V output. Connect a resistive
cycle at maximum load). Inductor values lower than this
voltage-divider at FB1 between OUT1 and GND to adjust the
grant no further size-reduction benefit.
respective output voltage between 0.7V and 5.5V (see
The ISL6236A pulse-skipping algorithm (SKIP = GND)
Figure 77). Choose R2 to be approximately 10k and solve for
initiates skip mode at the critical conduction point, so the
R1 using Equation 6.
inductor's operating point also determines the load
⎛ V OUT1 ⎞ current at which PWM/PFM switchover occurs. The
R 1 = R 2 ⋅ ⎜ ------------------- – 1⎟
⎝ V FB1 ⎠ optimum LIR point is usually found between 25% and
(EQ. 6)
50% ripple current.
where VFB1 = 0.7V nominal.
VIN
Likewise, connect REFIN2 to VCC to enable the fixed 3.3V
or tie REFIN2 to VREF3 to set the fixed 1.05V output. Set
UGATE1 Q3
REFIN2 from 0V to 2.50V for SMPS2 tracking mode (see
Figure 78).
OUT1
R 3 = R 4 ⋅ ⎛ ------------------- – 1⎞
VR
⎝V ⎠ ISL6236A
OUT2 (EQ. 7)
LGATE1 Q4
where:

• VR = 2V nominal (if tied to REF)

OUT1 R1
or
FB1
• VR = 3.3V nominal (if tied to VREF3)
R2
Design Procedure
Establish the input voltage range and maximum load current
before choosing an inductor and its associated ripple current
ratio (LIR). The following four factors dictate the rest of the FIGURE 77. SETTING VOUT1 WITH A RESISTOR DIVIDER
design:

1. Input Voltage Range. The maximum value (VIN(MAX))

must accommodate the maximum AC adapter voltage. VIN
The minimum value (VIN(MIN)) must account for the
lowest input voltage after drops due to connectors, fuses
Q1
and battery selector switches. Lower input voltages result UGATE2
UGATE2
UGATE
in better efficiency.
ISL88732
2. Maximum Load Current. The peak load current ISL6236A
ISL88733
OUT2
(ILOAD(MAX)) determines the instantaneous component ISL88734
stress and filtering requirements and thus drives output
capacitor selection, inductor saturation rating and the LGATE2
LGATE
LGATE2 Q2
design of the current-limit circuit. The continuous load
current (ILOAD) determines the thermal stress and drives
the selection of input capacitors, MOSFETs and other OUT2
VOUTOUT2
critical heat-contributing components.
3. Switching Frequency. This choice determines the basic FB
REFIN2
REFIN2 VR
trade-off between size and efficiency. The optimal R3
frequency is largely a function of maximum input voltage R4
and MOSFET switching losses.
4. Inductor Ripple Current Ratio (LIR). LIR is the ratio of
the peak-peak ripple current to the average inductor FIGURE 78. SETTING VOUT2 WITH A VOLTAGE DIVIDER FOR
current. Size and efficiency trade-offs must be TRACKING
considered when setting the inductor ripple current ratio.
Low inductor values cause large ripple currents, resulting
in the smallest size, but poor efficiency and high output
noise. Also, total output ripple above 3.5% of the output
regulation will cause controller to trigger out-of-bound

32 FN6453.3
March 18, 2008
ISL6236A

Inductor Selection Examining the 5A circuit example with a maximum

The switching frequency (on-time) and operating point rDS(ON) = 5mΩ at room temperature. At +125°C reveals the
(% ripple or LIR) determine the inductor value using following:
Equation 8: I LIMIT ( LOW ) = ( 25mV ) ⁄ ( ( 5mΩ × 1.2 ) > 5A – ( 0.35 ⁄ 2 )5A )
(EQ. 13)
V OUT_ ( V IN + V OUT_ )
L = --------------------------------------------------------------------- (EQ. 8)
V IN ⋅ f ⋅ LIR ⋅ I LOAD ( MAX ) 4.17A > 4.12A (EQ. 14)

Example: ILOAD(MAX) = 5A, VIN = 12V, VOUT2 = 5V, 4.17A is greater than the valley current of 4.12A, so the
f = 200kHz, 35% ripple current or LIR = 0.35: circuit can easily deliver the full-rated 5A using the 30mV
5V ( 12V – 5V ) (EQ. 9)
nominal current-limit threshold voltage.
L = ----------------------------------------------------------------- = 8.3μH
12V ⋅ 200kHz ⋅ 0.35 ⋅ 5A
Output Capacitor Selection
Find a low-loss inductor having the lowest possible DC The output filter capacitor must have low enough equivalent
resistance that fits in the allotted dimensions. Ferrite cores series resistance (ESR) to meet output ripple and
are often the best choice. The core must be large enough load-transient requirements, yet have high enough ESR to
not to saturate at the peak inductor current (IPEAK) shown in satisfy stability requirements. The output capacitance must
Equation 10: also be high enough to absorb the inductor energy while
IPEAK = I LOAD ( MAX ) + [ ( LIR ⁄ 2 ) ⋅ I LOAD ( MAX ) ] (EQ. 10) transitioning from full-load to no-load conditions without
tripping the overvoltage fault latch. In applications where the
output is subject to large load transients, the output
The inductor ripple current also impacts transient response
capacitor's size depends on how much ESR is needed to
performance, especially at low VIN - VOUT differences. Low
prevent the output from dipping too low under a load
inductor values allow the inductor current to slew faster,
transient. Ignoring the sag due to finite capacitance:
replenishing charge removed from the output filter capacitors
by a sudden load step. The peak amplitude of the output V DIP
R SER ≤ ---------------------------------- (EQ. 15)
transient (VSAG) is also a function of the maximum duty I LOAD ( MAX )
factor, which can be calculated from the on-time and
minimum off-time: where VDIP is the maximum-tolerable transient voltage drop.
⎛ ⎛ V OUT_ ⎞⎞ In non-CPU applications, the output capacitor's size
2
( ΔI LOAD ( MAX ) ) ⋅ L ⎜ K ⎜ ------------------- + t OFF ( MIN )⎟ ⎟ depends on how much ESR is needed to maintain an
⎝ ⎝ V IN ⎠⎠
VSAG = ---------------------------------------------------------------------------------------------------------------------------- acceptable level of output voltage ripple:
⎛ V IN – V OUT⎞
2 ⋅ C OUT ⋅ V OUT K ⎜ --------------------------------⎟ - t VP – P
⎝ V IN ⎠ OFF ( MIN ) R ESR ≤ ----------------------------------------------- (EQ. 16)
L IR ⋅ I LOAD ( MAX )
(EQ. 11)
where VP-P is the peak-to-peak output voltage ripple. The
where minimum off-time = 0.35µs (max) and K is from actual capacitance value required relates to the physical size
Table 2. needed to achieve low ESR, as well as to the chemistry of
Determining the Current Limit the capacitor technology. Thus, the capacitor is usually
selected by ESR and voltage rating rather than by
The minimum current-limit threshold must be great enough
capacitance value (this is true of tantalum, OS-CON, and
to support the maximum load current when the current limit
other electrolytic-type capacitors).
is at the minimum tolerance value. The valley of the inductor
current occurs at ILOAD(MAX) minus half of the ripple When using low-capacity filter capacitors such as polymer
current; therefore: types, capacitor size is usually determined by the capacity
required to prevent VSAG and VSOAR from tripping the
I LIMIT ( LOW ) > I LOAD ( MAX ) – [ ( LIR ⁄ 2 ) ⋅ I LOAD ( MAX ) ] (EQ. 12)
undervoltage and overvoltage fault latches during load
transients in ultrasonic mode.
where: ILIMIT(LOW) = minimum current-limit threshold
voltage divided by the rDS(ON) of Q2/Q4. For low input-to-output voltage differentials (VIN/ VOUT < 2),
additional output capacitance is required to maintain stability
Use the worst-case maximum value for rDS(ON) from the
and good efficiency in ultrasonic mode. The amount of
MOSFET Q2/Q4 data sheet and add some margin for the
overshoot due to stored inductor energy can be calculated
rise in rDS(ON) with temperature. A good general rule is to
as shown in Equation 17:
allow 0.2% additional resistance for each °C of temperature 2
rise. I PEAK ⋅ L
V SOAR = ------------------------------------------------ (EQ. 17)
2 ⋅ C OUT ⋅ V OUT_

where IPEAK is the peak inductor current.

33 FN6453.3
March 18, 2008
ISL6236A

Input Capacitor Selection

⎛ V OUT_ ⎞ 2
The input capacitors must meet the input-ripple-current PD ( Q H Resistance ) = ⎜ ------------------------⎟ ( I LOAD ) ⋅ r DS ( ON )
⎝ V IN ( MIN )⎠
(IRMS) requirement imposed by the switching current. The (EQ. 19)
ISL6236A dual switching regulator operates at different Generally, a small high-side MOSFET reduces switching
frequencies. This interleaves the current pulses drawn by losses at high input voltage. However, the rDS(ON) required
the two switches and reduces the overlap time where they to stay within package power-dissipation limits often limits
add together. The input RMS current is much smaller in how small the MOSFET can be. The optimum situation
comparison than with both SMPSs operating in phase. The occurs when the switching (AC) losses equal the conduction
input RMS current varies with load and the input voltage. (rDS(ON)) losses.
The maximum input capacitor RMS current for a single Switching losses in the high-side MOSFET can become an
SMPS is given by Equation 18: insidious heat problem when maximum battery voltage is
applied, due to the squared term in the CV2f switching-loss
⎛ V OUT ( V IN – V OUT_ )⎞
I RMS ≈ I LOAD ⎜ ------------------------------------------------------------⎟ (EQ. 18) equation. Reconsider the high-side MOSFET chosen for
⎝ V IN ⎠
adequate rDS(ON) at low battery voltages if it becomes
extraordinarily hot when subjected to VIN(MAX).
When V IN = 2 ⋅ V OUT_ ( D = 50% ) , IRMS has maximum
current of I LOAD ⁄ 2 . Calculating the power dissipation in NH (Q1/Q3) due to
switching losses is difficult since it must allow for quantifying
The ESR of the input-capacitor is important for determining
factors that influence the turn-on and turn-off times. These
capacitor power dissipation. All the power (IRMS2 x ESR)
factors include the internal gate resistance, gate charge,
heats up the capacitor and reduces efficiency. Nontantalum
threshold voltage, source inductance, and PC board layout
chemistries (ceramic or OS-CON) are preferred due to their
characteristics. The following switching-loss calculation
low ESR and resilience to power-up surge currents. Choose
provides only a very rough estimate and is no substitute for
input capacitors that exhibit less than +10°C temperature
bench evaluation, preferably including verification using a
rise at the RMS input current for optimal circuit longevity.
thermocouple mounted on NH (Q1/Q3):
Place the drains of the high-side switches close to each
other to share common input bypass capacitors. 2 ⎛ C RSS ⋅ f SW ⋅ I LOAD⎞
PD ( Q H Switching ) = ( V IN ( MAX ) ) ⎜ -----------------------------------------------------⎟
⎝ I GATE ⎠
Power MOSFET Selection (EQ. 20)
Most of the following MOSFET guidelines focus on the
challenge of obtaining high load-current capability (>5A) where CRSS is the reverse transfer capacitance of QH
when using high-voltage (>20V) AC adapters. Low-current (Q1/Q3) and IGATE is the peak gate-drive source/sink
applications usually require less attention. current.
Choose a high-side MOSFET (Q1/Q3) that has conduction For the synchronous rectifier, the worst-case power
losses equal to the switching losses at the typical battery dissipation always occurs at maximum battery voltage:
voltage for maximum efficiency. Ensure that the conduction ⎛ V OUT ⎞ 2
losses at the minimum input voltage do not exceed the PD ( Q L ) = ⎜ 1 – --------------------------⎟ I LOAD ⋅ r DS ( ON ) (EQ. 21)
⎝ V IN ( MAX )⎠
package thermal limits or violate the overall thermal budget.
Ensure that conduction losses plus switching losses at the
The absolute worst case for MOSFET power dissipation
maximum input voltage do not exceed the package ratings
occurs under heavy overloads that are greater than
or violate the overall thermal budget.
ILOAD(MAX) but are not quite high enough to exceed the
Choose a synchronous rectifier (Q2/Q4) with the lowest current limit and cause the fault latch to trip. To protect
possible rDS(ON). Ensure the gate is not pulled up by the against this possibility, "overdesign" the circuit to tolerate:
high-side switch turning on due to parasitic drain-to-gate I LOAD = I LIMIT ( HIGH ) + ( ( LIR ) ⁄ 2 ) ⋅ I LOAD ( MAX ) (EQ. 22)
capacitance, causing cross-conduction problems. Switching
losses are not an issue for the synchronous rectifier in the
where ILIMIT(HIGH) is the maximum valley current allowed
buck topology since it is a zero-voltage switched device
by the current-limit circuit, including threshold tolerance and
when using the buck topology.
resistance variation.
MOSFET Power Dissipation
Rectifier Selection
Worst-case conduction losses occur at the duty-factor
Current circulates from ground to the junction of both
extremes. For the high-side MOSFET, the worst-case power
MOSFETs and the inductor when the high-side switch is off.
dissipation (PD) due to the MOSFET's rDS(ON) occurs at the
As a consequence, the polarity of the switching node is
minimum battery voltage, as shown in Equation 19:
negative with respect to ground. This voltage is
approximately -0.7V (a diode drop) at both transition edges

34 FN6453.3
March 18, 2008
ISL6236A

while both switches are off (dead time). The drop is where VDROP1 and VDROP2 are the parasitic voltage drops
I L ⋅ r DS ( ON ) when the low-side switch conducts. in the discharge and charge paths (see “ON-TIME ONE-
SHOT (tON)” on page 20), tOFF(MIN) is from “Electrical
The rectifier is a clamp across the synchronous rectifier that
Specifications” table on page 6 and K is taken from Table 2.
catches the negative inductor swing during the dead time
The absolute minimum input voltage is calculated with h = 1.
between turning the high-side MOSFET off and the
synchronous rectifier on. The MOSFETs incorporate a Operating frequency must be reduced or h must be
high-speed silicon body diode as an adequate clamp diode if increased and output capacitance added to obtain an
efficiency is not of primary importance. Place a Schottky acceptable VSAG if calculated VIN(MIN) is greater than the
diode in parallel with the body diode to reduce the forward required minimum input voltage. Calculate VSAG to be sure
voltage drop and prevent the Q2/Q4 MOSFET body diodes of adequate transient response if operation near dropout is
from turning on during the dead time. Typically, the external anticipated.
diode improves the efficiency by 1% to 2%. Use a Schottky
Dropout Design Example:
diode with a DC current rating equal to one-third of the load
current. For example, use an MBR0530 (500mA-rated) type ISL6236A: With VOUT2 = 5V, fsw = 400kHz, K = 2.25µs,
for loads up to 1.5A, a 1N5817 type for loads up to 3A, or a tOFF(MIN) = 350ns, VDROP1 = VDROP2 = 100mV, and
1N5821 type for loads up to 10A. The rectifier's rated h = 1.5, the minimum VIN is:
reverse breakdown voltage must be at least equal to the
maximum input voltage, preferably with a 20% derating ( 5V + 0.1V ) (EQ. 24)
V IN ( MIN ) = ---------------------------------------------- + 0.1V – 0.1V = 6.65V
0.35μs ⋅ 1.5
factor. 1 – ⎛ -------------------------------⎞
⎝ 2.25μs ⎠

Applications Information Calculating with h = 1 yields:

Dropout Performance
( 5V + 0.1V ) (EQ. 25)
V IN ( MIN ) = ----------------------------------------- + 0.1V – 0.1V = 6.04V
The output voltage-adjust range for continuous-conduction 0.35μs ⋅ 1
1 – ⎛ --------------------------⎞
operation is restricted by the nonadjustable 350ns (max) ⎝ 2.25μs ⎠
minimum off-time one-shot. Use the slower 5V SMPS for the
higher of the two output voltages for best dropout Therefore, VIN must be greater than 6.65V. A practical input
performance in adjustable feedback mode. The duty-factor voltage with reasonable output capacitance would be 7.5V.
limit must be calculated using worst-case values for on- and PC Board Layout Guidelines
off-times, when working with low input voltages.
Careful PC board layout is critical to achieve minimal switching
Manufacturing tolerances and internal propagation delays
losses and clean, stable operation. This is especially true when
introduce an error to the tON K-factor. Also, keep in mind that
multiple converters are on the same PC board where one
transient-response performance of buck regulators operated
circuit can affect the other. Refer to the ISL6236 Evaluation Kit
close to dropout is poor, and bulk output capacitance must
Application Notes (AN1271 and AN1272) for a specific layout
often be added (see Equation 11 on page 33).
example.
The absolute point of dropout occurs when the inductor
Mount all of the power components on the top side of the board
current ramps down during the minimum off-time (ΔIDOWN)
with their ground terminals flush against one another, if
as much as it ramps up during the on-time ( ΔIUP). The ratio
possible. Follow these guidelines for good PC board layout:
h = ΔIUP/ΔIDOWN indicates the ability to slew the inductor
current higher in response to increased load, and must • Isolate the power components on the top side from the
always be greater than 1. As h approaches 1, the absolute sensitive analog components on the bottom side with a
minimum dropout point, the inductor current is less able to ground shield. Use a separate PGND plane under the
increase during each switching cycle and VSAG greatly OUT1 and OUT2 sides (called PGND1 and PGND2). Avoid
increases unless additional output capacitance is used. the introduction of AC currents into the PGND1 and PGND2
ground planes. Run the power plane ground currents on the
A reasonable minimum value for h is 1.5, but this can be top side only, if possible.
adjusted up or down to allow trade-offs between VSAG,
• Use a star ground connection on the power plane to
output capacitance and minimum operating voltage. For a
minimize the crosstalk between OUT1 and OUT2.
given value of h, the minimum operating voltage can be
calculated in Equation 23: • Keep the high-current paths short, especially at the
ground terminals. This practice is essential for stable,
( V OUT_ + V DROP ) jitter-free operation.
V IN ( MIN ) = --------------------------------------------------- + V DROP2 – V DROP1 (EQ. 23)
t OFF ( MIN ) ⋅ h
1 – ⎛ ------------------------------------⎞ • Keep the power traces and load connections short. This
⎝ K ⎠
practice is essential for high efficiency. Using thick copper
PC boards (2oz vs 1oz) can enhance full-load efficiency
by 1% or more. Correctly routing PC board traces must be

35 FN6453.3
March 18, 2008
ISL6236A

approached in terms of fractions of centimeters, where a Group the gate-drive components (BOOT capacitor, VIN
single mΩ of excess trace resistance causes a bypass capacitor) together near the controller device.
measurable efficiency penalty.
Make the DC/DC controller ground connections as follows:
• PHASE (ISL6236A) and GND connections to the
synchronous rectifiers for current limiting must be made 1. Near the device, create a small analog ground plane.
using Kelvin-sense connections to guarantee the 2. Connect the small analog ground plane to GND and use
current-limit accuracy with 8 Ld SO MOSFETs. This is best the plane for the ground connection for the REF and VCC
done by routing power to the MOSFETs from outside using bypass capacitors, FB dividers and ILIM resistors (if any).
the top copper layer, while connecting PHASE traces 3. Create another small ground island for PGND and use
inside (underneath) the MOSFETs. the plane for the VIN bypass capacitor, placed very close
• When trade-offs in trace lengths must be made, it is to the device.
preferable to allow the inductor charging path to be made 4. Connect the GND and PGND planes together at the
longer than the discharge path. For example, it is better to metal tab under device.
allow some extra distance between the input capacitors On the board's top side (power planes), make a star ground
and the high-side MOSFET than to allow distance
to minimize crosstalk between the two sides. The top-side
between the inductor and the synchronous rectifier or
star ground is a star connection of the input capacitors and
between the inductor and the output filter capacitor.
synchronous rectifiers. Keep the resistance low between the
• Ensure that the OUT connection to COUT is short and star ground and the source of the synchronous rectifiers for
direct. However, in some cases it may be desirable to accurate current limit. Connect the top-side star ground
deliberately introduce some trace length between the OUT (used for MOSFET, input, and output capacitors) to the small
connector node and the output filter capacitor.
island with a single short, wide connection (preferably just a
• Route high-speed switching nodes (BOOT, UGATE, via). Create PGND islands on the layer just below the
PHASE, and LGATE ) away from sensitive analog areas top-side layer (refer to ISL6236 Evaluation Kit Application
(REF, ILIM, and FB ). Use PGND1 and PGND2 as an EMI Notes, AN1271 and AN1272 for an example) to act as an EMI
shield to keep radiated switching noise away from the IC's shield if multiple layers are available (highly recommended).
feedback divider and analog bypass capacitors. Connect each of these individually to the star ground via,
• Make all pin-strap control input connections (SKIP, ILIM, which connects the top side to the PGND plane. Add one
etc.) to GND or VCC of the device. more solid ground plane under the device to act as an
additional shield, and also connect the solid ground plane to
Layout Procedure
the star ground via.
Place the power components first with ground terminals
adjacent (Q2/Q4 source, CIN, COUT ). If possible, make all Connect the output power planes (VCORE and system
these connections on the top layer with wide, copper-filled ground planes) directly to the output filter capacitor positive
areas. and negative terminals with multiple vias.

Mount the controller IC adjacent to the synchronous rectifier

MOSFETs close to the hottest spot, preferably on the back
side in order to keep UGATE, GND, and the LGATE gate
drive lines short and wide. The LGATE gate trace must be
short and wide, measuring 50 mils to 100 mils wide if the
MOSFET is 1” from the controller device.

All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems.
Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.

For information regarding Intersil Corporation and its products, see www.intersil.com

36 FN6453.3
March 18, 2008
ISL6236A

Package Outline Drawing

L32.5x5B
32 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
Rev 2, 11/07

4X 3.5
5.00 A 28X 0.50
6
B
25 32 PIN #1 INDEX AREA

6
PIN 1 24 1
INDEX AREA

5.00
3 .30 ± 0 . 15

17 8
(4X) 0.15
16 9
0.10 M C A B

+ 0.07
32X 0.40 ± 0.10 4 32X 0.23 - 0.05

TOP VIEW BOTTOM VIEW

SEE DETAIL "X"

0.10 C
0 . 90 ± 0.1 C
BASE PLANE
SEATING PLANE
0.08 C
( 4. 80 TYP )
( 28X 0 . 5 )
SIDE VIEW
( 3. 30 )

(32X 0 . 23 )

C 0 . 2 REF 5
( 32X 0 . 60)

0 . 00 MIN.
0 . 05 MAX.

TYPICAL RECOMMENDED LAND PATTERN DETAIL "X"

NOTES:

1. Dimensions are in millimeters.

Dimensions in ( ) for Reference Only.
2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994.
3. Unless otherwise specified, tolerance : Decimal ± 0.05

4. Dimension b applies to the metallized terminal and is measured

between 0.15mm and 0.30mm from the terminal tip.

5. Tiebar shown (if present) is a non-functional feature.

6. The configuration of the pin #1 identifier is optional, but must be

located within the zone indicated. The pin #1 identifier may be
either a mold or mark feature.

37 FN6453.3
March 18, 2008

High-Efficiency, Quad-Output, Main Power Supply Controllers For Notebook Computers Features
No ratings yet
High-Efficiency, Quad-Output, Main Power Supply Controllers For Notebook Computers Features
35 pages
Notebook Power Supply Guide
No ratings yet
Notebook Power Supply Guide
36 pages
Notebook Power Controller Guide
No ratings yet
Notebook Power Controller Guide
36 pages
37A65
No ratings yet
37A65
16 pages
Isl 6532 A
No ratings yet
Isl 6532 A
18 pages
Isl95833bhrtz V4PDe1i0 zZr67m72Z
No ratings yet
Isl95833bhrtz V4PDe1i0 zZr67m72Z
22 pages
Datasheet PDF
No ratings yet
Datasheet PDF
16 pages
Features: Two-Phase Buck PWM Controller With Integrated MOSFET Drivers For VRM9, VRM10, and AMD Hammer Applications
No ratings yet
Features: Two-Phase Buck PWM Controller With Integrated MOSFET Drivers For VRM9, VRM10, and AMD Hammer Applications
30 pages
ISL6255 Renesas
No ratings yet
ISL6255 Renesas
22 pages
ACPI Regulator/Controller For Dual Channel DDR Memory Systems Features
No ratings yet
ACPI Regulator/Controller For Dual Channel DDR Memory Systems Features
15 pages
4-Channel Integrated LCD Supply
No ratings yet
4-Channel Integrated LCD Supply
20 pages
Features: Monolithic 1A Step-Down Regulator With Low Quiescent Current
No ratings yet
Features: Monolithic 1A Step-Down Regulator With Low Quiescent Current
12 pages
Isl8560 Datasheet
No ratings yet
Isl8560 Datasheet
17 pages
Isl6261-Acrz Datasheet
No ratings yet
Isl6261-Acrz Datasheet
34 pages
ISL6255, ISL6255A: Highly Integrated Battery Charger With Automatic Power Source Selector For Notebook Computers Features
No ratings yet
ISL6255, ISL6255A: Highly Integrated Battery Charger With Automatic Power Source Selector For Notebook Computers Features
22 pages
Datasheet
No ratings yet
Datasheet
20 pages
ISL6269 Data Sheet
No ratings yet
ISL6269 Data Sheet
14 pages
97644IRZ - LG 17inch SQ Monitor Panel IC
No ratings yet
97644IRZ - LG 17inch SQ Monitor Panel IC
15 pages
Low Cost Multi-Chemistry Battery Charger Controller: ISL6251, ISL6251A
No ratings yet
Low Cost Multi-Chemistry Battery Charger Controller: ISL6251, ISL6251A
20 pages
EL7532 fn7435
No ratings yet
EL7532 fn7435
8 pages
ISL8560IRZ Datasheetz
No ratings yet
ISL8560IRZ Datasheetz
17 pages
14-Channel Programmable Switchable I C TFT-LCD Reference Voltage Generator With Integrated 4-Channel Static Gamma Drivers Features
No ratings yet
14-Channel Programmable Switchable I C TFT-LCD Reference Voltage Generator With Integrated 4-Channel Static Gamma Drivers Features
10 pages
ISL6565A, ISL6565B: Multi-Phase PWM Controller With Precision R or DCR Current Sensing For VR10.X Application Features
No ratings yet
ISL6565A, ISL6565B: Multi-Phase PWM Controller With Precision R or DCR Current Sensing For VR10.X Application Features
28 pages
Isl 88731 A
No ratings yet
Isl 88731 A
24 pages
Intermediate Bus PWM Controller Features: ISL6744A
No ratings yet
Intermediate Bus PWM Controller Features: ISL6744A
18 pages
ISL6263C Intersil
No ratings yet
ISL6263C Intersil
18 pages
REN Isl6269a DST 20040119
No ratings yet
REN Isl6269a DST 20040119
15 pages
Three-Phase Buck PWM Controller With Integrated MOSFET Drivers For VRM9, VRM10, and AMD Hammer Applications Features
No ratings yet
Three-Phase Buck PWM Controller With Integrated MOSFET Drivers For VRM9, VRM10, and AMD Hammer Applications Features
29 pages
Features: High Efficiency Synchronous Boost Converter With 4.2A Switches and Output Disconnect
No ratings yet
Features: High Efficiency Synchronous Boost Converter With 4.2A Switches and Output Disconnect
12 pages
Isl6545 A PDF
No ratings yet
Isl6545 A PDF
17 pages
Single-Phase Core Regulator For Imvp-6 Mobile Cpus Features: Fn9251.1 Data Sheet September 27, 2006
No ratings yet
Single-Phase Core Regulator For Imvp-6 Mobile Cpus Features: Fn9251.1 Data Sheet September 27, 2006
34 pages
ISL 6520B Step Down Converter (6520 BCB)
No ratings yet
ISL 6520B Step Down Converter (6520 BCB)
11 pages
Monolithic 6A DC/DC Step-Down Regulator Features: FN7102.7 Data Sheet May 8, 2006
No ratings yet
Monolithic 6A DC/DC Step-Down Regulator Features: FN7102.7 Data Sheet May 8, 2006
14 pages
Isl6609, Isl6609a
No ratings yet
Isl6609, Isl6609a
12 pages
Smbus Level 2 Battery Charger Features: Fn9258.1 Data Sheet January 21, 2009
No ratings yet
Smbus Level 2 Battery Charger Features: Fn9258.1 Data Sheet January 21, 2009
22 pages
Renesas Isl28191
No ratings yet
Renesas Isl28191
19 pages
Datasheet PDF
No ratings yet
Datasheet PDF
7 pages
Icl7650s Cba-1z
No ratings yet
Icl7650s Cba-1z
13 pages
Smbus Level 2 Battery Charger Features: Isl88731A
No ratings yet
Smbus Level 2 Battery Charger Features: Isl88731A
23 pages
Features: 600Khz/1.2Mhz PWM Step-Up Regulator
No ratings yet
Features: 600Khz/1.2Mhz PWM Step-Up Regulator
12 pages
Synchronous Rectified MOSFET Driver With Pre-Biased Load Startup Capability
No ratings yet
Synchronous Rectified MOSFET Driver With Pre-Biased Load Startup Capability
11 pages
Synchronous Rectified MOSFET Driver Features: FN9240.0 Data Sheet November 2, 2005
No ratings yet
Synchronous Rectified MOSFET Driver Features: FN9240.0 Data Sheet November 2, 2005
10 pages
Isl 6326
No ratings yet
Isl 6326
31 pages
ISL88731 A
No ratings yet
ISL88731 A
23 pages
Isl97634 1529394
No ratings yet
Isl97634 1529394
13 pages
Features: Two-Phase Core Controller For Amd Mobile Turion Cpus
No ratings yet
Features: Two-Phase Core Controller For Amd Mobile Turion Cpus
23 pages
Single 12V Input Supply Dual Regulator - Synchronous Rectified Buck PWM and Linear Power Controller Features
No ratings yet
Single 12V Input Supply Dual Regulator - Synchronous Rectified Buck PWM and Linear Power Controller Features
18 pages
Features: High Speed, Monolithic Pin Driver
No ratings yet
Features: High Speed, Monolithic Pin Driver
9 pages
Super Voltage Converter Features: ICL7660S
No ratings yet
Super Voltage Converter Features: ICL7660S
12 pages
55V, 1A Peak Current H-Bridge FET Driver Features: December 20, 2006 Data Sheet FN6382.0
No ratings yet
55V, 1A Peak Current H-Bridge FET Driver Features: December 20, 2006 Data Sheet FN6382.0
12 pages
ISL6314
No ratings yet
ISL6314
32 pages
Isl6610 Isl6610a
No ratings yet
Isl6610 Isl6610a
11 pages
Octal Voltage Level Shifter For TFT/LCD Panels: FN6196.0 Data Sheet October 21, 2005
No ratings yet
Octal Voltage Level Shifter For TFT/LCD Panels: FN6196.0 Data Sheet October 21, 2005
9 pages
D D D D: Description/ordering Information
No ratings yet
D D D D: Description/ordering Information
24 pages
REN Isl1219 DST 20040811
No ratings yet
REN Isl1219 DST 20040811
24 pages
Features: 4-Channel Integrated LCD Supply With Dual V Amplifiers
No ratings yet
Features: 4-Channel Integrated LCD Supply With Dual V Amplifiers
25 pages
tps73201 Ep
No ratings yet
tps73201 Ep
22 pages
WiFi Halow
No ratings yet
WiFi Halow
14 pages
BST B550 EN CLS Pro 600 2017 Low
No ratings yet
BST B550 EN CLS Pro 600 2017 Low
2 pages
Misra Rule
No ratings yet
Misra Rule
67 pages
Attivo PP3 A1430E
No ratings yet
Attivo PP3 A1430E
170 pages
EME-PG01: Pulse Generator Card For VFD-E Series Instruction Sheet
No ratings yet
EME-PG01: Pulse Generator Card For VFD-E Series Instruction Sheet
2 pages
Grentech DCS1800 ICS Repeater 10W 20W
No ratings yet
Grentech DCS1800 ICS Repeater 10W 20W
3 pages
Debug 1214
No ratings yet
Debug 1214
3 pages
V6R1 Memo To Users Rzaq9 (New Version)
No ratings yet
V6R1 Memo To Users Rzaq9 (New Version)
100 pages
Mitsubishi Radio Security Code Guide
No ratings yet
Mitsubishi Radio Security Code Guide
3 pages
Java Lab Exam for CS Students
No ratings yet
Java Lab Exam for CS Students
6 pages
Abbott PCA Plus and Plus II Service Manual
No ratings yet
Abbott PCA Plus and Plus II Service Manual
206 pages
3.6 IOT Protocols
No ratings yet
3.6 IOT Protocols
24 pages
Types of Signals
No ratings yet
Types of Signals
34 pages
Debugging Breakpoints Guide
No ratings yet
Debugging Breakpoints Guide
94 pages
Monster Airmax XKT01 MKII
No ratings yet
Monster Airmax XKT01 MKII
8 pages
Introduction To A/D Conversion
No ratings yet
Introduction To A/D Conversion
20 pages
Transistor NPN - Mmbt3904, Br3dg3904m RC Blue Rocket
No ratings yet
Transistor NPN - Mmbt3904, Br3dg3904m RC Blue Rocket
8 pages
Avoiding Storage Violations in CICS
No ratings yet
Avoiding Storage Violations in CICS
4 pages
Unreal Engine Plugin Assets
No ratings yet
Unreal Engine Plugin Assets
734 pages
Tarjeta Comunicación DC2000
100% (1)
Tarjeta Comunicación DC2000
16 pages
Ch4.Fault Simulation
No ratings yet
Ch4.Fault Simulation
68 pages
VMware Infrastructure 3 Pricing, Packaging and Licensing Overview
No ratings yet
VMware Infrastructure 3 Pricing, Packaging and Licensing Overview
10 pages
Shivam Mittal Resume
No ratings yet
Shivam Mittal Resume
1 page
Programming Abstraction CHO CS179
No ratings yet
Programming Abstraction CHO CS179
8 pages
Quiz 3: Instructions
No ratings yet
Quiz 3: Instructions
8 pages
LINUX Privilege Escalation
No ratings yet
LINUX Privilege Escalation
5 pages
Janardhan Reddy Lingam
No ratings yet
Janardhan Reddy Lingam
7 pages
Electronic Circuit Schematics Guide
No ratings yet
Electronic Circuit Schematics Guide
45 pages
27.2.10 Lab - Extract An Executable From A Pcap
No ratings yet
27.2.10 Lab - Extract An Executable From A Pcap
6 pages