DESIGN AND ANALYSIS OF COMPACT

MICROSTRIP PATCH ANTENNA FOR LTE

APPLICATIONS USING METAMATERIALS

A Thesis submitted to Gujarat Technological University

for the Award of

Doctor of Philosophy
in

Electronics and Communication Engineering

By

Tusharkumar Pravinchandra Dave

[Enrollment No. 149997111015]

under supervision of
Dr. Jagdish M. Rathod

GUJARAT TECHNOLOGICAL UNIVERSITY

AHMEDABAD
February–2020
i
 Abstract
The work presented in this thesis channel through the development of novel
microwave structures and techniques for realizing compact planar antennas for LTE-band
operation. The preliminary antenna designs were investigated by modelling microwave
patch antenna and loading metamaterial complementary split ring structures. The single
and dual-band resonant of the antenna confirm the advantage of the metamaterial-based
antenna for multi-band operation and compact. Subsequently, defective grounding
technique was analysed and modeled in the antenna design for obtaining a narrow and
selective frequency band.

Further, the design of a miniaturized dual-band planar antenna for LTE application
and the generation of narrow triple bands using the stacking concept of dielectric resonator
and metamaterial unit-cell were developed. In the basic antenna design, split ring resonator
was loaded in the radiating plane of the patch and frequency of resonance was further
modified with the help of E-shaped stub. The antenna has been fabricated using FR-4
substrate and the measured dual bands at 2.11 GHz, and 2.665 GHz are found in a close
match with the simulated data. By placing a thin dielectric resonator of permittivity εr =
10.2 and thickness of 1.27 mm, the separation between the bands is reduced, and two
closely spaced narrow bands are obtained at 2.217 GHz and 2.28 GHz. A novel
metamaterial unit-cell having near-zero refractive index is designed and mounted above the
dielectric resonator. This stack configuration generates triple narrow frequency band in the
LTE 2 GHz spectrum range. The overall size of the proposed antenna is 20 × 25 mm 2 and
found suitable for narrowband communication in the LTE spectrum.

In the final stage, a compact antenna design with wide spectral frequency diversity in
a wireless system was implemented. Planar antenna using defective ground and coarse
frequency switching with connected split-ring resonator through PIN diodes are presented.
The antenna design is based on connected radial stubs which form a patch. Slots in the
bottom plane create a corrugation-like structure. This results in the defective ground, which
contributes to the localization of electric flux at the different areas in the antenna plane for
multiband operation. The antenna was simulated and fabricated on a 1.6 mm thick FR-4
substrate. The coarse tuning of the frequency band is tested and compared with the

xi
simulation data. The measured results depict broadside radiation pattern at different
frequency bands (like 0.96 GHz, 2.63 GHz, 3.22 GHz) and with a peak gain of 3.72 dBi.
The compact size (45 × 45 mm2) and diversity frequency resonance up to pentaband in the
antenna makes it suitable for LTE and WLAN/WiMAX application systems.

From the proposed antenna designing techniques and their practical

implementation, it is observed that modifying the antenna topology using metamaterial,
dielectric resonator, defect ground and metering stubs is proven as a goal-oriented
approach for planar LTE band microstrip antennas.

xii
 Acknowledgment
Firstly I would like to thank the almighty God for providing me the opportunity and
ability to complete this research work.
I am deeply gratified to have supervisor Dr. J.M.Rathod, Assoc. Prof. Electronics
Engineering Department, Birla Vishvakarma Engineering College, Vallabh Vidyanagar.
His continuous encouragement and guidance inspire me to work consistently. He shared
his valuable experience in so many occasions and guided me to complete this work
effectively.
I am very thankful to Doctoral Progress Committee (DPC) members: Dr. Jaymin K.
Bhalani, Professor, Babaria Institute of Technology, Baroda and Dr. K.G.Maradia,
Professor and Head, EC Department, Government Engineering College, Gandhinagar for
their inputs and support during reviews. I am very grateful for their suggestions and critical
reviews that enable me to improve this work.
I am also very thankful to Dr. Indrajit N. Patel, Principal, Birla Vishvakarma
Engineering College, Vallabh Vidyanagar, and Dr. Tanmay D. Pawar, Head of
Electronics Engineering Department, Birla Vishvakarma Engineering College, Vallabh
Vidyanagar to allow me to conduct my research and experimentation work at ELARC-
ELectromagnetics and Antenna Research Centre which is operated by Electronics
Department of BVM Engineering College, Vallabh Vidya Nagar, Gujarat, India. They
always provided me supporting staff and necessary measuring equipments whenever I
required for research work.
I am very thankful to my parents for all their support during the tenure of my
research. Nita, my wife who always stood by my side in all difficult times of my life and
provided support and encouragement to complete research work.
I would like to thank all staff members of Electronics and Communication
Engineering department of my institute who have supported by providing me extra time in
the institute and guided me for research work.

Tusharkumar Pravinchandra Dave

Research Scholar, Gujarat Technological University

xiii
 Table of Contents
Abstract ................................................................................................................................ xi

Acknowledgment............................................................................................................... xiii

Table of Contents ............................................................................................................... xiv

Abbreviations .................................................................................................................... xvii

List of Symbols .................................................................................................................. xix

List of Figures ..................................................................................................................... xx

List of Tables.................................................................................................................... xxiv

Ch apt er 1. Introduction .............................................................................................. 1

1.1 Overview ................................................................................................................... 1

1.1.1 Evolution of LTE Communication ...................................................................... 1

1.1.2 LTE Bands ......................................................................................................... 3

1.1.3 Theoretical Background of LTE Antenna ........................................................... 5

1.1.4 Requirements and Challenges of LTE Antenna ................................................... 6

1.2 Theoretical Background of Microwave Structure ....................................................... 6

1.2.1 Microstrip Transmission Line ............................................................................. 6

1.3 Antenna Theory and Performance Parameters .......................................................... 12

1.4 Microstrip Patch Antenna Configuration and Related Formulations ......................... 15

1.5 Metamaterials .......................................................................................................... 17

1.5.1 History of Metamaterials .................................................................................. 17

1.5.2 Theoretical Background of Metamaterial .......................................................... 18

1.5.3 Material Classification and Their Properties ..................................................... 19

1.5.4 Transmission Line Analysis of Metamaterial .................................................... 23

1.5.5 Analysis of Metamaterials ................................................................................ 27

xiv
 1.5.6 Retrieval Methods for Material Medium Parameters ......................................... 29

1.6 Thesis Organization ................................................................................................. 32

Ch apt er 2. Literature Survey ................................................................................... 34

2.1 Antenna Design and Techniques for LTE Bands ...................................................... 34

2.1.1 Dielectric Resonator Antenna ........................................................................... 34

2.1.2 MIMO Antenna ................................................................................................ 37

2.1.3 Frequency Reconfigurable Antenna .................................................................. 40

2.1.4 Metamaterial Based Antenna ............................................................................ 44

2.2 Research Gap and Motivation .................................................................................. 48

2.3 Research Objectives ................................................................................................. 50

2.4 Research Methodology............................................................................................. 52

Ch apt er 3. Development of Metamaterial-Based Single Band and Dual Band

Microstrip Antenna ........................................................................................................... 53

3.1 Single Band Antenna Design and Implementation .................................................... 53

3.1.1 Fabrication and Measurement Setup of CSRR Loaded Microstrip Patch

Antenna ....................................................................................................................... 55

3.1.2 Results and Discussion ..................................................................................... 56

3.2 Dual-Band Antenna Design and Implementation ...................................................... 59

3.2.1 Fabrication of Connected Ring Resonator loaded Microstrip Patch Antenna ..... 60

3.2.2 Results and Discussion ..................................................................................... 61

Ch apt er 4. Development of Metamaterial Based Dual Band & Triple Band

Compact Microstrip Antenna ........................................................................................... 64

4.1 Antenna Design........................................................................................................ 64

4.1.1 SRR-Based Microstrip Patch Antenna .............................................................. 64

xv
 4.1.2 Thin Layer Dielectric Resonator Loaded Patch Antenna ................................... 66

4.2 Design of Metamaterial Unit-Cell Stacked TL-DR Antenna ..................................... 67

4.2.1 Unit-Cell Modelling and Analysis .................................................................... 67

4.2.2 Prototype Antenna Configuration ..................................................................... 70

4.3 Parametric Analysis ................................................................................................. 71

4.4 Antenna Fabrication and Measurement Setup ........................................................... 72

4.5 Discussion on Results .............................................................................................. 73

4.6 Performance Analysis TL-DR Metamaterial Loaded LTE Antenna .......................... 77

Ch apt er 5. Development of Metamaterial-Based Compact Frequency

Reconfigurable Microstrip Antenna ................................................................................. 79

5.1 Antenna Design........................................................................................................ 79

5.2 Antenna Fabrication & Experimental Setup.............................................................. 81

5.3 Results and Discussion ............................................................................................. 82

Ch apt er 6. Conclusions & Future Scope .................................................................. 88

6.1 Conclusions ............................................................................................................. 88

6.2 Future Scope ............................................................................................................ 89

References .......................................................................................................................... 90

Publications ........................................................................................................................ 95

xvi
 Abbreviations
1G First Generation
2G Second Generation
3G Third Generation
3GPP Third Generation Partnership Project
ABW Absolute Bandwidth
CDF Cumulative Distribution Function
CRLH Composite Right-Left Handed
CSRR Complementary Split Ring Resonator
CST-MWS Computer Simulation Tool - Microwave Studio Suite
DGS Defective Ground
DNG Double Negative
DPS Double Positive
DR Dielectric Resonator
DRA Dielectric Resonator Antenna
EBG Electromagnetic Bandgap
EM Electromagnetic Field
ENG Epsilon Negative
FDD Frequency Division Duplex
FDMA Frequency Division Multiple Access
FM Frequency Modulation
FSS Frequency Selective Surface
IFA Inverted F-Antenna
LTE Long Term Evolution
LT-TL Left Handed Transmission Line
MIMO Multiple Input Multiple Output
MNG Mue Negative
MSA Microstrip Antenna
MTM Metamaterial
NRW Nicolson Ross Weir
RF Radio Frequency
RH-TL Right Handed Transmission Line

xvii
RI Refractive Index
SNG Single Negative
SNR Signal to Noise Ratio
SRR Split Ring Resonator
TDD Time Division Duplex
TDMA Time Division Multiple Access
TL Transmission Line
TL-DRA Thin Layer Dielectric Resonator Antenna
UMTS Universal Mobile Telecommunication Systems
VNA Vector Network Analyser

xviii
 List of Symbols
C Channel bandwidth
εr Relative permittivity
εeff Effective permittivity
Z0 Characteristic impedance
β Phase constant
ω Angular frequency
vp Phase velocity
c Velocity of light
h Height of substrate
W Width of microstrip line
Γ Reflection coefficient
λ Operating wavelength
𝐸⃗ Electric field

𝐻⃗ Magnetic field
κ Wave vector
μeff Effective permeability
μr Relative permeability
n Refractive index
𝑆⃗ Poynting vector
f0 Resonant frequency
λg Guided wavelength
𝛼 Attenuation constant
𝛾 Propagation constant
ZL Load impedance
ZB Bloch impedance
T Transmission matric

xix
 List of Figures
Fig. 1.1 Illustration of the evolution of telecom systems[1] ................................................... 2
Fig. 1.2 Cross-sectional view of microstrip transmission lines with their corresponding
electric field (solid lines) and magnetic field (dashed lines) [7] .................................. 7
Fig. 1.3 Diagrams and equivalent circuit models of various microstrip discontinuities: (a)
step, (b) open-end, (c) gap and (d) bend [10-11] ....................................................... 12
Fig. 1.4 (a) Antenna as a transition device, (b) Equivalent circuit model of the antenna [12] 13
Fig. 1.5 Circuit showing transmission line and antenna model [12] ...................................... 13
Fig. 1.6 Layout of microstrip patch antenna. ........................................................................ 16
Fig. 1.7 Representation of wave propagation in MTM structure [17].................................... 17
Fig. 1.8 Classification of materials based on the value of effective medium parameters. ...... 20
Fig. 1.9(a) Periodic arrangement of thin-metal rods for effective negative permittivity, (b)
Array of SRR elements for effective negative permeability, (c) Realization of DNG
MTM based thin wires and SRR’s. ........................................................................... 20
Fig. 1.10 Effective permeability response of SRR structure [26] .......................................... 23
Fig. 1.11 (a) Ideal model of a uniform transmission line, (b) Equivalent circuit model of a
lossless infinitesimal size (z∆) CRLH line [27] ........................................................ 23
Fig. 1.12 (a) Equivalent circuit model of a periodic RH-TL having an infinitesimal
incremental length (z∆), and (b) Dual configuration of RH-TL [27]. ........................ 24
Fig. 1.13 (a) Schematic of the propagation mechanism of a generalized CRLH-TL unit-cell
showing dispersion and attenuation profile, (b) Transmission characteristics [30]. ... 25
Fig. 1.14 (a) Equivalent circuit model of a periodic LH-TL having an infinitesimal
incremental length (z∆), and (b) Dual configuration of LH-TL [27] ......................... 26
Fig. 1.15 Network topology for a general periodic unit-cell. (a) T-Type, (b) π-Type. ........... 27
Fig. 2.1 (a) Layout of MIMO antenna loaded with dielectric resonator, (b) Image of the
fabricated MIMO antenna [43] ................................................................................ 34
Fig. 2.2 Comparison of simulated and measured S11 and S21 of the dielectric resonator
loaded MIMO antenna [43] ...................................................................................... 35
Fig. 2.3 (a) Prototype hydrid DRA loaded triple band antenna, (b) Ground plane, (b) Top
radiating plane [44] .................................................................................................. 36
Fig. 2.4 Simulated and measured S11 performance of the hybrid DRA [44]......................... 36
Fig. 2.5 (a) Matching circuit of DRA (b) Image of DRA based antenna [78] ........................ 37
Fig. 2.6 Plot of Measured and simulated S11 of the DRA [78] .............................................. 37
Fig. 2.7 (a) Configuration of dual band filtering antenna element (b) Image of fabricated
antenna in MIMO arrangement [45] ......................................................................... 37
Fig. 2.8 Measured performance of filtering antenna for LTE band operation [45] ................ 38
Fig. 2.9 (a) Geometry of dual U-slot antenna (b) Fabricated prototype antenna [46]............. 39
Fig. 2.10 S-parameter response of the Antenna-1 and Antenna-2 [46] .................................. 39
Fig. 2.11 Proposed four-element SRR loaded MIMO antenna prototype design [47] ............ 40
Fig. 2.12 Measured S-parameter of the metamaterial SRR loaded MIMO antenna [47] ........ 40

xx
Fig. 2.13 (a) Schematic of the reconfigurable antenna with dielectric resonator (b) Image of
fabricated prototype antenna [48] ............................................................................. 41
Fig. 2.14 S11 response of the antenna for different switching states of PIN diodes [48]......... 42
Fig. 2.15 (a) Front view of the reconfigurable antenna (b) Ground plane view (c) Image of
the fabricated prototype antenna [49] ....................................................................... 42
Fig. 2.16 Simulated and measured S11 result of the proposed reconfigurable antenna in four
different state of the PIN diode bias [49] .................................................................. 43
Fig. 2.17 Single feed triple-band antenna [79]...................................................................... 44
Fig. 2.18 S11 result of antenna with and without using LC tuning filter S11 variation in the
antenna due to change in the state of EBG unit-cell structures [79] .......................... 44
Fig. 2.19 Front view of the fabricated metamaterial coupled antenna with DGS (b) Back
view of the antenna with partial corrugated ground plane [50].................................. 44
Fig. 2.20 Simulated and measured S11 result of the proposed metamaterial based antenna
[50] .......................................................................................................................... 45
Fig. 2.21 (a) Layout of pentagon-shaped patch with EBG unit-cell in the partial ground
plane (b) Image of the fabricated prototype antenna [51] .......................................... 46
Fig. 2.22 S11 variation in the antenna due to change in the state of EBG unit-cell structures . 46
Fig. 2.23 (a) Schematic of the unit-cell structure (b) Magnitude response of the effective
material medium parameters [80] ............................................................................. 47
Fig. 2.24 Image of metamaterial based antenna (a) Top view (b) Bottom view [80] ............. 47
Fig. 2.25 Simulated and measured returnloss plot of the antenna with and without MTM
loading [80] ............................................................................................................. 48
Fig. 3.1 Microstrip patch antenna design iterations for achieving strong fundamental
resonance using metamaterial complementary ring resonator structure.. ................... 54
Fig. 3.2 Layout of proposed metamaterial loaded CSRR microstrip patch antenna for LTE
band operation. ........................................................................................................ 54
Fig. 3.3 Image of the fabricated prototype antenna. (a) Front view, (b) Back view. .............. 56
Fig. 3.4 Antenna measurement setup.................................................................................... 56
Fig. 3.5 Comparison of simulated S11 of the antenna designs of iteration-1, iteration-2 and
iteration-3. ............................................................................................................... 57
Fig. 3.6 Comparison of simulated and measured S11 of CSRR loaded microstrip patch
antenna for LTE band............................................................................................... 58
Fig. 3.7 Simulated VSWR response of the propose CSRR loaded microstrip patch antenna. 58
Fig. 3.8 (a) 3D radiation pattern of the antenna, (b) Polar plot of the radiation pattern in E-
Plane ........................................................................................................................ 59
Fig. 3.9 Simulated surface current of CSRR loaded microstrip patch antenna....................... 59
Fig. 3.10 Layout of metamaterial connected ring resonator loaded microstrip patch
antenna. (a) Top view, (b) Ground plane. ................................................................. 60
Fig. 3.11 Image of the fabricated prototype dual band metamaterial loaded microstrip patch
antenna (a) Top view, (b) Bottom view. ................................................................... 61
Fig. 3.12 Simulated and measured S11 response of connected metamaterial ring resonator
loaded microstrip patch antenna.. ............................................................................. 61

xxi
Fig. 3.13 Measurement of S11 performed using Anritsu vector network analyser. ................. 62
Fig. 3.14 3D radition pattern of connected metamaterial ring resonator loaded microstrip
patch antenna. .......................................................................................................... 62
Fig. 3.15 Simulated antenna surface current distribution at (a) 1.66 GHz, (b) 3.475 GHz ..... 63
Fig. 4.1 Layout of the SRR-based antenna with the reduced ground plane. (a) Top view,
(b) Bottom view ....................................................................................................... 65
Fig. 4.2 Perspective view of the SRR-based antenna loaded with TL-DR. ............................ 66
Fig. 4.3 Layout of the proposed unit-cell (UL = 12 mm, GL = U3 = 2 mm, U1 = a = 1 mm,
U2 = 1.8 mm, U4 = 10 mm, gt = 1.5 mm) ................................................................ 67
Fig. 4.4 Simulated scattering response of the unit-cell with inset image of boundary
condition .................................................................................................................. 68
Fig. 4.5 Plot showing the effect of gt on unit-cell (a) S11 and (b) S21. ................................... 69
Fig. 4.6 Retrieved plots of the unit-cell characteristic parameters. (a) Permeability, (b)
Permittivity, (c) Wave impedance (z), and (d) Refractive index (R.I). ...................... 69
Fig. 4.7 Perspective view of SRR-based antenna loaded with a stack of TL-DR and
metamaterial unit-cell. .............................................................................................. 70
Fig. 4.8 Plot of parametric variation on geometrical dimensions. (a) P L1, (b) Pw1, (c) S1, (d)
S2. ............................................................................................................................ 71
Fig. 4.9 Image of the fabricated antenna. (a) SRR-based antenna without TL-DR and unit-
cell. (b) SRR-based antenna loaded with a stack of TL-DR and metamaterial unit-
cell, (c) Bottom ground plane.. ................................................................................. 72
Fig. 4.10 Image of the antenna measurement setup. ............................................................. 73
Fig. 4.11 Comparison of the antenna reflection coefficient after loading TL-DR and unit-
cell. .......................................................................................................................... 74
Fig. 4.12 Comparison of the measured and simulated plot of S11. (a) Bare antenna, (b) TL-
DR loaded antenna, (c) Unit-cell loaded on TL-DRA. .............................................. 74
Fig. 4.13 Simulated current distributions of the antenna at different resonant frequencies.
(a) 2.031 GHz, (b) 2.461 GHz, (c) 2.145 GHz, (d) 2.247 GHz, (e) 2.304 GHz. ......... 75
Fig. 4.14 Simulated radiation pattern in E-plane and H-plane. (a) 2.031 GHz, (b) 2.29 GHz,
(c) 2.304 GHz. ......................................................................................................... 76
Fig. 4.15 Comparison of simulated and measured wideband gain of the antenna.. .............. 76
Fig. 4.16 Plot of the antenna radiation efficiency ................................................................ 77
Fig. 5.1 Antenna Layout. (a) Top view of the radiating plane, (b) Bottom view of the
ground plane; where L = W = 45, h = 1.55, f1 = 3.318, f2 =10, C1 = 7, C2 = 8, Sw =
5.5, Sl = 11, g = 4, u1 = 1.5, r1 = 7, r2 = 5.9, r3 = 10, r4 = 14, M1 = 25, M2 = 40, M3 =
M4 = 10 S0 = S1 = 5, P = 17 (all dimension are in mm) ............................................. 80
Fig. 5.2 Image of the fabricated antenna ............................................................................. 81
Fig. 5.3 (a) Experimental setup for antenna measurement, (b) RF/DC isolation network
configuration (Rs = 5 Ω & Ls = 4 nH are series resistance and inductance of PIN
diode in forward bias respectively) ........................................................................... 82
Fig. 5.4 Material response of SRR. (a) Plot of permittivity, (b) Plot of permeability ............ 83

xxii
Fig. 5.5 Comparative plot of antenna reflection coefficient (S 11) for diode ON-OFF
Conditions. (a) D1-D2 OFF, (b) D1-ON, (c) D2-ON, (d) D1-D2-OFF...................... 83
Fig. 5.6 Simulated surface current distribution of the antenna at (a) 0.96 GHz, (b) 2.63
GHz, (c) 3.22 GHz ................................................................................................... 85
Fig. 5.7 Simulated E-plane radiation pattern of the antenna at (a) 0.96 GHz, (b) 2.63 GHz,
(c) 3.22 GHz. ........................................................................................................... 85
Fig. 5.8 Measured plot of antenna gain in comparison to the simulated. ............................... 86

xxiii
 List of Tables

Table 1.1 Details of frequency bands of FDD LTE [4] ........................................................... 3

Table 1.2 Details of frequency bands of TDD LTE [4]........................................................... 4
Table 1.3 Summary of the transmission line properties. ....................................................... 26
Table 3.1 Illustration of antenna dimension obtained from Iteration-3.................................. 55
Table 3.2 Dimensions of the metamaterial loaded dual-band microstrip patch antenna. ........ 60
Table 4.1 Illustration of the antenna layout dimensions. ....................................................... 66
Table 4.2 List of the antenna resonant frequencies. .............................................................. 75
Table 4.3 Comparison of the proposed TL-DRA with other reported antenna design
techniques for LTE application. ............................................................................... 77
Table 5.1 Measured and simulated resonant bands of the proposed antenna ......................... 84
Table 5.2 Performance comparison of the proposed antenna with other reported LTE band
antennas. .................................................................................................................. 87

xxiv
 Overview

CHAPTER - 1

Introduction

1.1 Overview

The planar antenna has always been a necessary element in the evolution of
sophisticated wireless communication systems, and it provides an effective means for free-
space communication through electromagnetic waves. Recent advancements in the
wireless standards have increased the demand for multiband antennas that can operate at
multiple frequencies with desired bandwidth. The need for higher bandwidth or multiple
narrowband resonances in the antenna carter essential functions such as spectrum sensing,
smart allocation of data on multiple frequency channels and selective communication for
various applications related to the industrial, defence, and biomedical.

1.1.1 Evolution of LTE Communication Technology

The Long-Term Evolution (LTE) technology is a huge step in communication

technology. The continuous expansion of LTE into the upcoming systems is due to the
growing demand for high-speed packet and need for large channel capacity. The evolution
of LTE is illustrated in Fig. 1.1 [1]. The advancement in wireless communication
technology can be categorized into different generations in which each preceding
generation has overcome the limitations of the previous. The first generation (1G) mobile
communication devices consisted of analog systems. The 1G technology supported voice-
only services and no roaming. This technology featured frequency modulation (FM),
frequency division duplexing (FDD), and Frequency division multiple access (FDMA)
methods [2].

1
 Introduction

FIGURE 1.1 Illustration of the evolution of telecom systems [1].

Mainly monopole type antennas were used for single-band communication. First-
generation wireless systems were having several limitations such as low capacity,
inconsistent voice quality, cross-talk between users, and bulky equipment [3].
The 2G systems were mainly dependent on digital signal modulation to provide a much
higher bandwidth capacity as well as digital encryption [4]. The combination of consistent
voice quality along with relatively small-sized devices attracted a wide variety of
applications. Time-division multiple access methods (TDMA) was one of the prominent
channel access methods in the 2G wireless systems. GSM (Global System for Mobile) and
PDC (Personal Digital Cellular) were some of the 2G standards based on this time-division
method [3].
Third generation (3G) mobile technology was based on standards that comply with IMT-
2000. Universal Mobile Telecommunication System (UMTS) is one of the examples of 3G
technology. This was defined by the Third Generation Partnership Project (3GPP) based
on updated GSM specifications [3]. Some other technologies that comply with the IMT-
2000 or 3G standards are CDMA2000, WiMAX, and EDGE. Third-generation mobile
devices provided much higher data rates than previous technologies. Originally peak data
rates of 200 Kbits/s were observed; however, later releases of 3G such as 3.5G and 3.75G
featured data rates capable of 10 Mbits/s.
One of the fourth-generation communication technologies is 3GPP’s Long Term
Evolution. Similar to the 3GPP, which used updated GSM standards to define UMTS,
3GPP was used to update UMTS and called it LTE [5]. A key driver in LTE
standardization was the spectrum flexibility. The Frequency spectrum is a very limited
resource and efficient use of spectrum has become a focus in the telecommunication

2
 Overview

industry. LTE mobile communication devices must support networks across a wide range
of allocated frequencies. In LTE, new high-speed access methods have been defined for
the smart mobile communication systems. Flexibility and interoperability with current
technology are additionally combined with the following features like;
 High data rates of 100 Mbps for downlink and 50 Mbps for the uplink.
 Improved frequency spectral efficiency.
 The system transceiver operation takes place at the same time using full-duplex
methodology.

1.1.2 LTE Frequency Bands

LTE technology offers not only higher performance but also reduced capital and operating
costs. The primary goal of LTE was to evolve within the existing infrastructure. Thus,
several existing frequency bands of different wireless standards were adopted for the
allocation of the LTE frequency band. The LTE spectrum can be split into two main
categories based on the duplexing methods i.e., Frequency Division Duplexing (FDD) and
Time Division Duplexing (TDD). In Table 1.1 and Table 1.2, the allocation of uplink and
downlink frequency bands are mentioned.

TABLE 1.1 Details of frequency bands of FDD LTE [4]

Band Number Uplink (MHz) Downlink (MHz) Main Regions

1 1920-1980 2110-2170 All
2 1850-1910 1930-1990 NA
3 1710-1785 1805-1880 All
4 1710-1755 2110-2155 NA
5 824-849 869-894 NA
6 830-840 875-885 APAC
7 2500-2570 2620-2690 EMEA
8 880-915 925-960 All
9 1749.9-1784.9 1844.9-1879.9 APAC
10 1710-1770 2110-2170 NA
11 1427.9-1447.9 1475.9-1495.9 Japan
12 699-716 729-746 NA

3
 Introduction

Band Number Uplink (MHz) Downlink (MHz) Main Regions

13 777-787 746-756 NA
14 788-798 758-768 NA
17 704-716 734-746 NA
18 815-830 860-875 Japan
19 830-845 875-890 Japan
20 832-862 791-821 EMEA
21 1447.9-1462.9 1495.9-1510.9 Japan
22 3410-3490 3510-3590 EMEA
23 2000-2020 2180-2200 NA
24 1626.5-1660.5 1525-1559 NA
25 1850-1915 1930-1995 NA
26 814-849 859-894 NA
27 807-824 852-869 NA
28 703-748 758-803 APAC
29 - 717-728 NA
30 2305-2315 2350-2360 NA
31 452.5-457.5 462.5-467.5 CLA
32 - 1452-1496 EMEA
Asia and Pacific (APAC); Europe, Middle East, and Africa (EMEA); North
American (NA); Central Latin America (CLA)

TABLE 1.2 Details of frequency bands of TDD LTE [4]

Band Number Spectrum (MHz) Main Regions

33 1900-1920 EMEA
34 2010-2025 EMEA
35 1850-1910 NA
36 1930-1990 NA
37 1910-1930 NA
38 2570-2620 EMEA
39 1880-1920 China
40 2300-2400 China
41 2496-2690 All
42 3400-3600 -
43 3600-3800 -
44 703-803 APAC
Asia and Pacific (APAC); Europe, Middle East, and
Africa (EMEA); North American (NA); Central Latin
America (CLA)

4
 Overview

The full spectrum of LTE mainly consists of different regional and universal bands. It can
utilize a variety of different frequency bands from various regions and areas to provide
benefits to the end-user and service provider of the LTE network for better signal
connectivity, bandwidth, and quality of service. Most of the LTE bands are allocated to a
specific mobile communication device that can operate for more diverse reception
coverage.

1.1.3 Theoretical Background of LTE Band Antenna

Antenna diversity for LTE band operation is commonly realized by a Multiple Input
Multiple Output (MIMO) configuration, where two more antennas are integrated on a
common substrate and electronically biased for the desired frequency of operation. The
antennas in the mobile phone need diversity, so they are required to differ in the way they
send the signal to the base station. But the diversity performance can be affected by many
factors, such as the antenna radiation pattern and the antenna positioning. A MIMO system
also helps to overcome multipath fading, which is a major performance impairment of the
wireless communication device. In a MIMO system, there are N number of transmitting
antennas and M number of receiving antennas. The wireless channel can be written as an
M×N matrix with random independent elements; expressed as H. Now the capacity of the
channel without any prior transmit information becomes [6]:

𝑆𝑁𝑅
𝐶 = 𝑙𝑜𝑔 det 𝐼 + 𝐻𝐻 ∗ (1.1)
𝑁

where SNR now represents the average signal to noise ratio of the receiving antennas.
Since the capacity is a random quantity, it is useful to calculate the cumulative distribution
function (CDF) rather than any instantaneous values. Considering the average capacity Ca
when M=N the previous expression can be simplified to:

𝑆𝑁𝑅
𝐶 ≈ 𝑁𝑙𝑜𝑔 1+ (1.2)
𝑁

The capacity can be increased by deploying more than one antenna element since the
MIMO channel is subsequently split into N virtual parallel channels. MIMO systems

5
 Introduction

utilize what was traditionally a major impairment of wireless communications systems,

multipath fading, to their advantage. Multipath fading is the phenomenon that occurs when
a receiver sees multiple copies of a transmitted signal. Since the transmitted signal reflects
and transmits around and through various obstacles before reaching the receiver, the final
received signal is a superposition of the transmitted signal that may contain components
out of phase with one another causing significant attenuation in the total received signal.

1.1.4 Requirements and Challenges of LTE Band Antenna

There exist several essential requirements and challenges for the implementation of LTE
antenna in a communications device, and these are;
 Multiple resonant frequencies, which essentially requires multiband communication
and a wide range of frequency diversity.
 Omnidirectional radiation pattern without any performance degradation due to the
presence of other radiation sources.
 Mutual coupling reduction between two or more antenna elements.
 Selective polarization and radiation pattern of an electromagnetic wave from the
antenna for point-to-point or broad range communications.
 Another important design goal is the size miniaturization in comparison to the
existing antenna topologies like PIFA, MIMO based antenna, and frequency
reconfigurable antenna based on active switching between various stubs and
resonators.

1.2 Theoretical Background of Microwave Structure

1.2.1 Microstrip Transmission Line

The microstrip line is one of the most popular types of the planar transmission line. It
allows integration with an active and passive component in a microwave device and allows
easy fabrication using standard photolithography process. Microstrip transmission lines
comprises of a conductor printed on the top of a dielectric substrate with a ground. The
cross-sectional view of a microstrip transmission line geometry is shown in Fig. 1.2 [7].

6
 Theoretical Background of Microwave Structure

FIGURE 1.2 Cross-sectional view of microstrip transmission lines with their corresponding electric
field (solid lines) and magnetic field (dashed lines) [7].

The properties of the conductor in conjunction with the dielectric substrate define the
signal transmission characteristics of the transmission line. In microstrip lines, the
electromagnetic fields exist not only within the dielectric substrate, but they also propagate
into the air above. The relative permittivity (𝜀 ) of the substrate is always higher than
unity. This causes the fields in the air to propagate faster than the fields within the
dielectric substrate. Therefore, microstrip transmission lines are unable to support a pure
transverse electromagnetic wave. Comparably weak longitudinal field elements allow a
quasi-static approximation to be applied in characterizing the microstrip transmission lines
up to a few GHz [8]. This technique considers the propagation of a pure transverse
electromagnetic mode which, in turn, simplifies the related calculations. Once the
capacitance per unit length with and without the dielectric - Cd and C0 respectively - are
known, the effective dielectric constant, 𝜀 eff, characteristic impedance in free space (Z0),
the phase constant (𝛽), angular frequency (𝜔), and the phase velocity (vp), can be
calculated using Eq. (1.3)-(1.6).

𝐶
𝜀 = (1.3)
𝐶

1
𝑍 = (1.4)
𝑐 𝐶 𝐶

𝜔 𝐶
𝛽= (1.5)
𝑐 𝐶

𝐶
𝑣 =𝑐 (1.6)
𝐶

7
 Introduction

1.2.1.1 Analysis of Microstrip Line Structure

For microstrip line-fed structure, as shown in Fig. 1.2, the closed-form expressions for
the effective dielectric constant (𝜀 eff), and the characteristic impedance (Z0) are given in
Eq. (1.7)-(1.9) [8]. In these equations, W - is the width of the microstrip transmission line,
h - is the height of the substrate, and n - is the free-space wave impedance and has a value
of 377 Ω. For narrow microstrip transmission lines, i.e., when W/h <1,

𝜀 +1 𝜀 −1 12ℎ 𝑊
𝜀 = + × 1+ + 0.04 1 − (1.7)
2 2 𝑊 ℎ

On the other hand, for wide microstrip transmission lines, i.e., when W/h >1,

𝜀 +1 𝜀 −1 12ℎ
𝜀 = + × 1+ (1.8)
2 2 𝑊

𝑛 𝑊 𝑊
𝑍 = × + 1.393 + 0.677 ln + 1.444 (1.9)
𝜀 ℎ ℎ

1.2.1.2 Discontinuities in Microstrip Transmission Line

Discontinuities in microstrip transmission lines are generally experienced during the

design of practical antennas. The typical discontinuities include changes in the widths,
open-ends, gaps, bends, and junctions. These are represented in Fig. 1.3 along with their
respective equivalent circuits. Discontinuities in the microstrip line generate parasitic
reactance which affects the electromagnetic field distributions. An equivalent capacitance
can characterize the transformed electric field distribution, and an equivalent inductance
can characterize the transformed magnetic field distribution.
The equivalent circuit model of the region containing the discontinuity can be modeled
accurately by deducing the effective dimensions, and these, in turn, are taken into account
during the antenna design with full-wave electromagnetic simulations.

8
 Theoretical Background of Microwave Structure

A. Step Width

The discontinuity is named as steps in width and it is formed on the junction of two
transmission lines having different width. The equivalent circuit model of this
discontinuity consists of shunt capacitance (C) in the plane of the junction and two series
inductances (L1 and L2) representing the two transmission lines on either side of the shunt
capacitance. This is shown in Fig. 1.3 (a). For a symmetrical step, approximate values for
the capacitance, C, in pF and the inductances, L1 and L2, in nH can be calculated by using
the formulas from [9-11] as given in Eq. (1.10)-(1.12) below.

𝑊
𝜀 𝑊 𝜀 + 0.3 + 0.264
𝐶 = (0.00137ℎ) 1− ℎ
𝑍 𝑊 𝜀 − 0.258 𝑊 (1.10)
+ 0.8
ℎ

𝐿 (1.11)
𝐿 =𝐿
𝐿 +𝐿

𝐿 (1.12)
𝐿 =𝐿
𝐿 +𝐿

𝑍 𝜀
𝐿 = (0.000987ℎ) × 1 −
𝑍 𝜀

B. Open-Ends

The discontinuity of open ends is formed when a transmission line ends in an open
circuit. The equivalent circuit model of this discontinuity consists of an open-circuit
capacitance (Cp). This is represented in Fig. 1.3(b) [9]. The open-circuit capacitance
causes fringing effects [10] and, hence, there is an apparent increase by a margin of ∆l in
the length of the transmission line. The approximate values for the open-circuit
capacitance (Cp) and increase in the length (∆l) can be calculated by using the formulas
[10] as given in Eq. (1.13) and Eq. (2.14) respectively. For W/h ≥ 0.2 and 2 ≤ 𝜀 ≤50,
these equations give results that are within 4% of the numerical results.

9
 Introduction

𝑊
𝜀 + 0.3 + 0.264
∆𝑙 = (0.412ℎ) × × ℎ
𝜀 − 0.258 𝑊 (1.13)
+ 0.8
ℎ

𝜀 (1.14)
𝐶 = ∆𝑙 ×
𝑐𝑍

where 𝜀 r is the relative permittivity of the substrate, h is the height of the dielectric
substrate, 𝜀 eff is the effective dielectric constant, W is the width of the transmission line, c
is the speed of light and Z0 is the characteristic impedance.

C. Gap between Microstrip Lines

This type of discontinuity is formed when there is a gap between the two microstrip
lines. The equivalent circuit model of this discontinuity can be denoted by a pi-network
consisting of two capacitances, i.e., Cp and Cg. The circuit model is represented in Fig.
1.3(c). If the two microstrip transmission lines are identical, then a plane of circuit
symmetry exists, and it can be analysed using even and odd mode analysis. In the even
mode, the incident waves along the microstrip lines are symmetrical and equal, i.e., equal
in magnitude and phase, and the odd mode, they are anti-symmetrical and opposite, i.e.,
equal in magnitude, but 180° out of phase. The even and odd mode is expressed by Ce and
Co, respectively. The capacitances of the equivalent circuit model, Cp, and Cg, can be
determined once the values for the Ce and Co are known. These can be calculated using the
formulas from [10-11] as given in Eq. (1.15)-(1.18).

𝐶 = 0.5𝐶 (1.15)

𝐶 = 0.5𝐶 − 0.25𝐶 (1.16)

where, Ce and C0 can be calculated as,

𝜀 . 𝑠
𝐶 = (12𝑊) × × × (𝑒 ) (1.17)
9.6 𝑊

𝜀 . 𝑠
𝐶 = (𝑊) × × × (𝑒 ) (1.18)
9.6 𝑊

10
 Theoretical Background of Microwave Structure

where, me, mo, and ko can be calculated for the different conditions of s/W factor. When
0.1 ≤ ≤ 1.0,

𝑊 𝑊
𝑚𝑜 = × 0.619 log − 0.3853 (1.19)
ℎ ℎ

𝑊
𝑘𝑜 = 4.26 − 1.453 log (1.20)
ℎ

Moreover, when0.1 ≤ ≤ 0.3,

𝑚𝑒 = 0.8675 (1.21)

.
𝑊
𝑘𝑒 = 2.043 × (1.22)
ℎ

The type of discontinuity is formed when a transmission line changes by an angle of 90 0.

The equivalent circuit model of this discontinuity can be denoted by a T-network
consisting of shunt capacitance, C, and two series inductances, L, on either side of the
shunt capacitance. The representation of the bend and its circuit model is shown in Fig.
1.3(d). The approximate values for the shunt capacitance, C-pF and the series inductances,
L-nH can be calculated;

When W/h <1,

𝑊
(14𝜀 + 12.5) − 18𝜀 + 2.25 0.02𝜀
𝐶= ℎ + (1.23)
𝑊 𝑊
ℎ ℎ

When W/h ≥1
𝑊
𝐶 = 𝑊[ (9.5𝜀 + 1.25) + 5.2𝜀 + 7 (1.24)
ℎ

𝑊
𝐿 = 100ℎ × 4 − 4.21 (1.25)
ℎ

11
 Introduction

(a)

(b)

(c)

(d)

FIGURE 1.3 Diagrams and equivalent circuit models of various microstrip discontinuities: (a) step, (b)
open-end, (c) gap and (d) bend [10-11].

1.3 Antenna Theory and Performance Parameters

The antenna is defined as a transmitting or receiving system that is designed to radiate

or to receive electromagnetic waves. The above definition signifies that there are two types
of antennas: transmitting antennas and receiving antennas. Transmitting antennas are the
antennas that take the signals from a transmission line, convert them into electromagnetic
waves, and then broadcast them into free space. The receiving antennas operate in reverse,
i.e., they collect the electromagnetic waves from free space, convert them into signals and
then put them back in the transmission line. An example of a simple antenna is shown in
Fig. 1.4(a) [12]. The equivalent circuit model of antenna consists of a voltage source, a
current, impedance, and an RLC network. An illustration of the equivalent circuit model is
shown in Fig. 1.4 (b).

12
 Antenna Theory and Performance Parameters

(a) (b)

FIGURE 1.4 (a) Antenna as a transition device, (b) Equivalent circuit model of the antenna [12]

There are several standard metrics used to characterize the performance of an antenna. In a
single port system, the scattering parameter S11 i.e. reflection coefficient is used. The
reflection coefficient is a representation of the ratio of reflected power to supplied incident
power. Alternatively, the measurement of accepted power to incident power is known as
return loss. The reflection coefficient and return loss represent the same metric; they are
equivalent in magnitude but opposite in sign. This can be understood using the
transmission line model, as shown in Fig. 1.5.

FIGURE 1.5 Circuit showing transmission line and antenna model [12].

The smaller the value of the reflection coefficient, the more power the antenna is
accepting. A significant reflection coefficient corresponds to poor impedance matching at
the input terminal as seen from Eq. (1.26). In most cases, coaxial and microstrip
transmission lines feeding the antenna have an impedance of 50 Ω. To obtain maximum
power transfer between the antenna and transmission line, the characteristic impedance of
both should match. Both the antenna input impedance and return loss can be valuable and
relate how efficiently the antenna accepts the supplied power. It is important to note that

13
 Introduction

since the impedance is a function of frequency, the return loss also varies with frequency;
thus, these values are usually plotted over a range of frequencies.

𝑃 𝑍 −𝑍
Reflection Coefficient 𝑆 (𝑑𝐵) = 10𝑙𝑜𝑔 = 20 log (1.26)
𝑃 𝑍 +𝑍

𝑃 𝑍 −𝑍
Return Loss 𝑅𝐿(𝑑𝐵) = 10𝑙𝑜𝑔 = −20 log (1.27)
𝑃 𝑍 +𝑍

The radiation pattern is another metric used to characterize the behaviour of an antenna. A
radiation pattern is a three-dimensional representation of the normalized power measured
across an imaginary spherical surface in the antennas far field. The far-field is defined as
the region surrounding an antenna where the angular field distribution is independent of
the distance away from the antenna [10]. The far-field can be described as,

𝑅 > 2𝐷 /𝜆 (1.28)

where R is the distance away from the antenna, D is the largest dimension across the
antenna structure, and λ is the operating wavelength. Another quantitative description of
an antenna is the radiation efficiency. The radiation efficiency is a ratio of the radiated
power to input power [10].

𝑃 𝑅 (1.29)
𝑒 = =
𝑃 𝑅 +𝑅

Where Rr is the radiation resistance, and RL is the load resistance, i.e., antenna resistance.
Radiation efficiency is an important parameter to consider because even when an antenna
shows good return loss and is accepting power, that does not necessarily correlate to the
amount of power radiated. The next parameter is antenna gain, which is a representation of
how much energy is concentrated in a specific direction with respect to the average energy
radiated in all directions [10].

𝑈(𝜃, 𝜙) (1.30)
𝐺(𝜃, 𝜙) =
𝑃 /4𝜋

14
 Microstrip Patch Antenna Configuration and Related Formulas

where 𝑈(𝜃, 𝜙) represents the radiation intensity, and Pin is the net power accepted by the
antenna. The gain is inherently a function of θ and φ as shown in the formula, however
often it is beneficial to know the maximum value, which is known as peak gain. The peak
gain is useful because it can give a quick indication of an antenna efficiency and
directivity when plotted versus frequency. Another critical parameter is the bandwidth of
the antenna. In antennas, the bandwidth can be taken as the specific range of frequencies
on either side of the center/resonant frequency, within which an antenna performance
characteristic is within an acceptable value of those at the center/resonant frequency. The
bandwidth of antenna can be given in terms of either the absolute bandwidth or the
fractional bandwidth. The absolute bandwidth, ABW, is expressed as a difference between
the upper limit and the lower limit of the bandwidth. It is given as

𝐴𝐵𝑊 = 𝑓 − 𝑓 (1.31)

where fh and fl are the upper limit and the lower limit respectively of the frequency range at
a return loss of 10 dB. Whereas, the fractional bandwidth is a percentage function of the
upper limit of the bandwidth, the lower limit of the bandwidth, and the center frequency of
an antenna.

𝑓 −𝑓 (1.32)
𝐹𝐵𝑊 =
𝑓𝑓

1.4 Microstrip Patch Antenna Configuration and Related Formulas

The microstrip patch antenna consists of a metallic patch on the top side of a dielectric
substrate and ground on the bottom side. A substrate with a dielectric constant in the range
of 2.2 < 𝜀 <12 is used to create a loosely bounded EM field for efficient radiation
mechanism. The patch can be rectangular, circular, elliptical, triangular or any other shape.

For a rectangular patch, the length L of the element is usually <𝐿< . The basic

configuration of the microstrip patch antenna is shown in Fig. 1.6. The width W of the
rectangular patch is given in terms of the speed of light c and the frequency f0 [10] by,

15
 Introduction

𝑐 (1.33)
𝑊=
𝜀 +1
2𝑓 2

FIGURE 1.6 Layout of a microstrip patch antenna

The effective length of the rectangular patch can be calculated as

𝑐 (1.34)
𝐿= − 2Δ𝐿
2𝑓 𝜀

The effect of fringing fields can be described by Δ𝐿

𝑊 (1.35)
𝜀 +3 + 0.264
Δ𝐿 = 0.421ℎ ℎ
𝑊
𝜀 − 0.258 + 0.8
ℎ

Where,

(1.36)
𝜀 +1 𝜀 −1 ℎ
𝜀 = + 1 + 12
2 2 𝑊

In the above-discussed formulations, h is the height of the dielectric substrate, 𝜀 is the

relative permittivity of the substrate and 𝜀 is the effective dielectric constant.

16
 Metamaterials

1.5 Metamaterials

The Metamaterial (MTM) is a sub-wavelength structure composed of planar or three-

dimensional arrangement of metal and dielectric. In MTM structure, the traveling
electromagnetic wave experiences a backward wave propagation or evanescent modes
over a range of frequency bands [13]. The metamaterials are sub-wavelength substrates.

(a) (b)
FIGURE 1.7 Representation of wave propagation in MTM structure [14]

In MTM, the random but yet the most suitable combination of metal and dielectrics
results in negative response permittivity (ɛ), permeability (μ), and refractive index (n). In
these materials, the electric field (𝐸⃗ ), magnetic-field (𝐻⃗ ) and wave vector (κ) components
of the plane wave traveling enters into left-handed propagation with pointing vector (𝑆⃗) in
the opposite direction. As a result, phase velocity (vp) gets reversed as shown in Fig. 1.7.

1.5.1 History of Metamaterials

The first theoretical explanation of negative indexed material was proposed by V. G.

Veselago in 1968 [13]. In connection to this, J. B. Pendry et al. [14] experimentally
developed the metamaterial as Split-Ring Resonator (SRR) in 1999 which resulted in a
negative response of the effective negative permeability (-µeff). Next, in the early year of
2000, Smith and co-author practically demonstrated periodic SRR structures loaded with
thin wire rods to provide negative effective permittivity (-ɛeff) [15]. After that, many
configurations of the metamaterial were implemented which includes broadside coupled
SRR having anisotropic behavior [16], electric LC-resonators, and mesh arrangement of

17
 Introduction

different shaped MTM’s [17-19]. Single Negative (SNG) or Epsilon Negative (ENG) is
another class of MTM that has the potential capability to go beyond the current limitation
of double negative MTM structure in context to the sensitivity of metamaterial-based
sensor devices. Generally, negative permittivity exists naturally in the plasmonic material
such as noble nanoparticles of gold and silver, and the polar dielectrics below the plasma
frequency. At this frequency, the incident waves are attenuated, and no propagation
occurs. Thus, a stopband response is the characteristic behavior of ENG materials. Another
essential phenomenon occurs when the electromagnetic wave impinges on the ENG
structure; all the fields get localized into the structure. A net change in the field
distribution then becomes more significant when it interacts with a homogenous or
inhomogeneous dielectric composite. In this scenario, the phase shift near the sample area
into the ENG structure is more, and the effective permittivity becomes dependent on the
sample.

1.5.2 Theoretical Background of Metamaterials

The electromagnetic metamaterials are artificially engineered homogeneous structures

whose properties do not exist in nature. The MTM’s are designed to have unit-cell size
(say P) smaller than the guided wavelength (λg), i.e. P << λg or at least P < λg/4 [20].
These conditions assure that when a wave is traveling through the MTM structure,
refractive phenomena dominates over scattering/diffraction phenomena. The theory of
double-negative parameters studied by V. G. Veselago derived an vital implication that
propagation of electromagnetic wave in a complex media does not affect the dispersion
equation. The dispersion equation for the effective isotropic medium can be expressed as
[21];

2
2  n2 (1.37)
c2

where κ indicates the propagation constant, c is the velocity of light, n denotes the
refractive index, and ω is the angular frequency in rad/sec. The refractive index of the
medium which is written as 𝑛 = √𝜀 𝜇 explains that in a lossless medium, a simultaneous
change in the sign of ɛr and µr of the composite structure doesn’t alter the Eq. (1.37). But
for SNG medium, the opposite sign of ɛr and µr led to the imaginary value of κ and n and

18
 Metamaterials

inhibition of the incident wave vector. Based on these observations, materials are
classified depending upon the effective nature of permittivity and permeability.

1.5.3 Classification of Metamaterial and Their Properties

Considering the different possible relation of permittivity and permittivity, the

medium or composite materials are classified into four categories as Double Positive
(DPS) material (+ɛ, +µ) into the first quadrant, Epsilon Negative (ENG) material (-ɛ, +µ)
in the second quadrant, Double Negative (DNG) material (-ɛ, -µ) entering into third
quadrant, and Mue Negative (MNG) material (+ɛ, -µ) in the fourth quadrant. If the
medium is source free, isotropic, homogeneous and linear, the Maxwell equations can be
transformed to phasor representation as [22];

 E   j H (1.38)

 H  j E (1.39)

Now, Eq. (1.38) and Eq. (1.39) can be expressed for the monochromatic plane wave as

 (1.40)
kE  H
c


kH  E (1.41)
c

Eq. (1.40) and Eq. (1.41) shows that positive value of ε and μ forms wave vector and its
transverse 𝐸⃗ and 𝐻⃗ Field towards right-hand. On the other hand, the simultaneous negative
nature of ε and μ results in the left-handed propagation of the vector quantities. Therefore,
the Poynting vector (𝑆⃗ = 𝐸⃗ × 𝐻⃗ ) points opposite to the direction of k. So, for left-handed
material, the energy propagates in the reverse direction of the wave vector. Another
important observation was proposed that ε and μ cannot become simultaneously negative
without energy (W) being negative. Therefore, the dispersion equation relating total energy
as a sum of electric energy (ɛE2) and magnetic energy (µH2) must be modified as [22];

19
 Introduction

( ( )) 2 (  ( )) 2
W E  H (1.42)
 

The dispersion equation represented in Eq. (1.42) describes the distribution of energy in
the propagating EM wave in the form of the electric and magnetic field. Eq. (1.42) is in
decomposed form for material medium parameter i.e. ε and μ, which varies with respect to
frequency and depicts the total energy content as electric (εE2) and magnetic energy (μH2).
The material classification is shown in Fig. 1.8.

FIGURE 1.8 Classification of materials based on the value of effective medium parameters.

(a) (b)

(c)
FIGURE 1.9 (a) Periodic arrangement of thin-metal rods for effective negative permittivity [25], (b) Array
of SRR elements for effective negative permeability, (c) Realization of DNG MTM based thin wires and
SRR’s.

20
 Metamaterials

Most of the materials like dielectrics which occur naturally are designated as DPS material
or medium. The existence of ENG response can be naturally seen at optical frequencies
where noble metal nanoparticles tend to resonate in evanescent mode. Plasma also exhibits
similar characteristics at a specific range of frequencies [23]. The characteristic of MNG
structure is opposite to that of ENG structure. Few gyrotropic and ferrite materials are seen
to create an MNG medium. However, J. B. Pendry et al. [24] introduced a novel ENG
metamaterial in the form of a periodic arrangement of thin metallic rods which were
proven to exhibit negative permittivity. These thin wires act as electric dipole and they can
be artificially engineered using metallic wire lattice, and analyzed by Drude dielectric
model as [25];

  pe
2

 ( )   0 1   (1.43)
  (  je 

where ωpe is the damping frequency, Γe denotes the damping coefficient, and ɛ0 is the free-
space permittivity. Eq. (1.43) is derived from the Drude model, and it explains the
oscillatory behavior of electron inside the material which can produce negative
permittivity. This helps in understanding the evolution of negative indexed artificial
material called metamaterial. Eq. (1.43) was obtained from Drude oscillation equation
which is

2 r r (1.43a)
m  m  qE
t
2
t

where m – is the mass of an electron, 𝚪 – denotes the damping factor, q – is the charge per
particle and E – is the applied force. The first term shows the inertia of electron, and the
second term indicates the loss factor with respect to the applied electric force (qE). So, it
explains that when electric force is applied on metal, the electron acquires inertia and tends
to oscillate over a range of frequency. Gradually, the oscillation decreases and there is no
restoring force acting on electron. As a result, the permittivity profile approaches zero or
negative at a higher frequency range. Below ωpe, no EM waves were able to propagate,
and ωpe can be formulated as

2 c 2
 pe
2
 (1.44)
a 2 ln[a / (2r )]

21
 Introduction

where 2r is the diameter of the rod, a is the spacing between adjacent rod, and c is the
speed of light. Just like thin wire rods contributes to the resonant electric dipole, the split
rings were found to behave as resonant magnetic dipoles. In 2000, Pendry et al. [25]
realized artificial planar SRR to provide effective negative permeability. The geometry of
SRR is shown in Fig. 1.9(b). The size of SRR unit-cells was sub-wavelength of the
incident EM wave. The split rings can be made to resonate at a much higher wavelength in
comparison to their physical dimensions. The additional split ring concentric to the outer
ring generates high capacitance. The capacitance across the split gap is sufficient enough
to allow the flow of current. In the presence of a time-varying magnetic field transverse to
the plane of SRR, a large current is produced in the metal loops and a magnetic dipole
through its center. The structure becomes a resonant circuit, and several unit-cells can be
placed in different positions to form an array. The resonant frequency (f0) can be expressed
in terms of LC response of the SRR structure as

1 (1.45)
f0 
2 LC

The gap in the rings creates air-fringing field capacitance (C), and the current flowing
around the loop develops a self-inductance (L). The Drude polarization model is also used
to describe the permeability of artificially MTMs as [26];

  pm
2

 ( )  0 1   (1.46)
  (  j m 

Fig. 1.10 shows the plot of negative µeff resulted from the magnetic polarization of SRR.
The simultaneous negative value of permittivity and permeability is a combination of the
electric and magnetic response of rod and SRR structure. As shown in Fig. 1.9(c), the rod
is placed behind the SRR. Thus, for DNG medium the refractive index can be negative.

22
 Metamaterials

FIGURE 1.10 Effective permeability response of SRR structure [26].

1.5.4 Transmission Line Analysis of Metamaterials

Metamaterials are operated in the fundamental mode considering the effective media
as homogenous. This helps in defining the effective value of ɛ and µ. MTMs can also be
viewed as one dimensional (1D) transmission line (TL) because of their homogeneity [27,
28]. In comparison to the conventional TL, a homogeneous TL has an incremental length
(∆z  0) for wave propagation in the z-direction. However, the only restriction of
incremental length is ∆z << λg or ∆z < λg/4, where λg is the guided wavelength and ∆z is
equal to the average unit-cell size.

(a) (b)
FIGURE 1.11 (a) Ideal model of a uniform transmission line, (b) Equivalent circuit model of a lossless
infinitesimal size (z∆) CRLH line [27].

Fig. 1.11(a) shows the ideal incremented homogeneous line model for composite
Right/Left-handed metamaterial. The circuit can be viewed as a composite of Left-Handed
(LH) and Right-Handed (RH) circuit model (CRLH) of the conventional transmission line.
For MTM–TL with ɛeff and µeff, the impedance per unit-length is defined as 𝑍 / = Z/∆z and

23
 Introduction

admittance per unit-length is defined as 𝑌 / = Y/∆ [29]. The term LR (H.m) is a series right-
handed inductance, CL (F/m) is a series left-handed capacitance, LL (H/m) is parallel left-
handed inductance, and CR (F.m) denotes parallel right-handed capacitance. The complex
propagation constant (γ) is written as [29];

    j   Z / .Y / (1.47)

 1  ( / s ) 2  1  1  ( /  p )2  1
Z /  j   LR    j and Y /
 j 
 R C    j (1.48)
 C L  C L   LL   LL

where ωs and ωp are resonant frequency corresponding to series and the parallel branch of
the circuit respectively and defined as 𝜔 = 1/ 𝐿 𝐶 and 𝜔 = 1/ 𝐿 𝐶 . The resonance
factor κ (s/rad)2 is expressed as the sum of LRCL and LLCR. Therefore, expression of γ can
be written as [29];

2 2
    L 
  js( )       L
2
(1.49)
 R    

where ωR and ωL are right-handed and left-handed frequencies in rad/(m.sec) and defined
as 𝜔 = 1/ 𝐿 𝐶 and 𝜔 = 1/ 𝐿 𝐶 . s(ω) is a signed function expressed as

1;   min(s ,  p ) Left  handed 

 
s( )    (1.50)
1;   min(s ,  p ) Right  handed 
 

(a) (b)
FIGURE 1.12 (a) Equivalent circuit model of a periodic RH-TL having an infinitesimal incremental length
(z∆), and (b) Dual configuration of RH-TL [27].

If CL and LL are open-circuited, then CRLH–TL is reduced to purely RH–TL. Likewise,

when LR and CR are short-circuited, then CRLH–TL changes to purely LH–TL as shown in

24
 Metamaterials

Fig. 1.12. The s(ω) can be understood by considering phase velocity (vp) and group
velocity (vg). If the frequency ω < min (ωs, ωp), then vp = -vg shows antiparallel relation.
Therefore, phase constant (β) will be negative for LH–TL. In opposite to this, when
ω > max (ωs, ωp), vp = vg shows parallel behavior which means β is positive. The vp, vg
now can be expressed as

 
vp   s ( )
 2
    L 
2
(1.51)
     L
2

 R    
2 2
    L 
     L
2

  
1

 R    (1.52)
vp    
   | ( / R2 )  (L2 /  3 ) |

The further interpretation reveals that CRLH–TL circuit will have a bandpass response due
to the existence of RH and LH circuit elements [30]. At low frequencies, the RH–TL
elements result in a low-pass response, and the remaining circuit elements constitute to
high-pass filtering.

(a) (b)
FIGURE 1.13 (a) Schematic of the propagation mechanism of a generalized CRLH-TL unit-cell showing
dispersion and attenuation profile, (b) Transmission characteristics [30].

Altogether, the RH and LH components determine the transmission properties of the

line at all frequencies. It should be noted that the frequencies ωs, ωp are unbalanced, i.e.,
ωs ≠ ωp. The characteristic curve of CRLH–TL for dispersion (β) and attenuation (α) and

25
 Introduction

transmission coefficient (S21) profile is represented in Fig. 1.13. No propagation exists

between max (ωs, ωp) and min (ωs, ωp). Further, the characteristic impedance (Zc) of
CRLH–TL can be obtained as [30];

( / s ) 2  1
Zc  Z L (1.53)
( /  p ) 2  1

It can be deduced from Eq. (1.53) that unbalanced CRLH-TL has a purely imaginary
characteristic impedance in the frequency range of min (ωs, ωp) to max (ωs, ωp). Besides
the backward propagation of waves in LH media with simultaneously negative permittivity
and permeability (i.e., Z/ and -Y/), a similar analysis can be conducted to describes the TL
with the incremental circuit in which either of Z/ of Y/ is negative. In such a model,
evanescent mode is dominant and inhibits the propagation of the wave. Fig. 1.14 shows the
circuits for the existence of SNG response, where either of the constitutive parameters is
negative.

(a) (b)
FIGURE 1.14 (a) Equivalent circuit model of a periodic LH-TL having an infinitesimal incremental length
(z∆), and (b) Dual configuration of LH-TL [27].

In brief, depending upon the sign of Z/ of Y/, the TL can be categorized into four types
as RH-TL (forward propagation), LH-TL (backward propagation), CRLH-TL and
conventional TL. Table 1.3 summarized the properties of different transmission lines.

Table 1.3 Summary of the transmission line properties.

Impedance Admittance Permittivity Permeability Medium Propagation

(𝒁/ ) (𝒀/ ) (ε) (µ)
+ve +ve >0 >0 DPS Forward
+ve -ve >0 <0 MNG No
-ve +ve <0 >0 ENG No
-ve -ve <0 <0 DNG Backward

26
 Metamaterials

1.5.5 Network Theory Analysis of Metamaterials

In the previous section, the transmission line analysis was studied on the assumption
of homogeneity of metamaterial TL. The circuit requires a physical dimension of
distributed LC components to be much smaller than the λg. In many situations, the
condition of homogeneity may not hold. So, analyzing the periodicity of artificial
materials is more appropriate. The dispersion characteristics deduced by Collin [31]
describes the periodicity of any random shaped MTM cell in terms of ABCD matrix
elements as

A D
cos(  l )  (1.54)
2

Any MTM-TL can be modeled in general using T or π-network as shown in Fig. 1.15.

(a) (b)
FIGURE 1.15 Network topology for a general periodic unit-cell. (a) T-Type, (b) π-Type.

Here, Zs and Zp = 1/Yp denote series and parallel impedance respectively. Bloch-Floquet
analysis well describes the 1D-periodic microwave networks. The dispersion relation in
terms of S-parameters is expressed as [32];

1  S11S22  S21S12
cos(  l )  (1.55)
2S21

The Bloch impedance (ZB) is considered to account for the infinitesimal small unit-cell
which fails to adhere to the non-homogeneous nature of TL. Therefore, ZB is defined as the
characteristic impedance at the unit-cell terminal of periodic structure [33]. It can be
obtained directly from the equivalent ABCD matrix of asymmetrical periodic MTM as

27
 Introduction

2 BZ c
ZB  (1.56)
A D ( A  D) 2  4

For symmetric unit-cell, ZB = ∓𝐵𝑍 /√𝐴 − 1. In this connection, ZB for T and π-network
can be expressed as [33];

Z p2 ( ) Z s ( ) (1.57)
Z BT network  Z s ( )  [Z s ( )  2Z p ( )] and Z B network 
Z s ( )  2Z p ( )

where ZBT-network and ZBπ-network define the Bloch impedance of T-network and π-network
respectively. Let us now examine for Bloch impedance for LH-TL. Eq. (1.57) describes
the equivalent characteristic impedance for T-network and π-network topology of a
general unit-cell structure after simplifying the model using ABCD matrix. These
impedances are defined as the Bloch impedances. The Bloch impedance is characteristic
impedance defined at the unit-cell terminals of a periodic structure. It is also related to the
voltage and current at unit-cell terminals and plays a similar role as that of the
characteristic impedance in homogeneous transmission line. For periodic LH-TL, the Zs
and Yp over finite length (dl) is given by

1 1
Z s  Z / .dl   (1.58)
j (CL / dl ) jCLTotal

1 1
Yp  Y / .dl   (1.59)
j ( LL / dl ) j LL Total

In the above equations, CL-Total and LL-Total are the total inductance (H) and capacitance
(F). As periodic LH-TL does not exist naturally, it is therefore required to synthesize it
artificially. Consequently, the equation of dispersion is given as [34];

28
 Metamaterials

  .dl   .dl 1
sin  PLH   PLH 
 2  2 2 LL Total  CL Total
(1.60)
1 1
Therefore,  PLH  
2 ( LLTotal .dl )  (CLTotal .dl )  LL .CL

1.5.6 Retrieval Methods for Material Medium Parameters

As discussed in Section 1.5.4, the response of metamaterial is attributed by the effective

material medium quantities, i.e. permittivity and permeability. However, retrieval of
constitutive effective material medium parameters may not be possible directly. So far,
various metamaterial parameter extraction techniques have been reported in the literature
[35-38]. In these methods, permittivity and permeability are obtained and the behavior of
most of the MTM is anisotropic. A proper electric or magnetic tensor is a necessary
requirement and only after exciting the MTM plane by a plane wave with suitable electric
and magnetic polarizations, negative refractive index (R.I) is possible. In most of the case,
the near-field information of MTM structure is unavailable. Some of the popular extraction
techniques involve simulation scattering parameters (S11 and S21) and solving transfer T-
matrix [39, 40]. In this method, homogeneous model of MTM is considered. The effective
impedance and its refractive index are expressed using S-parameters as

(1  S11 ) 2  S212
z (1.61)
(1  S11 ) 2  S212

1  1 
n cos 1  (1  S112  S212 )  (1.62)
kd  2S21 

and

1  T21 
  ikT12  (1.63)
2  ik 
S11 
1 T21 
Ts   ikT12  
2 ik 

29
 Introduction

1
S21 
1 T  (1.64)
Ts   ikT12  21 
2 ik 

where Ts – T11 = T22, d is the thickness of homogenous slab or substrate, and k = 2πc/λ. By
using Eq. (1.61) and Eq. (1.62), permittivity and permeability are calculated as ɛ = n/z and
µ = n. In reality, solving Eq. (1.62) gives multiple branches of inverse cosine and creates
difficulty in determining the material medium parameters. In addition to this when the
MTM unit-cell is inhomogeneous, and then Eq. (1.63) and Eq. (1.64) led to an ambiguous
solution. This is because asymmetry in the model along the direction of propagation
results in the variation of S11 and S21 profile and retrieval method becomes even more
difficult. However, if the asymmetric unit-cell is in infinite repetition, the unique index
value can be extracted. First, T-matrix is expressed in terms of S-matrix and the index
value defined for symmetric structure through Eq. (1.65) gets updated to Eq. (1.66).

1
cos(nkd )  (1  S112  S212 )
2S21
(for symmetric, homogeneous model) (1.65)
1
or  (1  S222  S212 )
2S21

1  S11S21  S212
cos( d )  (for asymmetric, inhomogeneous model) (1.66)
2S21

Eq. (1.66) shows all the parameter extraction can be performed using all the S-parameter
elements irrespective of the wavelength to cell ratio or size of unit-cell. Therefore, the
index value for inhomogeneous can be obtained by averaged Savg value as 𝑆 𝑆 .
Further, impedance (zih) of the inhomogeneous medium is calculated as

T12
zih 2  (1.67)
T21

(1  S11 )(1  S22 )  S21S12 (1  S11 )(1  S22 )  S21S12

where T12  , and T21  .
2S21 2S21

30
 Metamaterials

A more robust extraction method of metamaterial parameters based on Karmers-

Kroing relationship algorithm offers an additional advantage where wave impedance and
refractive index can be uniquely solved using S-parameters [41]. Such an algorithm is
useful in determining an accurate value of electric permittivity, magnetic permeability, and
R.I. In this method, when the incident plane wave is transverse to the MTM surface, the
wave impedance and R.I are obtained by expressing reflection (S11) and transmission (S21)
coefficient to the complex wave impedance as

i 2 neff k0 d
R01 (1  e )
S11  2 i 2 neff k0 d
(1.68)
1 R e 01

i 2 neff k0 d
(1  R01
2
)e
S21  i 2 neff k0 d (1.69)
1  R01
2
e

where R01 = (z-1) / (z+1), z is the complex wave impedance, neff is the refractive index, ko
is the free space wavenumber and d is the maximum length of the unit-cell. Recalculating
Eq. (1.68) and Eq. (1.69) gives

(1  S11 ) 2  S21
z (1.70)
(1  S11 ) 2  S21

ineff k0 d S21
e  (1.71)
1  S11 R01

neff 
1
k0 d
Im[ln(e eff 0 )]  2 m  i Re[ln(e eff 0 )]
in k d in k d
 (1.72)

where ‘m’ is an integer number due to the branches of the logarithmic function. The
effective permittivity and effective permeability can be calculated as εeff = neff / z, and μeff =
neff * z respectively. This algorithm also explained that the phase continuity of S21 in the
range [-1800, 1800] led to no branching problem and ‘m’ is substituted as zero in Eq.
(1.72). On the contradictory side, phase change exceeding 1800 gives more branches of
logarithmic function contributing to negative refractive index. An important observation

31
 Introduction

between the slope of S21 and R.I states that whenever slope of S21 phase changes, R.I
becomes negative. This is not always true, but it may indicate such an occurrence.
Nicolson Rose Weir (NRW) method is another retrieval technique of MTM constitutive
parameters that is widely used for evaluating metamaterial properties of planar unit-cell
structure [42]. The simplified equations for calculating effective permittivity, permeability
and R.I are given as
2c(1  S21  S11 )
Effective relative permeability r  (1.73)
j d (1  S21  S11 )

2c(1  S21  S11 )

Effective relative permittivity  r  (1.74)
j d (1  S21  S11 )

2c ( S21  1) 2  S112
Effective refractive index n   (1.75)
j d ( S21  1)2  S112

where c is the velocity of light, d is the thickness of the unit-cell substrate, ω is the angular
frequency in rad/sec and S11, S21 are the reflection and transmission coefficient of unit-cell
respectively.

1.6 Thesis Organization

This thesis first introduces a brief discussion on the evolution of LTE technology and the
requirements for designing antenna for this wireless standard. The theory of metamaterial
and related mathematical analysis is explained in detail which has been subsequently used
for developing miniaturized LTE band planar antennas. The chapters presented in the
thesis routs through theoretical background of LTE, important antenna design objectives,
analysis of artificially engineered metamaterial structures, modeling and simulation of
metamaterial-based LTE antenna, and their practical implementation.
Chapter-2 highlights the state-of-art of different antenna designs, important research gaps,
and primary thesis objectives based on the literature survey and theoretical study. In
Chapter-3, the design of microstrip patch antenna and its geometrical modification are
presented for achieving the LTE-frequency band. The improvement in the frequency of
resonance and the number of bands are demonstrated by loading suitable metamaterial
structures. In Chapter-4, a novel antenna design method is presented which is based on the
concept of superstrate layer of thin dielectric and selectively designed MTM unit-cell

32
 Thesis Organization

structure. In Chapter-5, the research on the multiband antenna design approach is covered
by implementing a microstrip patch inspired by the Yagi-Uda antenna and then actively
coupling the MTM-SRR structure to the antenna. The proposed MTM-based
reconfigurable antenna offers multiple resonant frequencies over a wide range of wireless
standards. The fabrication and measurement results are discussed. The performance of the
antenna is compared with other reported LTE antennas. In the last chapter, important
points of the research work and findings are concluded. In addition to this, the direction of
work towards future research work is also highlighted in brief.

33
 Literature Survey

CHAPTER - 2

Literature Survey

In this chapter, different antenna designs and their microwave techniques are discussed
in detail which is reported for LTE communication technology. These include the use of
dielectric resonator in the antenna design for achieving wideband that is useful for the LTE
system in spectrum sensing. The MIMO-based antenna and frequency reconfigurable
antenna are also discussed which have been reported for obtaining multiple resonant
frequencies falling into the bands defined for LTE technology. Further, various
metamaterial-based antenna designs related to the proposed research work is also studied.
Based on the literature survey and research gaps reported in the field of LTE band antenna
design, the important objectives are formulated for the research work.

2.1 Antenna Design and Techniques for LTE Bands

2.1.1 Dielectric Resonator Antenna

Jamal Nasir et al. [43], proposed a two-port MIMO antenna loaded with rectangular
dielectric resonator (DR) as shown in Fig. 2.1.

(a) (b)
FIGURE 2.1 (a) Layout of MIMO antenna loaded with dielectric resonator, (b) Image of the fabricated
MIMO antenna [43].

34
 Antenna Design and Techniques for LTE Bands

The antenna has an area of 80×80 mm2 and fabricated on FR-4 substrate of relative
permittivity 𝜀 = 4.4. Along with the concept of using dielectric resonator for wideband
operation in S-band, a defective ground was also implemented. The ground plane has a
dimension of 80×25 mm2. A lower mutual coupling between the two 50 Ω ports was
achieved by using symmetrical slits in the ground plane. The antenna shows fundamental
resonance between 2.56 GHz to 2.64 GHz with S11 < -10 dB as shown in Fig. 2.2. This
resonating frequency can be used to operate on LTE Band No. 38.

FIGURE 2.2 Comparison of simulated and measured S11 and S21 of the dielectric resonator loaded MIMO
antenna [43].

Due to the presence of defective ground, an omnidirectional radiation pattern was obtained
and a higher diversity gain of 9.6 dB was achieved. Besides these, the antenna size was
relatively large and not suitable for multi-band operation in different LTE spectrum
applications.
In contrast to the above-reported design, Sadiq B. A. Rahim and co-authors [44]
presented a triple-band antenna based on hybrid dielectric resonator for 4G LTE band
applications. The antenna design consists of a simple microstrip line with a square slot in
the ground plane, as shown in Fig. 2.3.

35
 Literature Survey

FIGURE 2.3 (a) Prototype hybrid DRA loaded triple band antenna, (b) Ground plane, (c) Top radiating
plane [44].

The antenna has an overall dimension of 70 × 50 mm2 and fabrication on FR-4 substrate.
The combination of the radiating slot in the ground plane and higher dielectric layer in the
top plane excites triple-band resonance at 1.8 GHz, 2.6 GHz, 3.4 GHz (LTE band No. 33-
43) as shown in Fig. 2.4. The antenna has a broadside radiation pattern. However, the
presence of a thick dielectric layer makes antenna design bulkier and non-conformable for
wireless systems.

FIGURE 2.4 Simulated and measured S11 performance of the hybrid DRA [44].

T.-L. Chiu et al. [78] showed the antenna design technique for developing a compact DRA
for automotive LTE communication. In this, a 50×22×13 mm3 DR was placed on the FR-
substrate with two copper layer patterns were printed on the DR as shown in Fig. 2.5. By
using a matching circuit in the antenna LTE standard frequency bands i.e., 790 MHz–860
MHz, 1700 MHz– 2200 MHz, and 2500 MHz–2700 MHz were obtained as shown in Fig.
2.6.

36
 Antenna Design and Techniques for LTE Bands

(a) (b)
FIGURE 2.5 (a) Matching circuit of DRA, (b) Image of the DRA based antenna [78].

FIGURE 2.6 Plot of measured and simulated S11 of the DRA [78].

2.1.2 MIMO Antenna

A dual-band antenna design using a filtering patch antenna configuration with the
MIMO system was presented by Xin Yin Zhang et al. [45]. In the proposed antenna
design, two separate U-shaped patches were used to excite different frequencies. These
patches were excited by multi stub microstrip feed lines, as shown in Fig. 2.7.

(b)
(a)
FIGURE 2.7 (a) Configuration of the dual-band filtering antenna element, (b) Image of the fabricated
antenna in MIMO arrangement [45].

37
 Literature Survey

The antenna was modeled and fabricated using a substrate of relative permittivity 2.2. By
using multi stub lines, dual resonant modes were controlled and two nulls were realized in
the gain. The antenna resonates at 1.9 GHz and 2.6 GHz with a peak gain of 6.7 dBi and
7.3 dBi respectively, as shown in Fig. 2.8. The antenna also maintains a low-profile. A
lower mutual coupling (< -19.2 dB) was made possible by using a grounded stub line
between the two antenna elements. In addition to these, the size of the antenna was still
large in comparison to the other reported antenna, and limited bands were obtained which
are sufficient enough for wideband sensing and frequency scanning for large bandwidth in
LTE based systems.

FIGURE 2.8 Measured performance of the filtering antenna for LTE band operation [45].

Dual feed U-shaped open-slot antenna in PIFA design was another type of configuration
reported by Imee R. Barani et al. [46] for LTE MIMO operation in metal-framed
smartphones. The antenna was designed on a circuit board of 78 × 150 mm2, which is
comparable to the size of a smartphone antenna. The ground plane was made in contact
with the metal frame. In the first antenna, a microstrip line with loaded capacitance Cb is
used which is backed by a U-slotted ground plane. The second antenna was realized by
shorting the microstrip line to the ground with two inductance (L1 and L2) and capacitance
C1 and Cb between the gaps created in the feed line, as shown in Fig. 2.9. From the
measured results depicted in Fig. 2.10, the two antenna elements cover LTE band of 746-
960 MHz and 1710-2690 MHz respectively. The design also allowed MIMO operation.
The broadside radiation pattern makes its suitable candidate for mobile phone antenna.
However, the reported antenna design being large in the physical dimension fails to meet
many of the upcoming mobile phone models and system integration. Even though such

38
 Antenna Design and Techniques for LTE Bands

antenna has higher efficiency up to 60 %, the gain of the antenna system was around 2 dBi
and does not allow radome configuration for gain enhancement.

(a) (b)

FIGURE 2.9 (a) Geometry of the dual U-slot antenna, (b) Fabricated prototype antenna [46].

FIGURE 2.10 S-parameter response of the Antenna-1 and Antenna-2 [46].

Recently, monopole antenna with coupled metamaterial split ring resonator (SRR) has
shown great potential in achieving a low-profile MIMO antenna system. D. Sarkar and K.
V. Shrivastava [47], reported a compact four-element SRR-loaded MIMO antenna which
can operate on WLAN, WiMAX, Wi-Fi, and 4G-LTE wireless standard. The antenna
geometry is shown in Fig. 2.11.

39
 Literature Survey

FIGURE 2.11 Proposed four-element SRR-loaded MIMO antenna prototype design [47]

By the inductive coupling of SRR to the monopole antenna, high-frequency of resonance

for SRR was combined with the low-frequency resonance mode of the unloaded MIMO
antenna configuration. Thus, the antenna operates at 2.93 GHz and 5.65 GHz, as shown in
Fig. 2.12. Isolation of up to 14 dB was achieved between the antenna elements and yet
maintaining a compact size of 0.103 𝜆 .

FIGURE 2.12 Measured S-parameter of the metamaterial SRR loaded MIMO antenna [47].

2.1.3 Frequency Reconfigurable Antenna

Frequency reconfigurability allows many different antenna design implementations

using stubs, slots, and resonators. By using active elements like PIN diode or varactor
diode, coarse tuning or fine-tuning of frequency bands is achieved. The PIN diode
connects part of the antenna to either a resonator or a metering stub. This changes the total
electrical length of the antenna and proper frequency diversity is obtained. On the other
hand, by using varactor diode, the reverse junction capacitance of the diode contributes to
a minor or major change in electrical length of the antenna depending upon its position
with respect to the feed point.

40
 Antenna Design and Techniques for LTE Bands

S. Danesh et al. [48], reported a compact frequency-reconfigurable antenna using four

resonators, each loaded with a dielectric material and PIN diode was connected between
the two elements. The antenna schematic is shown in Fig. 2.13. The total dimension of the
antenna is 20×36 mm2 and fabricated using a low-loss material substrate (Taconic) of
relative permittivity 3.2. The dielectric resonators of dielectric constant 10 excited
𝑇𝐸 mode in the antenna.

(b)
(a)
FIGURE 2.13 (a) Schematic of the reconfigurable antenna with dielectric resonators, (b) Image of the
fabricated prototype antenna [48].

The different switching conditions of the PIN didoes results in four different resonant
frequencies at 1.89 GHz, 2.14 GHz, 2.53 GHz, and 2.77 GHz, as shown in Fig. 2.14. Also,
the partial ground plane helps in obtaining an eight-shaped radiation pattern, which is
broadside in nature. Such techniques are useful for designing a compact and performance
oriented antenna for various LTE band applications.

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 Literature Survey

FIGURE 2.14 S11 response of the antenna for different switching states of PIN diodes [48].

I. A. Shah et al. [49], developed a Hexa-band frequency reconfigurable antenna by

using various stubs and PIN diode connection, as shown in Fig. 2.15. The antenna size was
compact (36×16 mm2). The antenna was operated in four different modes of switching. In
State-1 and State-2, a single-band mode was obtained at 3.5 GHz and 4.8 GHz
respectively, whereas the other two States resulted in dual-band mode at 2.10 GHz, 4.15
GHz, and 2.4 GHz, 5.2 GHz respectively as shown in Fig. 2.16. Gain between 2.1-4.3 dBi
and a higher frequency bandwidth of 2.7-12 GHz was obtained with improved impedance
matching in the circuit topology. Based on the measured S11 results, the antenna depicts
excellent frequency diversity, which is an important design parameter for LTE band
antennas.

(c)
(a) (b)
FIGURE 2.15 (a) Front view of the reconfigurable antenna, (b) Ground plane view, (c) Image of the
fabricated prototype antenna [49].

42
 Antenna Design and Techniques for LTE Bands

FIGURE 2.16 Simulated and measured S11 result of the proposed reconfigurable antenna in four different
states of the PIN diode bias [49].

Recently, Abdullah J. Alazemi et al. [79] showed the applicability of planar inverted F-
antenna with shorted pins and stubs for designing multiband LTE antenna. Two varactor
diodes connect to different stubs in the resonating plane as shown in Fig. 2.17.

FIGURE 2.17 Single feed triple-band antenna [79].

Also, a parallel LC harmonic-trap filter is built inside the antenna to suppress the
harmonics generated by the low-band resonance, which enhances the tuning capabilities at
the high-band. Change in the capacitance of the antenna circuit excited triple frequency

43
 Literature Survey

band at low-band (0.7-1 GHz), mid-band (1.6-2.1 GHz), and high-band (2.1-2.7 GHz) as
shown in Fig. 2.18.

FIGURE 2.18 S11 result of antenna with and without LC tuning filter [79].

2.1.4 Metamaterial Based Antenna

The unique properties of metamaterial like -𝜀, -𝜇, and -𝜂 interestingly helps in
controlling the bandwidth, gain, and electrical size of the antenna. This has been recently
explored by many researchers for LTE antenna design and in other application areas. For
example, Md. Mehedi Hasan et al. [50] designed a dual-band metamaterial antenna for
LTE/Bluetooth/WiMAX wireless systems as shown in Fig. 2.19.

(a) (b)
FIGURE 2.19 (a) Front view of the fabricated metamaterial coupled antenna with DGS, (b) Back view of
the antenna with partial corrugated ground plane [50].

The size of the antenna is 42 × 32 mm2. In the top plane of the antenna, multiple bend
lines are connected to the 50 Ω impedance line. Additionally, two metamaterials of

44
 Antenna Design and Techniques for LTE Bands

selective resonance are placed near to the feed line. The coupling of metamaterials to the
antenna helps in obtaining triple-band resonance at 0.59 GHz, 2.55 GHz, and 3.17 GHz, as
shown in Fig. 2.20.

FIGURE 2.20 Simulated and measured S11 of the proposed metamaterial-based antenna [50].

Furthermore, the partial ground plane promotes a broadside radiation pattern. It is

seen that placing metamaterial ring resonators in the ground plane of the antenna not only
helps in improving broadband gain but also results in the slight tuning of the frequency
bands. These properties are undoubtedly useful for designing a low-profile planar antenna
for LTE band applications.
Similarly, another class of metamaterial called Electromagnetic Bandgap (EBG)
structure was explored in the antenna by R. Dewan et al. [51]. The Electromagnetic
Bandgap (EBG) material is defined as a sub-wavelength periodic artificial structure
composed of a planar or three-dimensional arrangement of metal and dielectric”. In EBG
structure, the traveling electromagnetic wave experiences a periodic variation of dielectric
permittivity, which causes destructive interference of the incident and scattered wave
vectors, and therefore, inhibits the propagation of microwave signals over a band of
frequencies. So, the EBG structure can be used to create selective frequencies of zero
transmission. The design of EBG unit-cells consists of circular ring structures and circular
patch with shorting via pins. The different states were defined based on the connection of
EBG structure to the partial ground plane of the antenna. The antenna and EBG structure
were realized on the FR-4 substrate as shown in Fig. 2.21.

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 Literature Survey

(a) (b)

FIGURE 2.21 (a) Layout of the pentagon-shaped patch with EBG unit-cell in the partial ground plane, (b)
Image of the fabricated prototype antenna [51].

Single and dual bands were obtained in different switching states of the EBG unit-cell
structures as shown in Fig. 2.22. The measured gain was relatively higher (3.24-5.79 dBi)
in comparison to the other reported defective ground (DGS) antennas. The advantage of
using EBG structure is that the presence of frequency bandgap helps the antenna in
proving efficiency. This is because the power gets completely inhibited in the bandgap
range, and maximum power can be directed to other frequency bands for efficient
radiation mechanism.

FIGURE 2.22 S11 variation in the antenna due to change in the state of EBG unit-cell structures [51].

The concept of improving antenna resonance and efficiency is seen in the work
reported by M. M. Hasan and co-authors [80]. Initially, a metamaterial unit-cell structure
comprising of a square split ring and circular split ring resonator was modeled, and its

46
 Antenna Design and Techniques for LTE Bands

material medium response was extracted from the measured S-parameters. The geometry
of unit-cell structure and its response is shown in Fig. 2.23. The refractive index is
negative at 3 GHz, and thus, it can be potentially used to improve the antenna resonance.

(a) (b)
FIGURE 2.23 (a) Schematic of the unit-cell structure, (b) Magnitude response of the effective material
medium parameters [80].

The loading of two unit-cells in the arms of the resonating patch is demonstrated. Further,
the concept of the defective ground was also used to improve the wideband resonance in
the antenna. Fig. 2.24 shows the image of the fabricated antenna.

(a) (b)
FIGURE 2.24 Image of the metamaterial-based antenna. (a) Top view, (b) Bottom view [80].

In the boundary conditions of the MTM unit-cell, H-field is tangent to the y-plane. Thus,
MTM unit-cell was placed in the top plane of the antenna in order to excite the MTM with

47
 Literature Survey

similar boundary conditions. From the result of the return loss shown in Fig. 2.25, the
antenna has fundamental resonance at 2.4-2.6 GHz. However, with MTM loading, an
additional frequency band near 3 GHz is excited. Also, the wideband response due to the
zig-zag ground plane pattern and T-stubs can be seen at a higher frequency range.

FIGURE 2.25 Simulated and measured return loss plot of the antenna with and without MTM loading [80].

2.2 Research Gap and Motivation

The antennas for fourth generation (4G) long-term evolution (LTE) are under
continuous development for achieving a compact design with multiple frequency bands. In
recent years, a high data rate in LTE technology has attracted many personal wireless
applications for industrial and scientific areas, especially in the LTE-A band (2 GHz) [52].
To support the channel capacity in the operating frequencies of LTE system, electrically
small antennas are needed in the customer premises equipment (CPE) and handheld
receivers. To meet the emerging requirements of LTE communication, various microwave
techniques, along with multiple-input multiple-output (MIMO) mode are reported. I.
Dioum et al. [53], introduced a novel compact design of 3D inverted-F-antenna (IFA) with
MIMO for dual-band of LTE 700 and LTE 2.5-2.7 GHz. Generally, single resonance is
obtained in such designs, and more than one frequency band is generated with the help of
parasitic elements [54]. The design of the MIMO antenna with a shaped dielectric
resonator (DR) solves the problem of achieving a low-frequency band of LTE band 12 and
17 [55]. In such a design, a coaxial probe and microstrip line are employed simultaneously
and placed in the close vicinity of the DR [56-58]. In addition to this, different shapes like
C-shape [59], F-shape [60], sectored conical [61] and tetraskelion [62] are either coupled

48
 Research Gap and Motivation

with the feed line or placed above the slotted patch for wideband applications. Recently,
dual-band antennas integrated into the last stage of the filter are being proposed for LTE
systems [63, 64]. However, in these filtering antennas, the insertion loss is higher due to
the presence of the filter circuit. X. Y. Zhang [45] developed a dual-band filtering patch
for MIMO LTE in which two U-shaped slots were incorporated into the patch and a multi
stub feed line was used to excite resonant modes for B39- and B38-band.
Another concept of providing a separate single path to a ring resonator is reported by
Y. Zhang [65], wherein good isolation between the antennas is achieved. D. Sarkar and
coauthor presented a split ring resonator (SRR) loaded four inverted L-monopole printed
antennas having 4G LTE band (3.4-3.6 GHz) with an omnidirectional radiation pattern and
isolation (> 14 dB) amongst the radiators [47]. Besides the MIMO configurations, LTE
tablet computer antenna and smartphone antenna in PIFA technology are widely utilized.
The antenna developed by K. –L. Wong and co-authors explore various designs of the
LTE antenna [66-68]. These involve coupled fed stripline antenna with inductive coupling
through microstrip and lumped elements, two antennas mounted on the edge of the metal
plate, and dual feed U-shaped open slot antenna at the top and bottom side of the ground
plane. The microwave technique comprising of the stacked dielectric resonator of different
permittivity is found to improve the quality factor (Q) of the antenna, which contributes to
the narrowband resonance. A. K Jyani [69] realized such an antenna using a coaxial probe
and two or more DR’s.
Additionally, reconfigurable frequency DRA with numerous frequency-tuning
methods is available in the literature, where multiple DR is used and tuned through
modifying the transmission line using a diode and parasitic lumped elements [48]. From
[70], it was noticed that MTM loading into the slotted ground of the antenna is a promising
technique for designing a low-profile dual-band antenna. In parallel, another artificial
structure EBG is seen to be integrated into the planar antenna for minimizing cross-
polarization and improving S11 [71]. Moreover, frequency selective surfaces (FSS) and
MTM as superstrate to the DRA antenna is useful in confining the electromagnetic
radiations [72, 73] and even controlling the resonant frequency.
In the framework of multiband communication, its application in the automotive system
and wireless sensor network, it demands a low-profile antenna with improved gain and
frequency switching capability [74]. The integration of wireless standards belonging to
Long Term Evolution (LTE), Wireless Local Area Network/Worldwide Interoperability

49
 Literature Survey

for Microwave Access (WLAN/WiMAX) into a single radio platform typically impacts
the transceiver design and complexity. The typical antenna designs fail to account for the
underlying challenges when it comes to their practical applicability. Open slot with an L-
shaped metallization strip and the metal frame is a standard LTE band antenna that offers
multiband resonances [75]. However, the requirement of large ground plane and precise
metal wire strip design often restricts its use to limited wireless systems, and it required an
additional matching network. Tuning of frequency band using multiple dielectric
resonators placed on differently interconnected stubs has also been reported [48].
Reconfigurable antenna design using shot stubs connected via PIN diode, slots in the
radiating plane and defective ground plane (DSG) are found suitable for operating at
different wireless standards and maintains a low-profile structure [49]-[76, 77]. A dual-
band antenna reported by M. M. Hasan et al. [50], utilizes square metallic strips along with
two coupled metamaterial ring resonators and a defective ground for radiation. But the
frequency bands are limited.
Another important parameter of interest is the omnidirectional radiation pattern,
which requires positioning of stub or radiator in various directions. As a result, the
conformability gets compromised during the antenna installation in routers or wireless
sensor nodes. Thus, a compact planar antenna design is necessary to overcome the above-
discussed practical difficulties in LTE band antenna technology.

2.3 Research Objectives

The research work reported in this thesis explores different design topologies of microstrip
patch antenna by incorporating stubs, slots, and defective grounds. In addition to this,
various methods are adopted to load the metamaterial structure at a suitable position in the
antenna for achieving a selective band for LTE communications. In this context, some of
the important objectives of the thesis are outlined as follows;

 Design of metamaterial-based microstrip antenna for selective LTE band

communication.
 Design of novel microwave techniques using MTM unit-cell and dielectric resonator
for an increased number of LTE frequency bands.

50
 Research Objectives

 Design of frequency reconfigurable antenna based on switching between MTM and

microstrip patch for simultaneous support of different LTE band and other wireless
standards
 Electromagnetic simulation and Fabrication of the proposed antennas.
 Testing and performance comparison of the proposed antenna with other reported LTE
band antennas.

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 Literature Survey

2.4 Research Methodology

Start Microstrip Patch Antenna Designing

Geometrical Optimization

T-Shaped Stubs Defective Ground Slots in Patch

Single Band Antenna Design - CSRR Loaded MSA

Dual Band Antenna Design - Connected Ring Resonator Loaded MSA

Triple Band Antenna Design - Thin-layer Dielectric and SRR Stacked MSA

Frequency Reconfigurable Antenna Design – SRR Switched MSA

Antenna Simulation Using CST-MWS

Fabrication and Testing of the Proposed Antennas

Performance Comparison Stop

52
 Single Band Antenna Design and Implementation

CHAPTER - 3

Development of Metamaterial Based Single Band

and Dual Band Microstrip Antenna

In this chapter, the initial study on the design of microstrip patch antenna and its
geometrical modification based on stubs, slots in the radiating plane, defective ground for
broader radiation pattern is presented. Next, different configurations of MTM like CSRR,
connected ring resonator were loaded at the center of the patch and its performance
simulation, and practically analyzed.

3.1 Single Band Antenna Design and Implementation

The design methodology of a microstrip patch for LTE band operation begins with the
numerical calculation of the width of the patch [10], based on the following assumptions
i.e.,
 Substrate Relative Permittivity (𝜀 ) = 4.4
 Height of substrate h = 1.6 mm
 And higher cut off frequency (fh) = 5 GHz
Following the MSPA calculations discussed in Chapter 1: Section 1.4, a patch was
modeled, as shown in Fig. 3.1(a) Iteration-1. Slots were created along the radiating side in
order to create discontinuities to excite a resonance mode in a miniaturized dimension.
Time-domain solver of CST Microwave Studio Suite was used to model and simulate the
antenna parameters. The simulated S11 of Iteration-1 is at 5.56 GHz with a magnitude of -
5.9 dB. Based on the resonance obtained from Iteration-1, additional T-shaped stub was
connected to the non-radiating side of the patch in order to increase the electrical length of
the structure and obtain a lower frequency of resonance near to the LTE bands.

53
 Development of Metamaterial Based Dual Band
and Triple Band Compact Microstrip Antenna

Iteration-1 Iteration-2 Iternation-3

FIGURE 3.1 Microstrip patch antenna design iterations for achieving strong fundamental resonance using
metamaterial complementary ring resonator structure.

The simulated S11 of Iteration-1 is at 5.56 GHz with a magnitude of -5.9 dB. Based
on the resonance obtained from Iteration-1, additional T-shaped stub was connected to the
non-radiating side of the patch in order to increase the electrical length of the structure and
obtain a lower frequency of resonance near to the LTE bands. As seen in Fig. 3.5, the
resonance was further decreased to 2.38 GHz with S11 = -7.5 dB. Subsequently,
metamaterial complementary split-ring structure (CSRR) was loaded in the patch, and it
resulted in a strong resonance at 2.38 GHz with S11 < -15 dB.

(a) (b)
FIGURE 3.2 Layout of proposed metamaterial loaded CSRR microstrip patch antenna for LTE band
operation.

54
 Single Band Antenna Design and Implementation

Table 3.1 Illustration of antenna dimension obtained from Iteration-3.

Parameter Dimensions (mm)

L 25
W 20
f 2
P1 12
g 1
r 2
P 8
t 0.5
S1 3
S2 6
S3 18
W1 5

It is important to note that CSRR structure is polarized generally at a higher frequency,

which inhibits the resonances localization of electric flux at higher frequencies. As a
result, the majority of the RF energy is directed towards resonance condition occurring at
2.38 GHz, and we obtain a strong resonance in the S11 profile. The complete layout of
metamaterial CSRR loaded microstrip patch antenna with the defective ground is depicted
in Fig. 3.2. The defective ground was essentially adopted for achieving an eight-shaped
radiation pattern which is broadside in direction. In Table 3.1, the important dimensions of
the antenna evolved from Iteration-3 are illustrated.

3.1.1 Fabrication and Measurement Setup of CSRR Loaded Microstrip Patch

Antenna

The proposed antenna obtained in iteration-3 was fabricated on FR-4 substrate using
photolithography process. The image of the fabricated antenna is shown in Fig. 3.3. A
50 Ω SMA female connector is soldered to the microstrip feed line. The antenna
measurement setup is shown in Fig. 3.4. In the measurement setup, the vector network
analyzer (VNA, Field-Fox-N9923A) was first calibrated using Open-Short-Load method
over a frequency span of 1-4 GHz with 201 frequency sweep points and 5 kHz IF
bandwidth for obtaining a good resolution in the measured data points.

55
 Development of Metamaterial Based Dual Band
and Triple Band Compact Microstrip Antenna

(a) (b)
FIGURE 3.3 Image of the fabricated prototype antenna. (a) Front view, (b) Back view.

FIGURE 3.4 Antenna measurement setup.

A coaxial cable was used to connect VNA to the antenna. A RF absorber material was
placed at the bottom plane of the antenna to avoid any ground reflections of EM waves
from the nearby metallic source.

3.1.2 Results and Discussions

The comparative response of reflection coefficient (S11) of the antenna from iteration-1,
iteration-2, and iteration-3 are clear indicator of the usefulness of metamaterial in the
antenna design applications. A single band obtained at 2.38 GHz is narrowband and lies
within the LTE spectrum. This selective frequency band can be obtained at various other

56
 Single Band Antenna Design and Implementation

frequencies by properly adding the stubs and CSRR dimension in the radiating plane of the
patch antenna. It is found that a compact microstrip patch antenna can be developed by
first designing the patch with higher cut-off frequency, and later modifying the geometry
by loading artificially engineering material to alter the effective permittivity response of
the antenna. The measured S11 result of the antenna is obtained at 2.4 GHz, which shows a
close matching with the simulated data at 2.38 GHz, as shown in Fig. 3.6. Minor ripples in
the S11 response are possibly due to the loss nature of the material, and slight over-etching
of the metallization in some of the specific areas in the patch. The plot of simulated
voltage standing wave ratio (VSWR) represented in Fig. 3.7 shows the value of VSWR
below 2. This indicates that higher impedance matching between the RF source and
proposed prototype antenna as a load. It also explains that minimum reflection travels back
from the antenna to the source, and hence, the maximum of the power gets radiated by the
antenna. The 3D and polar plot of the radiation pattern are represented in Fig. 3.8. Due to
the presence of defective ground plane and CSRR geometry at the center of the patch, an
eight-shaped pattern was obtained which fulfill the broadside radiation pattern requirement
for LTE wireless communication.

FIGURE 3.5 Comparison of simulated S11 of the antenna designs of iteration-1, iteration-2 and iteration-3.

57
 Development of Metamaterial Based Dual Band
and Triple Band Compact Microstrip Antenna

FIGURE 3.6 Comparison of simulated and measured S11 of CSRR loaded microstrip patch antenna for LTE
band.

FIGURE 3.7 Simulated VSWR response of the propose CSRR loaded microstrip patch antenna.

However, the antenna gain is just 2 dBi. Thus, gain enhancement techniques are like
metamaterial radome or artificial magnetic conductor is needed in such antenna designs.
From the simulated surface current distributions obtained at 2.38 GHz, it can be observed
that current density is higher towards the non-radiating plane and minimum near the
radiating edges. The current density gets lowered due to the presence of CSRR at the
center of the patch. As a result, the region of the current minima creates higher resistive
impedance and enhances the resonance mechanism.

58
 Dual Band Antenna Design and Implementation

(a) (b)
FIGURE 3.8 (a) 3D radiation pattern of the antenna, (b) Polar plot of the radiation pattern in E-Plane.

FIGURE 3.9 Simulated surface current of CSRR loaded microstrip patch antenna.

3.2 Dual Band Antenna Design and Implementation

The dual-band antenna design is also inspired by loading of metamaterial inside the
radiating patch. Instead of CSRR, a connected ring resonator is used to excite higher-order
resonance mode. The geometry of the proposed antenna is shown in Fig. 3.10. Here,
similar to the design shown in Iteration-3, tuning stub was used to lower the frequency of
resonance. In addition to this, the two-split ring in the geometry was connected to the inner
and outer metallization of the patch.

59
 Development of Metamaterial Based Dual Band
and Triple Band Compact Microstrip Antenna

(a) (b)
FIGURE 3.10 Layout of metamaterial connected ring resonator loaded microstrip patch antenna. (a) Top
view, (b) Ground plane.

Moreover, slots of four different sizes were selectively etched from the ground to create a
narrow band resonance with higher quality factor. All the important dimensions of the
antenna are described in Table 3.2.

Table 3.2 Dimensions of the metamaterial loaded dual band microstrip patch antenna.

Parameter Dimension (mm)

L 25
W 20
fw 3
fl 9
CL 14.5
Pw 5
t 0.5
Sw 3
St 1.5

3.2.1 Fabrication of Connected Ring Resonator Loaded Microstrip Patch Antenna

The prototype antenna was realized on RT/Duroid substrate of relative permittivity

𝜀 = 3.38, loss tangent 𝑡𝑎𝑛𝛿 = 0.0018, and height of 1.52 mm. SMA female connected
was soldered to the printed circuit board as shown in Fig. 3.11.

60
 Dual Band Antenna Design and Implementation

(b)
(a)
FIGURE 3.11 Image of the fabricated prototype dual band metamaterial loaded microstrip patch antenna (a)
Top view, (b) Bottom view.

The loss-low material was particularly selected for better resonance performance of
the antenna. The copper metallization was cleaned and coated with a thin layer of silicon
resin in order to avoid oxidation of the copper at room temperature.

3.2.2 Results and Discussion

The simulated result of antenna reflection coefficient (S11) depicts highly selective and
narrowband resonance at dual frequencies, i.e. 1.66 GHz, and 3.475 GHz. The plot of S11
is shown in Fig 3.12(a). It is observed that creating defects in the ground plane by
properly removing the metallization help to create narrow-band resonance.

(a) (b)
FIGURE 3.12 Simulated and measured S11 response of connected metamaterial ring resonator loaded
microstrip patch antenna.

61
 Development of Metamaterial Based Dual Band
and Triple Band Compact Microstrip Antenna

It should be noted that such a highly narrow bandwidth is difficult to achieve, unless the
material is lossless. In the simulation, lossless material definitions were assigned in order
to estimate the antenna performance in case of low-loss Rogers dielectric laminate.

FIGURE 3.13 Measurement of S11 performed using Anritsu vector network analyser.

This is because a defective structure in the antenna ground plane causes frequency
bandgap and only allows a particular band of frequency to propagate through the antenna
surface. The measurement of the resonant frequency of the prototype antenna was
recorded at 1.74 GHz and 3.41 GHz as shown in Fig. 3.12(b) and Fig. 3.13. The 3D
radiation pattern plotted in Fig. 3.14 shows the main lobe in the z-direction with gain of
1.12 dBi.

FIGURE 3.14 3D radiaton pattern of connected metamaterial ring resonator loaded microstrip patch
antenna.

62
 Dual Band Antenna Design and Implementation

(a)
(b)
FIGURE 3.15 Simulated antenna surface current distribution at (a) 1.66 GHz, (b) 3.475 GHz

The 3D radiation pattern is obtained at fundamental resonance, i.e. 1.66 GHz. Low gain is
one of the challenge in DGS antennas which needs attention. The radiation mechanism can
be understood from the plot of antenna surface current distribution, as shown in Fig. 3.15.
At 1.66 GHz, the majority of the current is centered at the ring resonance. As a result, the
T-shaped stub and left-right edges of the patch contribute to the radiation. On the other
hand, at higher-order resonance (i.e., 3.475 GHz), the effect of ring resonator can be seen
where current is minimum near the ring resonator and diverted towards the stubs in the
patch. Thus, resonance dominantly occurs due to ring-resonator.

Summary
The design of microstrip antenna and its geometrical modification by loading different
MTM resonators is analyzed. The weak fundamental mode of the MSA obtained at 2.38
GHz was significantly improved by loading a CSRR at the center of the patch. The S11 was
found to improve from -7.56 dB to around -17 dB. The antenna was fabricated and tested.
In this continuation, geometry of MTM ring resonator was changed to connected ring-
resonator and its effect was investigated. It was observed that loading the connected ring-
resonator provides an additional direct contact between the patch and the MTM. As a
result, resonant behavior of the MTM gets combined with the fundamental of MSA as
higher order mode.

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 Development of Metamaterial Based Dual Band
and Triple Band Compact Microstrip Antenna

CHAPTER - 4

Development of Metamaterial Based Dual Band

and Triple Band Compact Microstrip Antenna

In this part of the research work, an attempt has been made to achieve multiple narrow
bands for frequency selective communication in the LTE band. The antenna is inspired by
the metamaterial SRR and a stack consisting of thin-layer dielectric resonator and a novel
metamaterial unit-cell is mounted above the radiating patch. In the proposed antenna
instead of directly loading the MTM unit-cell in the radiating patch, a novel MTM unit-
cell is coupled through the dielectric resonator. Such a microwave device design
methodology provides opportunity for loading different MTM structures in order to get
multiple frequency bands without changing the basic antenna topology. In Section 4.2, the
antenna design process is discussed. Section 4.3 explains the selection criteria for
important antenna dimensions through parametric analysis. The antenna fabrication and
measurement setup are illustrated in Section 4.4. Section 4.5 discusses the measured
results, whereas, the antenna performance is analyzed in Section 4.6. In the conclusion
section, the important findings are summarized.

4.1 Antenna Design

4.1.1 SRR-Based Microstrip Patch Antenna

The geometry of the antenna is projected from the design of a conventional microstrip
patch antenna which is later modified in accordance with the specification of LTE system.
Initially, a rectangular patch is designed using cavity model [9] as;

64
 Design of Metamaterial Unit-Cell Stacked TL-DR
Antenna

c
Lp 
r 1 (4.1)
2 fo
2

c
Pw  (4.2)
2 f 0  eff

Where, Lp = 14 mm is the patch length, Pw = 16 mm is the patch width, fo ≈ 5 GHz is the

assumed resonant frequency, εr = 4.4 is the relative permittivity of the substrate, εeff is the
effective permittivity, and c = 3 × 108 m/sec is the velocity of light. The corner edges of
the patch are chipped away which in turn creates a discontinuity in the flow of surface
current. By doing so, the radiating side along the length Pw1 is reduced, and thereby
provides a provision for introducing additional stubs as frequency tuning elements. For
dual frequency operation, a connected split ring resonator [81] is loaded in the top plane of
the patch as shown in Fig. 4.1(a).

(a) (b)
FIGURE 4.1 Layout of the SRR-based antenna with the reduced ground plane. (a) Top view,
(b) Bottom view.

Further, tuning stub in the shape of inverted-E is connected to the non-radiation side of the
patch which helps in altering the electrical length of the antenna. The ground plane of the
antenna is reduced by W1 from the total length L in order to increase the frequency
bandwidth as shown in Fig. 4.1(b).

65
 Development of Metamaterial Based Dual Band
and Triple Band Compact Microstrip Antenna

The antenna is fed by a 50 Ω microstrip line and has an overall dimension of L × W mm2
i.e. λ0/3 × λ0/4, where λ0` is the center of operating wavelength. The geometrical
dimensions of the antenna after the parametric analysis are listed in Table 4.1.

Table 4.1 Illustration of the antenna layout dimensions.

Parameter Dimension (mm)

L 25
W 20
W1 5
fw 2
PL1 6
Pw1 8
Pw2 4
R 2
G 1
T 0.5
S1 3
S2 6
S3 18

4.1.2 Thin Layer Dielectric Resonator Loaded Patch Antenna

In the configuration of thin layer dielectric resonator antenna (TL-DRA), a bare dielectric
substrate of relative permittivity εrd = 10.2 and height of h1 = 1.27 mm is mounted above
the SRR patch. The TL-DR has an area of 12 × 12 mm2 and it is shown in Fig. 4.2.

FIGURE 4.2 Perspective view of the SRR-based antenna loaded with TL-DR.

66
 Design of Metamaterial Unit-Cell Stacked TL-DR
Antenna

The surface current flow in the SRR provides a magnetic current parallel to the plane of
DR substrate and excites TEδmn resonant modes in the TL-DR and can be obtained from
[44] as

c
f0  k x2  k y2  k z2 (4.3)
2  rd
2
m n  1  k0 
ky  , kz  , k x  2 tan 1 1     1
b 2h1   rd  k x 

where, k0 = 2π/λ0, kx, ky, and kz are the wave vector in x, y, and z-direction respectively and
b = 12 mm is the width of the TL-DR. The coupled resonance in TL-DR alters the
impedance of the patch antenna.

4.2 Design of Metamaterial Unit-Cell Stacked TL-DR Antenna

4.2.1 Unit-Cell Modeling and Analysis

The basic geometry of the unit-cell structure has been inspired from the S-shaped
resonator. The design of unit-cell represented in Fig. 4.3 consists of S-shape resonator in
the top plane with rectangular stubs of width U2. These two stubs are located in the gap
region of the resonator in order to increase the electrical length in comparison to the
antenna.

FIGURE 4.3 Layout of the proposed unit-cell (UL = 12 mm, GL = U3 = 2 mm, U1 = a = 1 mm, U2 = 1.8 mm,
U4 = 10 mm, gt = 1.5 mm)

67
 Development of Metamaterial Based Dual Band
and Triple Band Compact Microstrip Antenna

A microstrip line of width GL in the bottom plane acts as a rod and it gets associated with
the magnetic field when excited by a wave port. The microstrip line rod then excites the
top ring resonator with the fringing fields via the dielectric material. The structure is
modeled using a dielectric substrate of relative permittivity εu-c = 3 and height h2 = 1.52
mm. In the proposed unit-cell structure, a wave port is defined in the z-direction, electric
polarization in the x-direction and magnetic polarization in the y-direction is assigned to
solve by frequency domain solver of Microwave Studio Suite of Computer Simulation
Tool (CST-MWS v.16) [82]. Fig. 4.4 shows the simulated S-parameters of the unit-cell
structure.

FIGURE 4.4 Simulated scattering response of the unit-cell with inset image of boundary
condition.

It is important to indicate that desired scattering parameters (S11 and S21) can be obtained
by increasing or decreasing the distance gt with respect to the thin connecting slit between
the S-shaped resonator and rectangular stub. The graph of S11 and S21 shown in Fig. 4.5
validates the dependence of the unit-cell frequency response on gt. A frequency cross-over
can be observed at 2.137 GHz which indicates the frequency at which negative response is
likely to be exhibited by the unit-cell. The material medium parameters are verified by
substituting S-parameters into the Kramers-Kroing relations as discussed in Chapter 1.

68
 Design of Metamaterial Unit-Cell Stacked TL-DR
Antenna

(a) (b)
FIGURE 4.5 Plot showing the effect of gt on unit-cell (a) S11 and (b) S21.

From Fig. 4.6(a), permeability is positive (μ > 0) in the simulated spectrum. In Fig. 4.6(b),
the permittivity is more negative (ε < 0) in the band of 2.1 GHz to 2.5 GHz. The ε < 0
indicates the location of electric flux into the unit-cell which decides the transmission
frequencies allowed to pass through into the free-space.

(a) (b)

(c) (d)
FIGURE 4.6 Retrieved plots of the unit-cell characteristic parameters. (a) Permeability, (b) Permittivity, (c)
Wave impedance (z), and (d) Refractive index (R.I).

69
 Development of Metamaterial Based Dual Band
and Triple Band Compact Microstrip Antenna

As results of μ > 0 and ε < 0, the refractive index (R.I) appears near the zero line in
Fig. 4.6(d) and the material tend to behave as a near-zero metamaterial. The plot of wave
impedance (z) show in Fig. 4.6(c) indicates that EM propagation through the metamaterial
unit-cell will experience lower resistance. These results confirm the existence of unusual
properties in the unit-cell.

4.2.2 Prototype Antenna Configuration

The DR and unit-cell are placed in stack arrangement over the SRR antenna in order to
take advantage of frequency shifting by the resonance condition of DR and a new band
generation with the help of unit-cell excitation via the DR. Fig. 4.7 depicts the antenna
configuration employing DR and metamaterial unit-cell altogether for multiband
operations.

FIGURE 4.7 Perspective view of SRR-based antenna loaded with a stack of TL-DR and metamaterial unit-
cell.

It important to explain that in this type of microwave technique, it is essential that

frequency band after loading the DR in the antenna should lie in the transition band of
unit-cell where crossover between S11 and S21 takes place. The transverse electric mode of
DR easily excites the mounted unit-cell structure and propagates its resonant band. This
means either different unit-cell can be stacked or a unit-cell with wideband negative R.I
can be modeled to generate diverse frequency bands the LTE spectrum.

70
 Parametric Analysis

4.3 Parametric Analysis

The numerically solved patch antenna dimensions often undergo one or more
increment/decrement offset from the reference value in order to meet the design goals. In
this regard, an optimization of antenna dimensions is carried out using time-domain solver
of CST-MWS. Fig. 4.8(a) shows the effect of increasing length of Pw1 which defines the
radiating side.

(a) (b)

(c) (d)
FIGURE 4.8 Plot of parametric variation on geometrical dimensions. (a) PL1, (b) Pw1, (c) S1, (d) S2.

It is found that the increase in the length of Pw1 alters the frequency and magnitude of S11
from -24.8 dB at 2.46 GHz to -12.20 dB at 2.41 GHz. This happened due to the reduction
in the slot area in the patch corners which inhibits the circular flow of surface current
around the SRR and increases the current density along the radiation edges of the patch. A
current maximum in the radiating side tends to suppress the resonant mode. A similar
effect can be seen in Fig. 8b when the dimension of PL1 increases. In the SRR patch, the E-
shape stub has been selectively connected to the non-radiating side to modify the overall
electrical length and hence the resonant modes.

71
 Development of Metamaterial Based Dual Band
and Triple Band Compact Microstrip Antenna

From the response plotted in Fig. 4.8(c), it is found that the length S1 of the E-shape stub
dominantly control the frequency band. S1 = 1 mm drastically inhibits the resonance
condition in the antenna. However, increasing the value of S1 helps to obtain a resonant
band in the LTE spectrum. In case of the S3 arm of the E-shape stub, the fundamental
resonance is weak at 3 mm length and low Q-factor affect the higher order band near 3
GHz. Increase in the length of S3 up to 6 mm gradually improves the dual band resonance
as shown in Fig 4.8(d). Beyond 6 mm, the impedance matching of patch reduces and the
magnitude of S11 decreases to -11.29 dB with minor frequency shift towards the lower side
of the microwave band. Thus, Pw1 = 16 mm, PL1 = 8 mm, S1 = 4 mm and S2 = 6 mm were
chosen better impedance matching and high-Q performance.

4.4 Antenna Fabrication and Measurement Setup

The proposed SRR-loaded patch antenna is fabricated on FR-4 substrate of relative

permittivity 4.3 and loss tangent tanδ = 0.017 using photolithography process. A 50 Ω sub-
miniature version-A (SMA) connected is soldered to the edge of microstrip feed line. Fig.
4.9(a) shows the image of the fabricated antenna. For realizing the DRA, the top and
bottom side metallization of RO3010 substrate of relative permittivity 10.2 and tanδ =
0.004 was chemically etched and used as TL-DR. Thereafter, unit-cell geometry is
patterned on RO3003 substrate relative permittivity εr = 3 and tanδ = 0.0013. The TL-DR
and unit-cell were first bonded together using silicone adhesive and then to the center of
the patch as shown in Fig. 4.9(b). The reduced ground plane of the antenna is shown in
Fig. 4.9(c).

(a) (b) (c)

FIGURE 4.9 Image of the fabricated antenna. (a) SRR-based antenna without TL-DR and unit-cell. (b)
SRR-based antenna loaded with a stack of TL-DR and metamaterial unit-cell, (c) Bottom ground plane.

72
 Antenna Fabrication and Measurement Setup

The measurement setup for validating the antenna performance is shown in Fig. 4.10.
Agilent Vector Network Analyzer (VNA) is used for the measurement of S-parameters.
Before recording the data, the input power level was set to 0 dBm, intermediate frequency
bandwidth to 10 KHz, and 201 frequency sweep points were taken. Next, the VNA was
calibrated for open-short-load conditions and thereafter the fabricated antenna was
connected to the VNA through 50 Ω coaxial cable for measurements.

FIGURE 4.10 Image of the antenna measurement setup.

4.5 Discussion on Results

In Fig. 4.11, the simulated S-parameter of the SRR-based patch is plotted and the response
after loading TL-DR and metamaterial unit-cell is compared. The impedance bandwidth of
the bare antenna is almost identical at the center frequency of 2.031 GHz and 2.461 GHz.
The TL-DR placed above the patch antenna gets coupled to the SRR structure through
magnetic current parallel to the plane of TL-DR. This excites the dielectric resonator and
new resonant modes are created at 2.217 GHz and 2.28 GHz.

73
 Development of Metamaterial Based Dual Band
and Triple Band Compact Microstrip Antenna

FIGURE 4.11 Comparison of the antenna reflection coefficient after loading TL-DR and unit-cell.

Subsequently, a metamaterial unit-cell above the TL-DR in stack configuration

resonates at 2.145 GHz and triple bands are obtained in the LTE spectrum above 2 GHz.
The measured S11 of bare SRR-based patch antenna, TL-DR loaded antenna, and
metamaterial unit-cell stacked on TL-DR antenna are plotted in Fig. 4.12. A close
agreement between simulated and practical results depicts proper functioning of the
fabricated antenna.

(a) (b)

(c)
FIGURE 4.12 Comparison of the measured and simulated plot of S11. (a) Bare antenna, (b) TL-DR loaded
antenna, (c) Unit-cell loaded on TL-DRA.

74
 Discussion on Results

However, the lossy nature of the FR-4 substrate affects the signal to radiation conversion
efficiency of the antenna which lowers the |S11| and introduces ripples. Unwanted ripples
in the plot of S11 mainly occurs from the slight mismatch caused because of the single feed
line and the patch which occurs from the presence of minor discontinuities in the
fabricated feed line, presence of solder bridges at the feed pin, and oscillatory behavior of
microstrip line. In Table 4.2, the simulated and measured data of the proposed antenna is
summarized.

Table 4.2 List of the antenna resonant frequencies.

Frequency SRR-Based TL-DR loaded Unit-cell Loaded TL-

(GHz) Patch Patch DDRA
Simulated 2.031, 2.461 2.217, 2.28 2.145, 2.247, 2.304
Measured 2.11, 2.665 2.204, 2.292 2.134, 2.259, 2.364

The mechanism of notched resonances in the antenna is studied by simulating the surface
current distributions. In Fig. 4.13, the current distribution at different resonant frequencies
is plotted. At the fundamental frequency of patch, strong current flow circulates in the
SRR and patch metallization as shown in Fig. 4.13(a).

(a) (b)

(c) (d) (e)

FIGURE 4.13 Simulated current distributions of the antenna at different resonant frequencies. (a) 2.031
GHz, (b) 2.461 GHz, (c) 2.145 GHz, (d) 2.247 GHz, (e) 2.304 GHz.

75
 Development of Metamaterial Based Dual Band
and Triple Band Compact Microstrip Antenna

The higher order mode at 2.461 GHz of the bare patch is solely due to the effect of current
concentration in the SRR. On the other hand, weak coupling of RF energy from the TL-
DR results in sparse current flow in the top plane of the unit-cell as shown in Fig. 4.13(c).
The radiating frequencies from the TL-DRA are well coupled. The current induced in the
S-shape resonator is evident which can be seen in the current plot represented by Fig.
4.13(d-e). Moreover, the circular path of the current still exists in the plane of the patch
antenna.
The simulated radiation patterns of the antenna at different resonant frequencies in E-
plane and H-plane are compared in Fig. 4.14. It can be observed that broadside radiation
pattern has been obtained at 2.031 GHz and 2.304 GHz. At 2.29 GHz, the field intensity in
E-plane is relatively lower because of the less reflection magnitude and impedance match.
The measured wideband gain of the antenna in the frequency span of 1 GHz to 4 GHz was
found in close match with the simulated data. The plot of gain performance is shown in
Fig. 4.15. The peak gain of the antenna was found to be 3.97 dBi.

(a) (b) (c)

FIGURE 4.14 Simulated radiation pattern in E-plane and H-plane. (a) 2.031 GHz, (b) 2.29 GHz, (c) 2.304
GHz.

FIGURE 4.15 Comparison of simulated and measured wideband gain of the antenna.

76
 Performance Analysis of TL-DR Metamaterial Loaded
LTE Antenna

It is possible to further enhance the overall gain of the antenna by using a metamaterial
superstrate with negative refractive index (R.I) at desired frequency band. The negative R.I
will enable selective propagation of electromagnetic radiation from the antenna through its
plane. The radiation efficiency of the antenna at 2.145 GHz, 2.247 GHz, 2.304 GHz was
found to be 47 %, 62.5 %, and 85 % respectively and plotted in Fig. 4.16.

FIGURE 4.16 Plot of the antenna radiation efficiency.

4.6 Performance Analysis of TL-DR Metamaterial Loaded LTE Antenna

The overall features of the antenna are further analysed by comparing its size and resonant
bands with the previously reported antennas. It is found that the proposed antenna
designing technique is suitable for multi-band operation along with higher miniaturization
in the size.

Table 4.3 Comparison of the proposed TL-DRA with other reported antenna design techniques for LTE
application.

Reference Technique Substrate Size Bands for Total

Permittivity (mm2) LTE Bandwidth
(MHz)
Ref [59] Tunable DR Taconic (ɛr = 3.2) 20×36 4 200
array antenna
Ref [45] Filtering patch (ɛr = 2.2) 80×100 2 -
antenna
Ref [47] SRR-Loaded FR-4 (ɛr = 4.4) 40×40 2 ~ 850
MIMO antenna
Ref [72] Metamaterial Teflon (ɛr = 2.1) 60×60 2 2100
superstrate DRA
Ref [44] Hybrid DRA FR-4 (ɛr = 4.4) 70×50 3 629
This work Unit-cell loaded FR-4 (ɛr = 4.4) 20×25 3 105
TL-DRA

77
 Development of Metamaterial Based Dual Band
and Triple Band Compact Microstrip Antenna

Summary
A triple band antenna using stack configuration of thin-layer dielectric and metamaterial
unit-cell is realized. It was found that superstrating or stacking the unit-cell need it's cross-
over frequency at which S11 and S12 coincide in accordance with the excitation frequency
range from the dielectric resonator. Thus, with the help of stacked TL-DR and unit-cells,
multiple narrow bands can be excited in the antenna. The other frequency bands of the
LTE spectrum can also be obtained by stacking different metamaterial unit-cell structures
with multi-band operation. In such antenna configuration, the transverse electromagnetic
wave propagation will excite MTM unit-cells in a progressive manner. As a result,
different MTM structures tend to resonate and desired LTE bands can be obtained. The
measured results show close agreement with the simulation data. Moreover, the proposed
antenna is highly compact in comparison with the other reported LTE antenna and the size
can be further reduced by engineering the design of metamaterial loaded in the antenna
radiation plane.

78
 Antenna Design

CHAPTER - 5

Development of Metamaterial-Based Compact

Frequency Reconfigurable Microstrip Antenna

In this Chapter, a compact planar antenna is presented to overcome the practical

difficulties in LTE band antenna technology as discussed in Chapter 1 , Section Research
Gap. The design of the antenna is inspired by printed Yagi-Uda antenna. Defective ground
plane geometry in the shape of a comb is created for the generation of multiple frequency
bands. Additionally, Split Ring Resonator (SRR) is placed at a suitable position where
outer and inner rings are electrically connected to the patch for achieving frequency
diversity in the desired spectrum. Unlike MTM-based antenna discussed in Chapter 3 and
Chapter 4 where either MTM was directly loaded in the radiating patch or coupled through
the DR layer, SRR has been connected to the main radiating patch which can be
electrically switched ON-OFF for desired frequency bands. The frequency response, gain,
and radiation pattern of the antenna was experimentally tested to validate its performance
compatibility with LTE and WLAN/WiMAX applications.

5.1 Antenna Design

The antenna design for LTE band operation is initiated with the modeling of a circular
patch and its radius a was calculated using [74],

1.8412 × 𝑐
𝑓𝑟 = (5.1)
2𝜋𝑎 √𝜀

/
where, 𝑎 = 𝑎 1 + 𝑙𝑛 + 1.7726 (5.2)

79
 Development of Metamaterial-Based Compact
Frequency Reconfigurable Microstrip Antenna

𝐹
𝑎= /
2ℎ 𝜋𝐹 (5.3)
1 + 𝜋𝜀 𝐹 𝑙𝑛 + 1.7726
2ℎ

In Eq. (1)-(3), fr – denotes the resonant frequency for transverse magnetic (TM10) mode,
a – is the radius of the patch, F – is the inset feed line length, h – is the height of the
substrate, and 𝜀 – is the relative permittivity of the substrate. The assumption of fr = 5.75
GHz, 𝜀 = 4.4 and h = 1.55 mm leads to the numerical value of a ~ 14 mm. Initially, a
higher resonant frequency was selected to achieve a smaller dimension of the antenna.
Later, resonant bands in the LTE spectrum were obtained by trimming top plane and
ground plane of the circular patch and incorporating split ring resonator in close vicinity to
the patch. The presence of multiple discontinuities in the top metallization helps in
perturbing the flow of surface current to create multiple resonances. Therefore, multiple
radial stubs in ± x-direction and ± y-direction were evolved from the circular patch as
shown in Fig. 5.1(a). The square slot was created at the center of the patch to direct RF
energy in different directions. The orientation of radial stubs resembles the Yagi-Uda
antenna configuration and its operation. It is important to highlight that conventional
microstrip patch antenna with defective ground emits EM wave radiation from top plane as
well as bottom plane due to the presence of partial ground.

(a) (b)
FIGURE 5.1 Antenna Layout. (a) Top view of the radiating plane, (b) Bottom view of the ground plane;
where L = W = 45, h = 1.55, f1 = 3.318, f2 =10, C1 = 7, C2 = 8, Sw = 5.5, Sl = 11, g = 4, u1 = 1.5, r1 = 7, r2 =
5.9, r3 = 10, r4 = 14, M1 = 25, M2 = 40, M3 = M4 = 10 S0 = S1 = 5, P = 17 (all dimension are in mm)

80
 Antenna Fabrication and Experimental Setup

As a result, lower pattern directivity is obtained. However, by using Yagi-Uda

configuration in the top plane, improvement in the directivity of the radiation pattern is
possible. Furthermore, radiation in the broadside direction will also take place because of
the director elements of Yagi-Uda inside the circular patch. Therefore, beam coverage of
the antenna can be enhanced. A stepped impedance feed line of width 1.5 mm and
characteristic impedance Z0 = 190 Ω was connected to the antenna for maximum power
transfer. The characteristic impedance of 190 Ω between 50 Ω microstrip line and the
antenna structure was selected to improve the matching of the source to load for lower
reflection. The ground plane is partially corrugated with depth (P) and ridge (S1) as shown
in Fig. 5.1(b), and it makes the ground defective which is responsible for selective
frequency bandgap [44]. Moreover, the SRR structure is placed near to the patch. The
excitation to SRR from x-direction and y-direction using Diode D1 and D2 respectively,
changes its electric field (𝐸⃗ ) polarization opposite to the direction of excitation. Therefore,
LC-resonance of the SRR modifies the electrical length of the patch and different
resonances are obtained.

5.2 Antenna Fabrication and Experimental Setup

The antenna was fabricated using FR-4 substrate of relative permittivity εr = 4.4 and
loss tangent tan δ = 0.07 as shown in Fig. 5.2. The layout of the antenna was printed using
photolithography, and unwanted metallization was removed with chemical etching. Two
PIN diodes (D1 and D2) of model BAP64 from NXP semiconductors are connected the
top plane patch and SRR rings.

FIGURE 5.2 Image of the fabricated antenna.

81
 Development of Metamaterial-Based Compact
Frequency Reconfigurable Microstrip Antenna

The PIN diode D1 and D2 were operated in forward bias using DC bias voltage
(Vbias = 0.8). Surface mount RF choke and DC blocking lumped elements from Murata
were utilized in the biasing network. In the experimental setup, Vector Network Analyzer
(Agilent-NN23A; 4 GHz) was calibrated with Open-Short-Load in the span of 0.5-4 GHz
with 201 sweep points and an intermediate frequency bandwidth of 2.5 KHz. A 50 Ω
coaxial cable was connected to the antenna, and proper DC supply was given, as shown in
Fig. 5.3(a). The circuit represented in Fig. 5.3(b) was used for isolating RF and DC
signals. A capacitor (Cblock = 220 pF) soldered between the gap in the feed line blocks DC
signal from the supply source. Lchoke = 47 nH and 68 nH acts as RF choke and it provides
higher impedance (~ 250 Ω - 1 KΩ) in the selected frequency range, i.e., 0.5-4 GHz.

(a)
(b)
FIGURE 5.3 (a) Experimental setup for antenna measurement, (b) RF/DC isolation network configuration
(Rs = 5 Ω & Ls = 4 nH are series resistance and inductance of PIN diode in forward bias respectively)

5.3 Results and Discussion

The permittivity and permeability response of the metamaterial split ring resonator (SRR)
was extracted using the Kramer’s-Kroing formulation as Discussed in Section 1.5.6. The
plot of permittivity shows negative epsilon at 3.5 GHz. This mean, at this frequency, the
formation electric flux within the substrate of the antenna is not strong and more loosely
bounded EM field can be created for efficient radiation mechanism. The response of
permeability remains positive throughout the frequency range from 1-4 GHz. Thus, the
designed MTM acts as pure epsilon negative (ENG) material. The result of permittivity
and permeability is plotted in Fig. 5.4. The reflection coefficient (S11) of the antenna in
different biasing states of the two PIN diodes was simulated using Time-Domain solver of
Computer Simulation Tool (Microwave Studio Suite).

82
 Results and Discussion

(a) (b)
FIGURE 5.4 Material response of SRR. (a) Plot of permittivity, (b) Plot of permeability

As shown in Fig. 5.5(a), the antenna offers pentaband resonance in all diode OFF
conditions, and the graph of S11 changes to dual-band resonance for D1-ON, quad-band
resonance for D2-ON, and again dual-band resonance for D1-D2-ON conditions.

(a) (b)

(c) (d)
FIGURE 5.5 Comparative plot of antenna reflection coefficient (S11) for diode ON-OFF Conditions. (a) D1-
D2 OFF, (b) D1-ON, (c) D2-ON, (d) D1-D2-OFF

83
 Development of Metamaterial-Based Compact
Frequency Reconfigurable Microstrip Antenna

In parallel to this, the measured results are found in coherence with the simulated data, as
shown in Fig. 5.5. A slight deviation in the frequency response for D2-ON and shift in the
second band (2.47-2.83 GHz) of D1-ON to 1.18-1.23 GHz exist because of the high-
frequency pickup by the biasing wires. However, by proper shielding of bias wires or
replacing them with high impedance microstrip lines can solve the problem to a greater
extent. It should also be noted that the presence of parasitic capacitance from the active
element often creates a slight impedance mismatch with respect to the 50 Ω source. As a
result, a minor shift in the measured S11 was observed for ON-state of D1 and D2.
Moreover, a weaker resonance was also seen in the antenna at the desired operating
frequency. Thus, a tunable matching network would be an essential need for an active
antenna. In Table 5.1, the simulated and measured frequencies of resonance are
summarized. It is evident from the surface current distribution profile (Fig. 5.6) that RF
energy is directed in both end-fire and broadside direction. In D1-D2-OFF state, the
resonance at 0.96 GHz is solely from the circular patch where the surface current is not
localized in all arms of the patch.

Table 5.1 Measured and simulated resonant bands of the proposed antenna

Unit-GHz Diode OFF D1-ON D2 ON D1-D2 ON

Simulated 0.68-1.2, 0.6-1.03, 1.47-2.29, 0.7-1.11,
1.3-1.5, 2.47-2.83, 2.57-2.9, 2.4-2.8,
1.68-1.82, 3.74-4 3.1-3.3, 3.75-4
1.94-2.5, 3.76-4
2.7-2.84
Measured 1.14-1.23, 1.18-1.23, 0.9-1, 1.25-1.27,
1.49-1.6, 3.1-4 1.48-1.76, 1.81-1.86,
1.7-1.76, 2.21-2.3, 3-4
1.9-1.93, 2.72-4
2.11-2.25,
2.62-2.79,
2.93-4

In D2-ON state, the outer ring of SRR and left-hand side radiating plane of the circular
patch contributes to majority of the surface current at 2.36 GHz. Thus, radiation takes from
the area of current minima. In D1-D2-ON state, the antenna surface current is uniformly
distributed in SRR and patch, and an eight-pattern radiation pattern is obtained.

84
 Results and Discussion

(a) (b) (c)

FIGURE 5.6 Simulated surface current distribution of the antenna at (a) 0.96 GHz, (b) 2.63 GHz, (c) 3.22
GHz

The same can be observed in the 𝐸⃗ -plane radiation pattern plot presented in Fig. 5.7. The
radiation pattern exists in the direction where the surface current is minimum. The current
minima create a high-impedance surface area in the antenna top plane for radiation
mechanism.

(a) (b) (c)

FIGURE 5.7 Simulated E-plane radiation pattern of the antenna at (a) 0.96 GHz, (b) 2.63 GHz, (c) 3.22
GHz.

The antenna gain was measured by fabricating two similar antenna structures and then
applying Friis-Transmission theory, that is applicable as per [83], as

𝜆 𝑃𝐺 (5.4)
𝑃 =
(4𝜋𝑑)

85
 Development of Metamaterial-Based Compact
Frequency Reconfigurable Microstrip Antenna

Where, Pr – is the power measured from the receiving antenna, Pt – is the power of
transmitting antenna, G denotes the antenna gain, d – is the distance kept between the two
antennas, and 𝜆 – is the operating wavelength. The transmitting and receiving antenna
were placed at a known distance, and the one of the antennas was excited using a
microwave power source at a selected frequency. The receiving antenna was connected to
the spectrum analyzer for measuring the received power (Pt). The gain was then
calculated. The measured gain in the band of 1-1.5 GHz, 2-2.5 GHz, and 2.6-2.9 GHz is
around 2.5 dBi, 3.25 dBi, and 3.72 dBi respectively, and it is represented in Fig. 5.8.

FIGURE 5.8 Measured plot of antenna gain in comparison to the simulated.

The performance of the proposed metamaterial based circular patch antenna in Yagi-Uda
configuration is in fair in comparison with other reported LTE band antennas, as shown in
Table 5.2.

86
 Results and Discussion

Table 5.2 Performance comparison of the proposed antenna with other reported LTE band antennas.

Ref. Technique Substrate Size (mm2) No. of Bands Peak Gain (dBi)
[48] Reconfigurable Taconic 20 × 36 4 1.9
DRA (𝜀 = 3.2)
[84] Multiple monopole FR-4 25 × 76 2 4.4
antenna (𝜀 = 4.4)
[85] MIMO Slotted FR-4 180 × 90 2 12.5
antenna with (𝜀 = 4.4)
Metamaterial
radome
[49] Tunable stubs with FR-4 33 × 16 6 3.46
truncated ground (𝜀 = 4.5)
[51] Metamaterial ring FR-4 42 × 32 2 3.69
resonator coupled (𝜀 = 4.3)
square strip
This SRR coupled Patch FR-4 45 × 45 12 3.72
work (𝜀 = 4.3)

It is evident that a circular microstrip patch inspired by Yagi-Uda design, DGS and SRR
mode switching is useful to multiband operation and offers compact size. The switching of
frequency bands using SRR demonstrates that frequency diversity can be implemented
using the proposed antenna.

Summary
Antenna design using DSG and SRR is found useful in generating multi-resonance modes
for frequency diversity applications. The switching of SRR using DC bias network also
allows steering of the radiation pattern, which addresses the challenge of beam coverage
using a compact antenna. The performance of the realized antenna is in a close match with
the simulation results.

87
 Conclusions and Future Scope

CHAPTER - 6

Conclusions and Future Scope

6.1 Conclusions

In this research work, various microwave antenna design techniques are developed and
analyzed to improve the resonance modes in a conventional microstrip patch antenna for
multi-Band LTE wireless operation. The prototype antenna design explores method of
loading CSRR, connected ring-resonator, and superstrate layer techniques. The following
are some of key observations and conclusion extracted from the simulation, testing and
measurement of the proposed antenna designs;

 A compact and low-profile MSA can be realized by first modeling the structure under
the assumption of higher cut-off frequency and later, the frequency of resonance can
be tune to lower microwave band connection stubs in the non-radiating plane of the
antenna.

 The resonance of the MSA can be directly enhanced by loading a CSRR structure
inside the patch metallization of the antenna.

 Along with the fundamental mode (TMmn) of MSA, second order mode can be
excited by loading the connected ring-resonant structure at the center of the patch. This
helps in obtaining dual-band antenna operation.

 More importantly, the isolation between the two or more frequency bands can be
controlled by placing a dielectric resonant above the patch. It is observed that
additional resonant mode can be excited in the MSA by designing a MTM unit-cell for

88
 Conclusions

selective frequency range and loaded a superstrate in the close vicinity to the antenna
coupled by dielectric resonator.

 Furthermore, loading of MTM unit-cell structure near to the patch and selective
switching its connection with different metallization areas of the patch is found
effective for achieving multiple frequency bands and broad side radiation pattern.

 The proposed metamaterial-based MSA designs are modeled, simulated, fabricated and
practically tested for validation of the design and proper functioning for LTE
applications.

6.2 Future Scope

The development of various metamaterial-based antennas shows potential applications in

controlling the resonant bands. It also allows several different microwave techniques to co-
exist in the design topology of metamaterial-based antenna. This provided numerous
opportunities in the development of compact multi-band LTE antennas. Based on the study
and analysis of the LTE band antennas reported in this thesis, some of the identified
challenges which still needs further research are;

 Gain enhancement techniques and maintaining the form factor of the antenna.
 Improvement in the RF and DC isolation network by using printed impedance
controlled line and suitable bias network for minimum signal losses.
 Reconfigurable radiation pattern in the antenna for addressing spectrum sensing in
broad range.
 Implementation of the LTE antenna on flexible substrate for specific wireless sensing
applications like IOT communication.

89
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94
 Publications

Publicatons

Journal Articles
[1] Tushar P. Dave, Jagdish M. Rathod, “A thin-layer dielectric and metamaterial unit-cell
stack loaded miniaturized SRR-based antenna for triple narrow band 4G-LTE
applications,” International Journal of RF Microwave Computer Aided Eng., vol. 29, no.
5, 2018. (SCI Indexed)

[2] Tushar P. Dave, Nita T. Dave, Jagdish M. Rathod, “Compact Microstrip Patch Antenna
Using Metamaterials to Perform as Narrow Band LTE Receiver Antenna”, Journal of
Communication Engineering & Systems, vol. 8, no. 3, pp. 24-29, 2018 (UGC Approved)

[3] Tushar P. Dave, Jagdish M. Rathod “Analysis of improvement in bandwidth and

multiband behavior of metamaterial based microstrip patch antenna”, International
Journal of Recent Trends in Electronics and Communication (IJRTEC), vol. 1, issue. 1,
August-2016.

[4] Tushar P. Dave, Jagdish M. Rathod “Compact frequency diversity DGS antenna with
SRR mode switching for LTE and WLAN/WiMAX wireless systems”, Wireless Personal
Communications, (2020). https://doi.org/10.1007/s11277-020-07035-5 (SCI Indexed)

Patent
[1] Dave T. P., Rathod J. M., Kashyap S. (2019) Microstrip Couplers Using Artificial
Dielectric Substrate, Indian Patent 201921031330 A.

Conference Article
[1] Tushar P. Dave, Jagdish M. Rathod, “Comparison of Spiral and Metamaterial Inspired
Patch Antenna For 4G LTE Applications”, International Conference on Emerging Trends
in Engineering, Science and Technology, CSPIT, Changa, Gujarat, pp. 25-29, Dec-2018.

95