sensors

Article
Vivaldi Antenna Arrays Feed by Frequency-Independent Phase
Shifter for High Directivity and Gain Used in Microwave
Sensing and Communication Applications
Jiwan Ghimire 1 , Feyisa Debo Diba 1,2 , Ji-Hoon Kim 3 and Dong-You Choi 1, *

1 Department of Information and Communication Engineering, Chosun University, 375 Susuk-dong, Dong-gu,
Gwangju 501-759, Korea; ghimire@chosun.kr (J.G.); feyisa@chosun.ac.kr (F.D.D.)
2 Department of Electronics and Communication Engineering, Adama Science and Technology University,
Adama 1888, Ethiopia
3 Department of Mechanical Engineering, Chosun University, 375 Susuk-dong, Dong-gu,
Gwangju 501-759, Korea; kjh@chosun.ac.kr
* Correspondence: dychoi@chosun.ac.kr

Abstract: This paper describes a novel feed system for compact, wideband, high gain six-slot Vivaldi
antenna arrays on a single substrate layer using a unique combination of power splitters based on
binary T-junction power splitter topology, frequency-independent phase shifter, and a T-branch. The
proposed antenna system consists of six Vivaldi antennas, three on the left, and three on the right arm.





 Each arm connects with T-junction power divider splitter topology, given that the right arm is linked
through a frequency-independent phase shifter. Phase shifters ensure that the beam is symmetrical
Citation: Ghimire, J.; Diba, F.D.; Kim,
without splitting in a radiating plane so that highly directive radiation patterns occur. The optimal
J.-H.; Choi, D.-Y. Vivaldi Antenna
return losses (S-parameters) are well enriched by reforming Vivaldi’s feeding arms and optimizing
Arrays Feed by Frequency-
Independent Phase Shifter for High
Vivaldi slots and feeds. A novel feature of our design is that the antenna exhibits the arrangements of
Directivity and Gain Used in a T-junction power splitter with an out-of-phase feeding mechanism in one of the arms, followed
Microwave Sensing and by a T-branching feeding to even arrays of proper Vivaldi antenna arrangement contributing high
Communication Applications. Sensors realized gain and front-to-back ratio up to 14.12 dBi and 23.23 dB respectively applicable for not
2021, 21, 6091. https://doi.org/ only ultra-wideband (UWB) application, also for sensing and position detecting. The high directivity
10.3390/s21186091 over the entire UWB frequency band in both higher and lower frequency ranges ensures that the
antenna can be used in microwave through-wall imaging along with resolution imaging for ground
Academic Editors: penetration radar (GPR) applications. The fabricated antenna parameters are in close agreement with
Mahdi Moosazadeh and
the simulated and measured results and are deployed for the detection of targets inside the voids of
Andrey Miroshnichenko
the concrete brick.

Received: 18 August 2021

Keywords: ultra-wideband antenna; Vivaldi antenna array; power dividers; ground penetra-
Accepted: 8 September 2021
Published: 11 September 2021
tion radar

Publisher’s Note: MDPI stays neutral

with regard to jurisdictional claims in
1. Introduction
published maps and institutional affil-
iations. In a modern context, wireless devices have seen immense growth in the field of radar,
radio astronomy, vehicular, high-speed communication, surveillance, and UWB imaging
systems, making communication tools and techniques easily accessible and fast [1–3]. Due
to certain benefits over conventional antennas such as their low profile, compactness, cheap
Copyright: © 2021 by the authors.
costs, planar configuration, suitability for arrays, and ease of integration [4,5], a microstrip
Licensee MDPI, Basel, Switzerland.
antenna has been considered as a potential alternative for detecting landmines [6], through-
This article is an open access article
wall imaging [7], biomedical imaging [8], breathing and heart rate detection [9,10], motion
distributed under the terms and and gesture detection [11], and a two-dimensional (2D) imaging of concrete block [12].
conditions of the Creative Commons Most of the UWB antennas existing in the literature had to overcome high propagation loss
Attribution (CC BY) license (https:// within microwave bands confined by a specific transmit spectrum. For microwave imaging
creativecommons.org/licenses/by/ of the buried targets, voids, and hidden cracks in concrete-based construction material and
4.0/). composite structure, the antenna with a wide frequency band, stable unidirectional end-fire

Sensors 2021, 21, 6091. https://doi.org/10.3390/s21186091 https://www.mdpi.com/journal/sensors

Sensors 2021, 21, 6091 2 of 17

radiation character, and a high directive gain is preferable to get desired penetration depth,
especially within high lossy material. The traditional destructive methods for checking
are time-consuming, dangerous, and expensive, so the use of non-destructive testing
(NDT) is preferred for subsurface monitoring where a high depth and range resolution
tracking antenna is required. Different antenna designs are implemented for UWB GPR
systems such as dipole [13], bow-tie antenna [14,15], the spiral antenna [16], tapered slot
antenna [17,18], and TEM horn antenna [19,20] depending upon whether the application
demand requires high penetration depth with low resolution or vice-versa. The GPR
antennas have either low gain or are bandwidth limited, can exhibit late-time ringing while
transmitting short-time pulse, and are bulky [21]. The bandwidth of the antenna plays
an important role in selecting the application requirement for a high depth penetration
with low resolution or vice-versa. The choice between penetration depth and resolution
is determined by the antenna transmission frequency, so a high bandwidth antenna is
required for the desired frequency selection. Ground-coupled and air-coupled antennas
are commonly employed for GPR assessment, however, for detection of layer interference
such as voids and targets, air-coupled antennas are preferable [22]. Because airborne
GPR antennas still have severe disputes with broad bandwidth, narrow beam width, light
weight, and high gain, these issues can be addressed by deploying a planner antenna
like Vivaldi [23]. The Vivaldi antenna and its array could be suitable for microwave
communication for their wide bandwidth, easy fabrication, high efficiencies and directivity,
simple structure, consistency in E-H plane, and lightweight characteristics [24]. However,
the conventional Vivaldi antenna in an array undergoes some weaknesses, like slanting
beam and changeable in directivity and gain at higher frequency [25,26]. To overcome
these problems, several methods have been proposed for achieving higher gain by using
methods like employing dielectric lens [26], metamaterial lens [27,28], parasitic elliptical
patch [29], and electromagnetic bandgap (EBG) [30]. Other methods include use of profiled
dielectric directors [31,32], modification of radiating arm slots [33], array structure [34,35],
negative index material (NIM) [36], zero-index material (ZIM) [37] and dielectric slab [38].
However, developing efficient gain enhancement with different existing techniques in an
area of limited size within a planner antenna system is still a costly, complicated, and
difficult task. Array structures are generally used to achieve high gain with inevitable
problems of the side lobes and mutual coupling in array design. The solutions to solve
the ambiguity between the side lobes and the mutual coupling of the corresponding
radiating elements arise while tuning the antenna’s performance. High coupling causes
significant degradation of antenna gain and bandwidth while increasing the side lobes can
cause a decrease in antenna radiation performance. The wideband performance can be
achieved if the array elements are electrically connected to adjacent elements [39], whereas
by increasing the aperture size of the radiating patch, the lower frequency response will
be satisfactory along with the improvement in the radiation characteristics [40,41]. The
performance of an antenna, particularly in Vivaldi arrays, is determined by its feeding
network. The size of the feeding structures should also be taken into consideration while
designing arrays of these antenna as most of the valuable space is occupied by the feeding
network portion. Different kinds of feeding array structures are proposed in various
studies for the overall increase in the antenna performance either in terms of optimizing
insertion loss or increasing the frequency bandwidth (e.g., SIW binary splitter, SIW power
dividers, grounded coplanar waveguide (GCPW), T-junction power divider, combined
T-type and Y-type dividers, four-way SIW power divider, a two-way power divider, and
a 1-to-8 power divider network [42–50]), making power divider the most popular ones
among all types of feeding network. However, most of the power dividers are in-phase
power dividers limited to feed a certain number of antenna arrays providing the same
amplitude and phase at the output which, when introduced at the system, provide the
splitting of the beam at a higher frequency. In [51], the Vivaldi antenna array is fed by
1-to-4 non-uniform T-junction power dividers, limiting the number of antennas to four
with a maximum measured gain of 11.3 dBi. In [47], a 1-to-8 power divider feed network
Sensors 2021, 21, 6091 3 of 17

is used for connecting to eight antipodal left and right arms for 5G mm wave application.
Two distinct types of Vivaldi antenna for see-through-wall applications were developed
in [52] for UWB using 8 × 1 and 16 × 1 Wilkinson power divider with a gain of more
than 12 dBi and 13 dBi, respectively. What we see from the reported design of the power
divider feeding network is that we can arrange 2n number of antenna arrays for n number
of rows of branching junction limiting the antenna to 2, 4, 8, 16... number of array sequence.
Getting a 6, 10, 12... even antenna array combination sequence is impossible, and a lot of
space is occupied by the feeding network within the array of antenna using this divider
topology. A V-shaped even mode power divider with T- junctions is presented in [53] that
limits the antenna array to four, constraining the maximum realized gain to approximately
11 dBi. Similarly, the microstrip feeding arrangement for this Vivaldi array antenna is in
the same row, as seen in most of the antenna designs. Different prototypes of phase power
dividers and impedance transformation are presented in the literature [54–56]. These bulky
power dividers are mostly SMA connected, need multi-source signal feeding, and require
power cables to connect to another system or antenna, which may lead to a loss in a system.
Designing in-phase and out-of-phase power dividers within a limited area on a single
substrate layer for direct feeding to antenna array system is still a challenging task. In
addition, designing the antenna array system with the desired antenna combination and
managing the feeding arrangement concerning the number of antennas in an array is still a
tough problem.
To overcome the limited arrangement of the antennas in an array, remove the constraint
of adding feeding sources in a single row of Vivaldi antenna arrays, and eliminate the
beam-splitting effect, and significantly enhance the radiation directivity of the antenna
arrays, this paper presents a feeding network of a T-junction power splitter topology with
the frequency-independent phase shifter added on one of the two arms. Each arm is then
linked to three distinct sub-branches as a feeding source for the three Vivaldi antennas
array. These three antennas at each arm are arranged compactly at multi-row locations
separated by a length of guided wavelength so that the directivity is significantly increased,
making the set up far more sensible for object detection within ultra-wideband region of
(2.5–6.8 GHz) and (7.5–9.5 GHz). The U-shaped feed structures are placed in the arms,
in between sub-branches, or the feeding source of the Vivaldi array, to compromise the
signal delay to each feeding line. The antenna has a maximum realized gain of up to
14.12 dBi, with a front-to-back ratio of 23.23 dB. The high gain and directivity assure that
the antenna can be used for microwave imaging of the hollow voids and objects buried
under the concrete brick structure, as well as for applications requiring broad bandwidth
communication. The paper is arranged as follows: Section 2 presents the design of the
proposed antenna and its feeding structure, Section 3 consists of the parametric study of
simulated and measurement results and discussion, Section 4 discusses the operation of
the antenna with experimental results. Finally, Section 5 is the conclusion of the work.

2. Antenna Design
2.1. Antenna Structure
The proposed geometry of the antenna is shown in Figure 1, where Figure 1a rep-
resents the top geometrical view of the substrate with the optimal dimension listed in
Table 1, while Figure 1b shows the feeding portion of the antenna, which consists of a
T-junction topology with an independent phase shifter at one of the arms. The antenna
was designed on a Taconic substrate (εr = 4.3, tan δ = 0.0035). The size of the antenna is
167.48 mm × 158.25 mm × 0.6 mm. The antenna comprises a microstrip feeding network
on the top side, and on the bottom side is a ground plane with an exponential tapering
radiating patch. The 50-ohm quarter-wave microstrip line has a width “Fw”, which is
followed by a T-junction base with a width twice that of the microstrip line where the
junction base yields a symmetric T-junction power branch. The feed line length is set at
λg/4, where λg is the guided wavelength of the center of the UWB frequency band. One of
the power branches is routed through the frequency-independent phase shifter. This phase
Sensors 2021, 21, 6091 4 of 19

junction base with a width twice that of the microstrip line where the junction base yields
Sensors 2021, 21, 6091 4 of 17
a symmetric T-junction power branch. The feed line length is set at λg/4, where λg is the
guided wavelength of the center of the UWB frequency band. One of the power branches
is routed through the frequency-independent phase shifter. This phase shifter changes the
armed phase such
shifter changes that each
the armed power
phase suchbranch
that each haspower180+−20 -degree
branch change
has 180+ −20in-degree
a phase shift
change
which aids shift
in a phase in thewhich
generation
aids inofthe
a directive
generation stableof aradiation
directive pattern. Each arm
stable radiation is branched
pattern. Each
with
arm is three additional
branched withmicrostrip feed slots
three additional to the Vivaldi
microstrip antenna
feed slots to thearray, with
Vivaldi a widtharray,
antenna half
the
withbreadth
a widthofhalf
the the
arm. Each feed
breadth slot
of the arm.is separated
Each feedfrom itsseparated
slot is constituent slotits
from byconstituent
a distance
slotλby
of a distance
g whereas its of
lengthλg whereas
varies by g ⁄2. The
its λlength varies basebyofλ the
g /2.Vivaldi
The baseslotofantenna
the Vivaldiarrayslotis
antenna array
determined byisthe
determined
length and byspacing
the length andfeeding
of the spacing microstrip
of the feeding slotmicrostrip
line. As shownslot line.
in
As shown
Figure in Figure
1a, the tapered1a, thewhich
slot, tapered slot,exponential
is four which is four exponential
curves curves
E1, E2, E3, and E1,
E4, E2, E3, and
is specified
E4,terms
in is specified in terms
of the values of of
thethe values ofas
parameter the parameter
listed in Table as1,listed
given inby:
Table 1, given by:
λg 
1 5λg ln( ) λg
E1 : x = 1( 5λg+ Vw (exp (y ln w 2V)))(0 (1)
2V
w ≤ y ≤ Lv ),
E1 : x = 2 ( 2 + Vw (exp (y Lv )))(0 ≤ y ≤ Lv ) , (1)
2 2 Lv
ln( λg ) 
1 5λg 2V λg
E2 : x 1= 5λ ( − Vw (exp (yln 2Vww )))(0 ≤ y ≤ (Lv − λg )),
(2)
2 g2 Lv −λg
E2 : x = ( − Vw (exp (y ))) 0 ≤ y ≤ (Lv − λg ) , (2)
2 2 Lv − λg
λg
1 3λg
ln(λg  )
2Vw
E3 : 1x =3λg( − Vw (exp (y ln 2Vw )))(λg ≤ y ≤ (Lv − λg )),
(3)
2 2 Lv −2∗λg
E3 : x = ( − Vw (exp (y ))) λg ≤ y ≤ (Lv − λg ) , (3)
2 2 Lv − 2∗λg
ln( λg )
1 λg λ
2Vg
E4 : x1 =λg( − Vw (exp (y 2Vww )))(2λg ≤ y ≤ (Lv − λg )),

ln (4)
2 2 (Lv −3∗λg
E3 : x = ( − Vw (exp (y ))) 3λg ≤ y ≤ (Lv − λg ) , (4)
2 2 Lv − 3∗λg

x
y
z

(a) (b)
Figure 1. Structure
Figure 1. Structure of
of antenna:
antenna: (a)
(a) top
top view
view of
of the
the antenna
antenna presenting
presenting feed,
feed, substrate,
substrate, and
and ground
ground layer
layer structure;
structure; (b)
(b)
enlargement of the feeding top view section of an antenna from Figure
enlargement of the feeding top view section of an antenna from Figure 1a.1a.

Table 1. Parametric dimensions of the proposed antenna structures.

Parameter mm Parameter mm
Fw 1.2 Sw 0.3
λg 21.1 Vw 0.3
Rs 0.125 × Ws Sl λg/2
Sr λg/8 S5 2 × Fw
Fr λg/8 L5 10.25
Vr λg/8 Ls 147.7
Vfr λg/8 Ws 158.25
Lv 4 × λg a+b λg/4
 of the proposed antenna. Both the arms are separated by a length twice the feed width,
giving the common mode and differential mode impedance of 46.70 and 92.45 Ohms, re-
spectively.
As shown in Figure 3a, the simulating S-parameter of the power divider is below 10
Sensors 2021, 21, 6091 dB, satisfying the operating frequency range of 2.5–10 GHz for the proposed antenna. The 5 of 17
simulated insertion loss ranges from 2.1 dB to 5.7 dB. The power divider provides equal
power divisions with a phase difference of 180 ± 20 degrees between the two output ports
at segments 4 and 5 as observed in Figure 3b. Figure 3c shows the sum of the amplitude
of 2.2.
the Design of Feeding
two segments of Structure
the power divider. The physical length of the arm of segment 4
differs because of fringing
The schematic of theatproposed
the end of the circular
T-junction stubdivider
power and around the slot
topology feedline
by a(segment
frequency-
2).independent
This new effective
phase length
shifter ofonthe
onearm andarms
of the power loss at in
is shown microstrip slot representing
Figure 2a,b, line transitionthe
fabricated
results antenna
in a change prototype.
in phase The junction
and magnitude provides
difference thetwo
at the slots with
ends ofathe
signal uniform
segment. The in
magnitude
change andtilt
in beam phase
angle with characteristic
at the E-plane ofslot
theline impedance
proposed of 50inOhms.
antenna The phase
the entire shifter
operating
makes ais180+
frequency −20-degree
below phase
nine degrees shift
with on the microstrip
a directive beam such line because
that a smallofchange
the microstrip
in phase to
slot-line transition. As illustrated in Figure 2a, E-field lines at the output
and magnitude of two feeding ports in this proposed antenna design is insignificant and of segments 4 and
5 propagate
does not hamper in the
the opposite
directivity direction, which isAintegrated
of the antenna. considerable withchange
the leftinand right
phase orarm of the
magni-
proposed
tude antenna.
difference Both
may lead tothe
beamarms areand
split separated by ainlength
a decrease twice
gain and the feed width, giving
directivity.
the common mode and differential mode impedance of 46.70 and 92.45 Ohms, respectively.

3
3
5

5
1
1

4
4

2
2

Sensors 2021, 21, 6091 6 of 19

(a)

Top Layer Bottom Layer

(b)
Figure
Figure2.2.Proposed
Proposedantenna
antennawith
withpower
powerdivider:
divider:(a)
(a)power
powerdivider
dividerwith
withE-field
E-fieldlines
linesdistribution
distributionatat
each
each segment 1, 2, 3, 4, 5 of a microstrip line; (b) top and bottom layer of the fabricatedantenna.
segment 1, 2, 3, 4, 5 of a microstrip line; (b) top and bottom layer of the fabricated antenna.

As shown in Figure 3a, the simulating S-parameter of the power divider is below
10 dB, satisfying the operating frequency range of 2.5–10 GHz for the proposed antenna.
0
The simulated insertion loss ranges
180 from 2.1 dB to 5.7 dB. The power divider provides equal
-5 power divisions with a phase difference of 180 ± 20 degrees between the two output ports
-10
at segments 4 and 5 as observed in 90
Figure 3b. Figure 3c shows the sum of the amplitude of
the two segments of the power divider. The physical length of the arm of segment 4 differs
S-parameter (dB)

Phase (deg)

-15
because of fringing at the end of the circular stub and around the slot line (segment 2). This
0
-20 new effective length of the arm and power loss at microstrip slot line transition results in a
-25
change in phase and magnitude difference at the two ends of the S41 segment. The change in
S51
-90
[S41]–[S51]
S11 -Reflection coefficient (power divider)
-30 S51 -Insertion loss (power divider)
S41 -Insertion loss (power divider)
-180
-35
2 3 4 5 6 7 8 9 10 2 3 4 5 6 7 8 9 10

Frequency (GHz) Frequency (GHz)

Sensors 2021, 21, 6091 6 of 17

Top Layer Bottom Layer

beam tilt angle at the E-plane of the proposed (b)antenna in the entire operating frequency is
below nine degrees with a directive beam such that a small change in phase and magnitude
Figure 2. Proposed antenna with power divider: (a) power divider with E-field lines distribution at
of two feeding ports in this proposed antenna design is insignificant and does not hamper
each segment 1, 2, 3, 4, 5 of a microstrip line; (b) top and bottom layer of the fabricated antenna.
the directivity of the antenna. A considerable change in phase or magnitude difference
may lead to beam split and a decrease in gain and directivity.

0
180
-5

-10 90
S-parameter (dB)

Phase (deg)
-15

0
-20

S41
-25 S51
-90
[S41]–[S51]
S11 -Reflection coefficient (power divider)
-30 S51 -Insertion loss (power divider)
S41 -Insertion loss (power divider)
-180
-35
2 3 4 5 6 7 8 9 10 2 3 4 5 6 7 8 9 10

Frequency (GHz) Frequency (GHz)

(a) (b)

1.0

0.8

0.6
Magnitude

0.4

0.2
[S41]+[S51]

0.0
2 3 4 5 6 7 8 9 10

Frequency (GHz)

(c)
Figure3.3.S-parameter
Figure S-parameterwith
withphase
phase and
and magnitude
magnitude of the
of the power
power divider
divider withwith phase-shifter
phase-shifter at of
at one onetheofarms:
the arms: (a) output
(a) output port
port return
return loss;
loss; (b) (b) output
output port phase,
port phase, and (c)and (c) magnitude.
magnitude.

3. Results and Discussion

The commercially available high-frequency structure simulator (HFSS) software is
used for the optimization and simulation of the proposed antenna. The simulation and
measurement results are shown in Figure 4. Figure 4a shows impedance bandwidth of the
proposed antenna below 10 dB from (2.5–6.8 GHz) and (7.5–9.5 GHz). The realized gain is
shown in Figure 4b which is below 14.12 dB for the entire bandwidth with a close agreement
between the measured and simulated results. The simulated and measured results vary
slightly except at 4, 8, and 8.5 GHz, and can be endorsed by the losses taking place in
the connector due to dimension imperfection and parasitic effect, imperfect soldering of
the feed line with the connector, and fabrication errors during the etching process and
characterization of the parameters of the substrate.
 ment between the measured and simulated results. The simulated and measured results
vary slightly except at 4, 8, and 8.5 GHz, and can be endorsed by the losses taking place
in the connector due to dimension imperfection and parasitic effect, imperfect soldering
of the feed line with the connector, and fabrication errors during the etching process and
Sensors 2021, 21, 6091 characterization of the parameters of the substrate. 7 of 17

–5 15

14
`
–10 13

12

–15 11

Realized gain (dBi)

S- parameter (dB)

10

–20 9

8
–25 7

6
–30 Measured
Measured 5
Simulated
Simulated
4
–35 3
2 3 4 5 6 7 8 9 10 2 3 4 5 6 7 8 9 10
Frequency(GHz) Frequency (GHz)

(a) (b)

Figure4.4.Simulated
Figure Simulatedand
andmeasured
measuredresults
resultsofofthe
theantenna:
antenna:(a)
(a)return
returnloss;
loss;(b)
(b)realized
realizedgain.
gain.

The
Themeasured
measured2D 2Dradiation
radiationpatterns
patternsatat3,3,4,4,5.5,
5.5,7,7,and
and8.58.5GHz
GHzfrequencies
frequenciesofofthethe
fabricated
fabricated antenna are shown in Figure 5. The radiation patterns of theantenna
antenna are shown in Figure 5. The radiation patterns of the antennawithwitha a
frequency-independent phase shifter at one of the arms are almost directive in both the
frequency-independent phase shifter at one of the arms are almost directive in both the E-
E-plane (x–y plane) and H-plane (z–y plane) which is one of the required characteristics for
plane (x–y plane) and H-plane (z–y plane) which is one of the required characteristics for
a Vivaldi antenna array.
a Vivaldi antenna array.
Figure 6a shows the simulated electric field distribution at 4.5 GHz frequency; it can
Figure 6a shows the simulated electric field distribution at 4.5 GHz frequency; it can
be seen that the electric field radiated due to a change in surface current at each tapered slot
be seen that the electric field radiated due to a change in surface current at each tapered
of the Vivaldi array. These fields are superimposed to form the plane-like wave transmitted
slot of the Vivaldi array. These fields are superimposed to form the plane-like wave trans-
in the direction of wave propagation whose directionality is further enhanced by a semi-
mitted in the direction of wave propagation whose directionality is further enhanced by
elliptical substrate of Length Rs. It can be seen from Figure 6b that the gain of the antenna
a semi-elliptical substrate of Length Rs. It can be seen from Figure 6b that the gain of the
rises with an increase in the number of antenna array segments achieving a maximum gain
antenna rises with an increase in the number of antenna array segments achieving a max-
Sensors 2021, 21, 6091
of up to 15.3 dBi. Figure 6c represents the simulated radiation pattern of the antenna when
8 of 19 of the
imum gain of up to 15.3 dBi. Figure 6c represents the simulated radiation
the frequency-independent phase shifter is replaced by a simple feed line. As shown in the
pattern
antenna when
radiation thebeam
plot, the frequency-independent phaselosing
is split into two halves, shiftertheis tendency
replaced by to abesimple feedand
directive, line.
As shown in the
has a reduced gain. radiation plot, the beam is split into two halves, losing the tendency to
be directive, and has a reduced gain.
90 90

120 60 120 60
10 10

0 0

150 30 150 30
-10 -10

-20
-20

180 -30 0
180 -30 0

210 330
210 330

240 300
240 300
270
270

(a) (b)

Figure 5. Cont.
90 90

120 60 120 10 60
10

0
0
-10
150 30 150 30
-10 -20

-30
-20
-40

180 -30 0 180 -50 0

 210 330
210 330

240 300
240 300
270
270
Sensors 2021, 21, 6091 8 of 17
(a) (b)

90 90

120 60 120 10 60
10

0
0
-10
150 30 150 30
-10 -20

-30
-20
-40

180 -30 0 180 -50 0

210 330 210 330

Sensors 2021, 21, 6091 9 of 19

240 300 240 300

270 270

(c) (d)
90 90

120 60 120 60
10 10

0 0

150 30 150 30
-10 -10

-20 -20

180 -30 0 180 -30 0

210 330 210 330

240 300 240 300

270 270

(e) (f)

90
90
120 60
120 60 10
10

0
0

150 30 150 30
-10 -10

-20 -20

180 -30 0 180 -30 0

210 330
210 330

240 300
240 300
270
270

(g) (h)

Figure 5. Cont.
 90 90

120 60 120 60
10 10

Sensors 2021, 21, 6091 0 0 10 of 19

Sensors 2021, 21, 6091 150 30 150 30 9 of 17
-10 -10

-20 -20

180 90 -30 0 180 90 -30 0

120 60 120 60
10 10

0 0
210 330 210 330
150 30 150 30
-10 -10

-20 -20
240 300 240 300

180 270-30 0 180 270-30 0

(i) (j)
― ∙ ― Measured Cross-Pol.
210 ――― 330Measured Co-Pol.
210 330

Figure 5. Measured far-field radiation pattern at E-plane (a,c,e,g,i) and H-plane (b,d,f,h,j) at frequency 3, 4, 5.5, 7, 8.5 GHz,
respectively.
240 300 240 300

Figure 7a shows the front-to-back ratio and 3 dB 270

270 beamwidth, while Figure 7b repre-
sents
(i) the beam tilt angle of the proposed antenna in (j) the E-plane. The measured result of
high front-to-back ― ∙ratio at the operating
― Measured bandwidth and minimum tilt angle of E-plane
Cross-Pol.
beam ensures that ―――the antenna
Measured radiates maximum energy at the desired direction to enable
Co-Pol.
penetration through the substrate, which is n for handling GPR and microwave imaging
Figure Figure 5. Measured
5. Measured far-field
far-field radiation
radiation pattern
pattern
application. at at E-plane(a,c,e,g,i)
E-plane (a,c,e,g,i) and
and H-plane
H-plane(b,d,f,h,j)
(b,d,f,h,j)at frequency 3, 4,3,
at frequency 5.5,
4, 7,5.5,
8.57,
GHz,
8.5 GHz,
respectively.
respectively.
Figure 7a shows the front-to-back ratio and 3 dB beamwidth, while Figure 7b repre-
sents the beam tilt angle of the proposed antenna in the E-plane. The measured result of
16
high front-to-back ratio at the operating bandwidth and minimum tilt angle of E-plane
15
beam ensures that the antenna radiates
14
maximum energy at the desired direction to enable
penetration through the substrate,
13 which is n for handling GPR and microwave imaging
application. 12
Realized gain (dBi)

11
10
9
8
16
7
15
6
14 Simulated six vivaldi array
5 Simulated eight vivaldi array
13
Sensors 2021, 21, 6091 4 11 of 19
12
Realized gain (dBi)

3
11 2 3 4 5 6 7 8 9 10
10
Frequency (GHz)
9

(a) 8 (b)
7
90
6
120 60 Simulated six vivaldi array
105
Simulated eight vivaldi array
4
0
3
150 2 3 4 30 5 6 7 8 9 10
-10
Frequency (GHz)
-20

(a) (b)
180 -30 0

210 330

H- plane
E- plane

240 300

270

(c)

FigureFigure 6. Simulated
6. Simulated electricfield,
electric field,gain,
gain, and
and radiation
radiationpattern of the
pattern antenna:
of the (a) electric
antenna: field distribution
(a) electric at 4.5 GHz;
field distribution at (b)
4.5 GHz;
simulated realized gain of six and eight Vivaldi antennas array; (c) simulated radiation pattern at 4.5 GHz without phase
(b) simulated realized gain of six and eight Vivaldi antennas array; (c) simulated radiation pattern at 4.5 GHz without phase
shifter.
shifter.

60 24

50
22
 180 -30 0

210 330

Sensors 2021, 21, 6091 H- plane

10 of 17
E- plane

240 300

270

Figure 7a shows the front-to-back ratio and 3 dB beamwidth, while Figure 7b repre-
(c)
sents the beam tilt angle of the proposed antenna in the E-plane. The measured result of
Figure 6. Simulated electrichigh
field,front-to-back
gain, and radiation
ratio pattern of the antenna:
at the operating (a) electric
bandwidth andfield distribution
minimum tiltatangle
4.5 GHz; (b)
of E-plane
simulated realized gain of six andensures
beam eight Vivaldi antennas
that the array;
antenna (c) simulated
radiates maximumradiation pattern
energy at 4.5
at the GHz without
desired phase
direction to en-
shifter. able penetration through the substrate, which is n for handling GPR and microwave
imaging application.

60 24

50
22

Front to back ratio (dB)

40
Beam width (dB)

20

30

18
20

16
10
Beam width
Front to back ratio
0 14
2 3 4 5 6 7 8 9 10

Frequency (GHz)

(a) (b)

Figure7.7.The
Figure Themeasured
measuredvariation
variationof
ofbeam
beamcomponents
componentswith
withfrequency:
frequency:(a)
(a)front-to-back
front-to-backratio
ratioand
andbeam
beamwidth;
width;(b)
(b)E-plane
E-plane
beamtilts.
beam tilts.

The
Theradiation
radiation performance and feed
performance and feed system
systemofofthe
thesuggested
suggestedantenna
antennaisiscompared
compared to
to that of previous known Vivaldi antennas array in Table 2. As indicated in
that of previous known Vivaldi antennas array in Table 2. As indicated in the table,the table,
most
most antenna
antenna systems
systems employ
employ a T-junction
a T-junction powerpower divider.
divider. In comparison
In comparison to thetoexisting
the existing
ultra-
ultra-wideband Vivaldi antennas array, the suggested antenna offers enhanced antenna
wideband Vivaldi antennas array, the suggested antenna offers enhanced antenna array
array flexibility, feed management, and gain.
flexibility, feed management, and gain.
Table 2. Comparison table.

Number of Vivaldi Frequency Range

Ref. Feed System Gain (dBi)
Antenna Elements (GHz)
T-junction power divider and
[45] 8 9–11 12
GCPW transition
[47] 8 24.75–28.5 T-junction power divider 6.96–11.32
[51] 4 1–6 T-junction power divider 5.7–11.3
V-shaped power divider
[53] 4 2–11 3.75–11
with T-junction
[57] 4 25–31 4 × 4 butler matrix 10.2
T-junction power divider with
Proposed antenna 6 2.5–6.8 and 7.5–9.5 3.2–14.12
independent phase shifter

4. Experimental Study and Results

After validating the proposed antenna design, the experiment was performed in a
room, in a controlled configuration measuring setup for microwave imaging. The objective
of the test is to analyze and evaluate the change in the GPR images regarding the hidden
targets placed within these hollow structures. These experiments are designed to enhance
GPR survey techniques for quantifying the depth and width of hollow surfaces within a
concrete block and locating running cables, pipes, or any metallic surface within it.
Sensors 2021, 21, 6091 11 of 17

4.1. Specimens
The setup consists of a UWB radar module (NVA-R661 of Xethru Co., Oslo, Norway),
RF cables, supporting frame, and connectors connecting to PC and antenna modules
as shown in Figure 8a. The specimens represent an aerated concrete brick and hidden
substrate plates. The concrete block is 38.5 cm × 18.5 cm × 10 cm in size, with three
hollow-spaced structures measuring 9 cm × 5 cm each 2.5 cm apart are shown in Figure 8b.
A substrate plate with dimensions of 5.5 cm × 4.5 cm × 0.18 cm was taken as a target.
The X2 chip of the NVA-R661, as shown in the Figure 8c module, generates and transmits
UWB pulses of high-order Gaussian impulse signal with several GHz bandwidths of signal
duration in the order of nanoseconds [58]. The high-frequency signal was chosen based
on the maximum gain and return loss of the antenna, as well as the radar ability to have
high-resolution depth throughout the object. The impulse signal was set to around the
operating bandwidth of the antenna by selecting number 10 of the PGSelect command,
which has a center frequency (f c ) at 8.8 GHz and 3.1 GHz bandwidth with peak-to-peak
output amplitude of 0.54 volts. This satisfied the minimum separation of antenna distance,
d = 5 cm facing parallel to each other by the condition of d > λ c /4. The radar resolution
per frame depth was set to 0.20 cm on air, and when accounting for the concrete brick of
relative permittivity εr = 2.5, at 8.8 GHz, a resolution depth of 0.16 cm was considered.
Sensors 2021, 21, 6091 Matlab is used for signal analyzing and processing. Figure 9 shows the time and frequency
13 of 19
domain response of transmitted signal pulse in order of nanoseconds for the selected (f c )
at 8.8 GHz.

Figure
Figure Experimental
8. 8. measurement
Experimental measurementsetup: (a)(a)
setup: setup arrangement
setup with
arrangement antenna
with scanning
antenna thethe
scanning concrete beam
concrete withwith
beam the the
target
within; (b) proposed antenna with radar module support and concrete brick and; (c) UWB radar module.
target within; (b) proposed antenna with radar module support and concrete brick and; (c) UWB radar module.
gure 8. Experimental measurement
Sensors 2021, 21, 6091 setup: (a) setup arrangement with antenna scanning the concrete beam with the 12 of 17
rget within; (b) proposed antenna with radar module support and concrete brick and; (c) UWB radar module.

Sensors 2021, 21, 6091 (a) (b) 14 of 19

Figure 9. The transmitted

Figure pulse shapepulse
9. The transmitted and frequency
shape andspectrum
frequency ofspectrum
IR-UWB radar for PGSelect
of IR-UWB = 10;
radar for (a) transmitted
PGSelect = 10; signal in
the time domain and; (b) transmitted
(a) transmitted signal in thesignal pulse inand;
time domain the frequency domain.
(b) transmitted signal pulse in the frequency domain.
4.2. Signal Analyzing
4.2. Signal Analyzing
The reflected signal from the target consists of a clutter signal, target signal, and noise
is combinedly represented
The reflected signal from as: the target consists of a clutter signal, target signal, and noise
is combinedly represented as: ri = rc,i + rt,i + rn, (5)
ri = rc,i + rt,i + rn, (5)
where ri is the i-th received signal, consisting of clutter signal rc,i, the target signal rt,i, and
where ri is
the noise rn.the
Thei-th received
received rawsignal,
signal,consisting
as shown in of Figure
clutter10a,
signal rc,i, cross-correlated
is then the target signalwith
rt,i,
and the noise rn. The received raw signal, as shown in Figure 10a, is then cross-correlated
the template signal (Figure 9a). These template signals are the transmitted signal gener-
with the template signal (Figure 9a). These template signals are the transmitted signal
ated in the X2 chipset. This helps the signal to achieve the best enhancement in the signal-
generated in the X2 chipset. This helps the signal to achieve the best enhancement in the
to-noise ratio and correlated the pattern of the received signal concerning target position.
signal-to-noise ratio and correlated the pattern of the received signal concerning target
A correlated signal is shown in Figure 10b. The n-th correlated signals are taken and
position. A correlated signal is shown in Figure 10b. The n-th correlated signals are taken
grouped covering the whole specimen as a single bin and then the Fourier transform is
and grouped covering the whole specimen as a single bin and then the Fourier transform is
applied to those signals. The Welch’s power spectral density is estimated from the Fourier
applied to those signals. The Welch’s power spectral density is estimated from the Fourier
transform targeted signals and finally, holographical slices of 2-D space image are ob-
transform targeted signals and finally, holographical slices of 2-D space image are obtained
tained detecting the position of the hidden target.
detecting the position of the hidden target.

(a) (b)

Figure 10.
Figure 10. The received pulse
pulse shape
shape and
and correlated
correlated signals
signals from
from IR-UWB
IR-UWB radar
radar for
for PGSelect
PGSelect == 10; (a) received raw
raw signal
signal
strength and;
strength and; (b)
(b) correlated
correlated signal
signal pulse.
pulse.

4.3. Measurement Setup

The experiment is carried out with three hollow-shaped aerated concrete brick struc-
tures stacked on top of one another, with some metallic objects inserted within. The main
objective is to locate the concealed targets within these hollow structures. The proposed
antenna was placed 2.5 cm above the concrete structure’s surface, and scans were per-
formed manually with an interval of 0.25 cm per step. By summing the reflected echo
 Sensors 2021, 21, 6091 13 of 17

4.3. Measurement Setup

The experiment is carried out with three hollow-shaped aerated concrete brick struc-
tures stacked on top of one another, with some metallic objects inserted within. The main
objective is to locate the concealed targets within these hollow structures. The proposed an-
tenna was placed 2.5 cm above the concrete structure’s surface, and scans were performed
manually with an interval of 0.25 cm per step. By summing the reflected echo signals for
each antenna position, each scan progressively forms an image. The echo thus obtained by
each antenna position is considered as a pixel that is shifted to match or align with the next
scan pixel elements in the image map. A total of 176 scans were conducted to generate
a complete 2-D image of the scanned surface, as shown in Figure 11, where the figure
Sensors 2021, 21, 6091 represents the raw scanned data taken from the radar module before further applying 15 of 19
Sensors 2021, 21, 6091 15 of 19
signal processing. A substrate plate is placed as a target within the hollow-shaped concrete
brick at the depth of 7.5 cm from the surface. The experiment is made in three scenarios.

Figure 11. Scanned raw data from the radar module.

Figure
Figure11.
11.Scanned
Scannedraw
rawdata
datafrom
fromthe
theradar
radarmodule.
module.
First, the pair of antennas made the scans throughout the concrete brick without plac-
First,the
First, the pair
pair of antennas
of antennas mademade the scans
the scans throughout
throughout the concrete
the concrete brick without
brick without plac-
ing the target plates. Figure 12 shows the power spectral density of the three hollow 2-D
placing
ing the target
the target plates.plates.
FigureFigure 12 shows
12 shows the power
the power spectral
spectral densitydensity
of theof the three
three hollowhollow
2-D
cross surfaces
2-D surfaces
cross of theof
surfaces brick without
thewithout targetstargets
brick without where itwhere
can be seen that the signals are reflected
cross of the brick targets where it can be itseen
canthat
be seen that the
the signals aresignals are
reflected
strongly from
reflectedfrom the from
strongly air-brick surface. On
the surface.
air-brick moving
surface. further down
Onfurther
moving further through the section
down through of the
the of
section
strongly the air-brick On moving down through the section the
brick, the
of thethe three
brick, hollow
thehollow surfaces
three hollow can be easily
surfaces seen with significant penetration depth.
brick, three surfaces can be can be seen
easily easily seensignificant
with with significant penetration
penetration depth.depth.

Figure
Figure12. 2-Dcross-surface
12.2-D cross-surfacescanned
scannedimage
imagewithout
without the
the target.
target.
Figure 12. 2-D cross-surface scanned image without the target.

Second, all the three target plates are placed inside the hollow concrete brick, and
Second, all the three target plates are placed inside the hollow concrete brick, and
scans were made across the surface. As shown in Figure 13, signals are massively reflected
scans were made across the surface. As shown in Figure 13, signals are massively reflected
below the first layer of the air-surface interference indicating the presence of three hidden
below the first layer of the air-surface interference indicating the presence of three hidden
Sensors 2021, 21, 6091 16 of 19

Sensors 2021, 21, 6091 14 of 17

gain and narrow beam width over a wider frequency range, suitable for air-coupled GPR.
The limitation of the proposed antenna is that it cannot be utilized as a ground-coupled
Second,
antenna all the
because three
of the target plates
orientation are placed
of radiated inside
beam andtheantenna
hollow concrete brick, and
shape, making scans
it unable
were
to made
obtain acrossdata
clearer the and
surface. As shown
greater in inspection
depth of Figure 13, signals aresurface
for small massively reflectedsuch
anomalies below
as
the first layer of the air-surface interference indicating the presence of three hidden targets.
cracks.

Sensors 2021, 21, 6091 16 of 19

gain and narrow beam width over a wider frequency range, suitable for air-coupled GPR.
The limitation of the proposed antenna is that it cannot be utilized as a ground-coupled
antenna because of the orientation of radiated beam and antenna shape, making it unable
to obtain clearer data and greater depth of inspection for small surface anomalies such as
cracks.

Figure
Figure 13.
13. 2-D
2-D cross-surface
cross-surface scanned
scanned image
image with
with three
three targets.
targets.

After this, one middle target is removed from the hollow concrete brick, and scans
are again performed where the remaining two targets can be seen as shown in Figure 14.
From all those three cases, looking at the scanned image we can easily predict the hollow
surfaces and the hidden targets at the position around 7.5 cm from the antenna. The slight
discrepancy between the real and the predicted target position of the scanned image is
due to the variation in permittivity of hollow concrete brick. Poor reflectivity and strong
attenuation in the hollow interior of the brick hinder further tracking of in-depth microwave
imaging. The main advantage of this lightweight antenna is that it facilitates a higher gain
and narrow beam width over a wider frequency range, suitable for air-coupled GPR. The
limitation of the proposed antenna is that it cannot be utilized as a ground-coupled antenna
because of the orientation of radiated beam and antenna shape, making it unable to obtain
clearer13.data
Figure 2-D and greater depth
cross-surface ofimage
scanned inspection for small
with three surface anomalies such as cracks.
targets.

Figure 14. 2-D cross-surface scanned image with one target removed.

5. Conclusions
A novel feed system consisting of high gain six-slot Vivaldi antenna arrays on a single
substrate layer using a power splitter based on binary T-junction power splitter topology
and frequency-independent phase shifter has been presented. The antenna exhibits an
out-of-phase feeding mechanism in combination with proper Vivaldi antenna array lay-
out on separate rows causing high realized gain and a front-to-back ratio up to 14.12 dBi
and 23.23 dB, respectively, within ultra-wideband regions of (2.5–6.8 GHz) and (7.5–9.5
GHz). Broad bandwidth, high gain, and strong directivity are all benefits of the suggested
antenna, making it a viable option for applications that need broad bandwidth communi-
cation. The feeding system overcomes the limited arrangement of the antenna in an array
reported in the literature, removes the constraint of adding feeding sources of Vivaldi an-
Figure14.
Figure
tenna 2-Dcross-surface
14.2-D
arrays cross-surface
in scanned
a single scanned
row, imagewith
image
eliminateswith one
one
the targetremoved.
target removed.
beam-splitting effect, and significantly en-
hances
5. the radiation directivity of the antenna arrays. The fabricated antenna is deployed
Conclusions
5. Conclusions
for the detection of substrate plates as targets placed inside the concrete brick. To begin,
AAnovel
novelfeed
feedsystem
systemconsisting
consistingof
ofhigh
highgain
gainsix-slot
six-slotVivaldi
Vivaldiantenna
antennaarrays
arraysononaasingle
single
substrate layer using a power splitter based on binary T-junction power splitter topology
substrate layer using a power splitter based on binary T-junction power splitter topology
and frequency-independent phase shifter has been presented. The antenna exhibits an
out-of-phase feeding mechanism in combination with proper Vivaldi antenna array lay-
out on separate rows causing high realized gain and a front-to-back ratio up to 14.12 dBi
and 23.23 dB, respectively, within ultra-wideband regions of (2.5–6.8 GHz) and (7.5–9.5
Sensors 2021, 21, 6091 15 of 17

and frequency-independent phase shifter has been presented. The antenna exhibits an
out-of-phase feeding mechanism in combination with proper Vivaldi antenna array layout
on separate rows causing high realized gain and a front-to-back ratio up to 14.12 dBi and
23.23 dB, respectively, within ultra-wideband regions of (2.5–6.8 GHz) and (7.5–9.5 GHz).
Broad bandwidth, high gain, and strong directivity are all benefits of the suggested antenna,
making it a viable option for applications that need broad bandwidth communication. The
feeding system overcomes the limited arrangement of the antenna in an array reported in
the literature, removes the constraint of adding feeding sources of Vivaldi antenna arrays in
a single row, eliminates the beam-splitting effect, and significantly enhances the radiation
directivity of the antenna arrays. The fabricated antenna is deployed for the detection of
substrate plates as targets placed inside the concrete brick. To begin, scans were taken on
the concrete brick surface. The power spectral density of the scanned image reviled the
three hollow spaces within the brick. Second, all three target plates were placed into a
hollow concrete brick revealing the scanned image with three targets, and finally, one of the
targets was removed and scans were conducted, revealing the scanned image without one
of the targets. The 2-D imaging of the systems detecting the target location and a hollow
space inside the concrete brick confirms that the proposed antenna is suitable for use in
microwave imaging and GPR application.

Author Contributions: Conceptualization, J.G.; methodology, J.G.; software, J.G.; writing—original

draft preparation, J.G.; writing—review and editing, J.G.; supervision, D.-Y.C.; funding acquisition,
D.-Y.C. measurement, F.D.D.; resources/material assembly: J.-H.K. All authors have read and agreed
to the published version of the manuscript.
Funding: The BrainKorea21Four Program supported this research through the National Research
Foundation of Korea (NRF) funded by the Ministry of Education (4299990114316).
Institutional Review Board Statement: Not applicable.
Informed Consent Statement: Not applicable.
Data Availability Statement: Data sharing not applicable.
Conflicts of Interest: The authors declare no conflict of interest.

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