TI Designs

100-W, 0.1% Dimmable DC-DC LED Driver With Daylight

Harvesting and Wireless Connectivity

Description Features
This TI Design is a tested DC-DC LED driver • 100-W Synchronous Buck LED driver With Average
subsystem for high-power, high-efficiency dimmable of 97.3% Efficiency Over 100% to 50% Brightness
LED luminaires. The design is built on a wireless With Analog Dimming
system-on-chip (SoC) platform which can enable • 1:1000 Contrast Ratio With Analog Dimming, and
intensity adjustment through analog, PWM dimming, PWM Dimming (200 Hz to 5 kHz)
and control using any Bluetooth® low energy (BLE)
smart device. • Ambient Light Sensor OPT3001-Based Light
Measurement, Enabling Daylight Harvesting and
At the time of this writing, high-bay and low-bay LED Constant Lumen Implementations
lighting luminaires are replacing the fluorescent and • MCU PWM Used as 12-bit DAC for IADJ Setting in
HID lights because they cut energy consumption in Analog Dimming
half, and nearly eliminate maintenance costs. Daylight
harvesting using the dimming feature, combined with • Overcurrent and Overtemperature Protection for
an ambient light sensor, can provide up to an Driver and LED Module
additional 50% in energy savings, depending on the • CC2650 SimpleLink™ Multi-Standard 2.4-GHz
application. Ultra-Low-Power Wireless MCU Enables
The TIDA-01095 TI Design provides high-efficiency Connected Lighting With Bluetooth® Smart or
DC-DC conversion, allows dimming and daylight ZigBee®
harvesting, and enables wireless connected lighting Applications
control.
• Indoor LED Lighting (Industrial High-Bay, Low-Bay
Resources Lighting)
• Outdoor LED Lighting (Area Light, Street Light)
TIDA-01095 Design Folder
TPS92641 Product Folder • Distributed DC Lighting
CSD18537NQ5A Product Folder
CSD18563Q5A Product Folder
OPA376 Product Folder ASK Our E2E Experts
OPT3001 Product Folder
LMT84 Product Folder
LAUNCHXL-CC2650 Tools Folder

LAUNCHXL-CC2650 TIDA-01095
5V 36- to 50-V DC input

High side MOSFET

TPS79601
Ultra-Low-Noise, NexFET
N-Ch
RF LDO
MOSFET
RC Low Analog Dim
TPS92641 CSD18537NQ
pass OPA376 5A
Filter
Sync-Buck
PWM DIM Controller for NexFET
LED Driver N-Ch 33-V,100-W
MOSFET LED COB
CC2650 CSD18563Q5
A

Low-side MOSFET
Analog LMT84
Temperature Sensor for LED

I2C OPT3001
Ambient Light Sensor

Copyright © 2016, Texas Instruments Incorporated

SimpleLink, PowerPAD, NexFET, e-trim, LaunchPad, BoosterPack, Code Composer Studio are trademarks of Texas Instruments.
WEBENCH is a registered trademark of Texas Instruments.
Cortex is a trademark of ARM Holdings.
Bluetooth, Bluetooth are registered trademarks of Bluetooth SIG.
Windows is a registered trademark of Microsoft Corporation.
ZigBee is a registered trademark of Zigbee Alliance.
All other trademarks are the property of their respective owners.

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An IMPORTANT NOTICE at the end of this TI reference design addresses authorized use, intellectual property matters and other
important disclaimers and information.

1 System Overview

1.1 System Description

At the time of this writing, industrial indoor lighting is increasingly adopting light-emitting diodes (LEDs) for
high-bay and low-bay lighting. LED lighting offers several advantages, such as energy savings greater
than 50%, long lifetime, controllable light output, and extremely low maintenance. Integration of light
sensors with connected lighting enables daylight harvesting, which increases the energy efficiency.
Daylight harvesting is achieved by measuring the light conditions in the given work area, which could be
partially from the light source and partially from the daylight. Based on that data, the lights are
automatically dimmed when the daylight level increases.
Besides optical design and thermal management, LED luminaires must also ensure constant lumen output
from the luminaire over its lifetime (as the light output from the LED is expected to reduce over a few
years), and smooth and efficient dimming to enable daylight harvesting using light sensors.
Figure 1 shows an example of high-bay lighting in an industrial environment.

Figure 1. High-Bay Lighting in Industrial Environment

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1.2 Key System Specifications

Table 1. Key System Specifications

TEST CONDITIONS AND
PARAMETERS MIN TYP MAX UNITS
NOTES
INPUT CHARACTERISTICS
Input Voltage — 36 — 50 (1) V
Input UVLO setting — 30.25 31.9 33.55 V
OUTPUT CHARACTERISTICS
Output (LED) Current — 0 — 3 A
Inductor ripple current Peak-to-peak — — 500 mA
LED ripple current Peak-to-peak — — 350 mA
OVP Threshold — 53.7 57.5 61.3 V
SYSTEMS CHARACTERISTICS
Switching frequency — — 222 — kHz
Current sensing resistor — — 0.05 (2) — Ω
LOAD CHARACTERISTICS (3)
Forward voltage at 1900 mA, TC = 85°C — 38.5 — V
Forward voltage at 1900 mA, TC = 25°C — — 42 V
Forward current — — — 2800 mA
Reverse current — — — 0.1 mA
(1)
Maximum input voltage is limited by the MOSFET voltage rating. The board can work with 60-V MOSFET with up to 50-V input,
and it also can work with 100-V MOSFET with up to an 80-V input.
(2)
For deep dimming or dimming in the range of less than 10 mA, the current sense resistor value must be around 0.25 Ω.
(3)
LED load used is Cree CXA3070-0000-000N0HAB57F-ND.

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1.3 Block Diagram

LAUNCHXL-CC2650 TIDA-01095
5V 36- to 50-V DC input

High side MOSFET

TPS79601
Ultra-Low-Noise, NexFET
N-Ch
RF LDO
MOSFET
RC Low Analog Dim
TPS92641 CSD18537NQ
pass OPA376 5A
Filter
Sync-Buck
PWM DIM Controller for NexFET
LED Driver N-Ch 33-V,100-W
MOSFET LED COB
CC2650 CSD18563Q5
A

Low-side MOSFET
Analog LMT84
Temperature Sensor for LED

I2C OPT3001
Ambient Light Sensor

Copyright © 2016, Texas Instruments Incorporated

Figure 2. Block Diagram of Dimmable White LED Driver Augmented by

CC2650 LaunchPad and Sensors

1.4 Highlighted Products

1.4.1 TPS92641
The TPS92640 and TPS92641 devices are high-voltage, synchronous NFET controllers for buck-current
regulators (see Figure 3). Output current regulation is based on valley current-mode operation using a
controlled on-time architecture. This control method eases the design of loop compensation while
maintaining nearly constant switching frequency.
The TPS92640 and TPS92641 devices include a high-voltage start-up regulator that operates over a wide
input range of 7 V to 85 V. The PWM controller is designed for high-speed capability, including an
oscillator frequency range up to 1 MHz. The deadtime between the high-side and low-side gate driver is
optimized to provide very high efficiency over a wide input operating voltage and output power range.
The TPS92640 and TPS92641 devices accept both analog and PWM input signals, resulting in
exceptional dimming control range. Linear response characteristics between input command and LED
current is achieved with true zero LED current using low off-set error amplifier and proprietary PWM
dimming logic. Both devices also include precision reference capable of supplying current to a low-power
microcontroller. Protection features include cycle-by-cycle current protection, overvoltage protection, and
thermal shutdown. The TPS92641 device includes a shunt FET dimming input and MOSFET driver for
high-resolution PWM dimming.
Features:
• VIN range from 7 V to 85 V
• Wide dimming range
• 500:1 analog dimming
• 2500:1 standard PWM dimming
• 20000:1 shunt FET PWM dimming
• Adjustable LED current sense voltage
• 2-Ω, 1-A peak MOSFET gate drivers
• Shunt-dimming MOSFET gate driver (TPS92641)
• Programmable switching frequency
• Precision voltage reference 3 V ±2%
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• Input UVLO and output OVP

• Low-power shutdown mode and thermal shutdown
• MSOP-10 package with PowerPAD™

VIN 370mV
TPS92640, TPS92641 1.276V
VOLTAGE 2.54V 3.03V
VCC BIAS VCC REFERENCES VREF
VIN REGULATOR VDD

COMP THERMAL
VCC UVLO VCC
SHUTDOWN
IADJ VSW
PWM_DIM
2.54V 9R EA GATE DRIVE UVLO BOOT
+
R - SD

FSW
+
CS - tON DEAD TIME / H.S.
R Q Driver HG
370mV -
13ms FILTER
Shutdown tOFF LEVEL SHIFT
+ S Q
LGATE Enable VSW SW
21µA
UDIM + PWM_DIM / UVLO VCC
1.276V -

LOGIC DEAD L.S.

RON Driver LG
TIME
tON_ Reset
+ End tON
-
tON_Reset GND

LEB TIMER
VOUT
+ OVP
3.05V -

TPS92641 ONLY
VDD
VCC

SDIM + PWM
SDRV
1.276V - LOGIC

Copyright © 2016, Texas Instruments Incorporated

Figure 3. TPS92640, TPS92641 Block Diagram

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1.4.2 TPS92640
The TPS92640 device is a high-voltage, synchronous NFET controller for buck-current regulators from the
TPS9264x device family, and can be used instead of the TPS92641 in the same design. The TPS92641
device includes a shunt FET dimming input and MOSFET driver for high-frequency and high-resolution
PWM dimming, in addition to all the features provided by the TPS92640. The TPS92640 can fit in end
applications which do not exploit the shunt dimming feature of the TPS92641.

1.4.3 CSD18537NQ5A
This 10-mΩ, 60-V, SON 5×6-mm NexFET™ power MOSFET from TI is designed to minimize losses in
power conversion applications. This MOSFET is recommended to be used as the high-side MOSFET in
synchronous buck converter applications.
Features:
• Ultra-low Qg and Qgd
• Low thermal resistance
• Avalanche rated
• Pb-free terminal plating
• RoHS compliant

1.4.4 CSD18563Q5A
This 5.7-mΩ, 60-V SON 5×6-mm NexFET power MOSFET is designed to pair with the CSD18537NQ5A
control FET, and act as the sync FET for a complete industrial buck-converter chipset solution.
Features:
• Ultra-low Qg and Qgd
• Soft body diode for reduced ringing
• Low thermal resistance
• Avalanche rated
• Logic level
• Pb-free terminal plating
• RoHS compliant
• Halogen free
• SON 5×6-mm plastic package

1.4.5 CSD18563Q5A
This 100-V, 12.6-mΩ, SON 5x6-mm NexFET power MOSFET is designed to minimize losses in power
conversion applications.
Features:
• Ultra-low Qg and Qgd
• Low thermal resistance
• Avalanche rated
• Pb-free terminal plating
• RoHS compliant
• Halogen free
• SON 5×6-mm plastic package

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1.4.6 OPT3001
The OPT3001 is a sensor that measures the intensity of visible light. The spectral response of the sensor
tightly matches the photopic response of the human eye, and includes significant infrared rejection (see
Figure 4).
The OPT3001 is a single-chip LUX meter that measures the intensity of light as visible by the human eye.
The precision spectral response and strong IR rejection of the device enables the OPT3001 device to
accurately meter the intensity of light as seen by the human eye, regardless of the light source. The strong
infrared (IR) rejection also aids in maintaining high accuracy when industrial design calls for mounting the
sensor under dark glass for aesthetics. The OPT3001 is designed for systems that create light-based
experiences for humans, and an ideal preferred replacement for photodiodes, photoresistors, or other
ambient light sensors with less human eye matching and IR rejection.
Measurements can be made from 0.01 lux up to 83k lux without manually selecting full-scale ranges, by
using the built-in, full-scale setting feature. This capability allows light measurement over a 23-bit effective
dynamic range.
The digital operation is flexible for system integration. Measurements can be either continuous or single-
shot. The control and interrupt system features autonomous operation, allowing the processor to sleep
while the sensor searches for appropriate wake-up events to report through the interrupt pin. The digital
output is reported over an I2C- and SMBus-compatible, two-wire serial interface.
The low-power consumption and low-power-supply voltage capability of the OPT3001 enhances the
battery life of battery-powered systems.
Features:
• Precision optical filtering to match human eye
• Rejects > 99% (typ) of IR
• Automatic full-scale setting feature simplifies software and ensures proper configuration
• Measurements: 0.01 lux to 83k lux
• 23-bit effective dynamic range with automatic gain ranging
• 12 binary-weighted full-scale range settings
• < 0.2% (typ) matching between ranges
• Low operating current: 1.8 µA (typ)
• Operating temperature range: –40°C to +85°C
• Wide power-supply range: 1.6 V to 3.6 V
• 5.5-V tolerant I/O
• Flexible interrupt system
• Small-form factor: 2.0 mm × 2.0 mm × 0.65 mm
VDD

VDD
OPT3001 Digital Processor
SCL SCL
Ambient I 2C SDA SDA
Optical ADC Interface INT
Light INT or GPIO
Filter ADDR

GND

Copyright © 2016, Texas Instruments Incorporated

Figure 4. OPT3001 Block Diagram

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1.4.7 CC2650
The CC2650 device is a wireless MCU targeting Bluetooth® Smart, ZigBee®, 6LoWPAN, and ZigBee
RF4CE remote control applications (see Figure 5).
The device is a member of the CC26xx family of cost-effective, ultra-low-power, 2.4-GHz RF devices.
Very-low active RF and MCU current and low-power mode current consumption provide excellent battery
lifetime and allow for operation on small coin cell batteries and in energy-harvesting applications.
The CC2650 device contains a 32-bit ARM Cortex™-M3 processor that runs at 48 MHz as the main
processor, and a rich peripheral feature set that includes a unique ultra-low-power sensor controller. This
sensor controller is ideal for interfacing external sensors, and for collecting analog and digital data
autonomously while the rest of the system is in sleep mode. Thus, the CC2650 device is ideal for
applications within a whole range of products including industrial, consumer electronics, and medical.
The BLE controller and the IEEE 802.15.4 MAC are embedded into ROM, and are partly running on a
separate ARM Cortex-M0 processor. This architecture improves overall system performance and power
consumption, and frees up flash memory for the application.

Figure 5. CC2650 Block Diagram

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1.4.8 OPA376
The OPA376 family represents a new generation of low-noise operational amplifiers with e-trim™, offering
outstanding DC precision and AC performance. Rail-to-rail input and output, low offset (25 µV maximum),
low noise (7.5 nV/√Hz), quiescent current of 950 µA (maximum), and a 5.5-MHz bandwidth make this part
attractive for a variety of precision and portable applications. In addition, this device has a reasonably-
wide supply range with excellent power supply rejection ratio (PSRR), which makes it desirable for
applications that run directly from batteries without regulation.
The OPA376 (single version) is available in MicroSIZE SC70-5, SOT-23-5, and SOIC-8 packages. The
OPA2376 (dual) is offered in the DSBGA-8, VSSOP-8, and SOIC-8 packages. The OPA4376 (quad) is
offered in a TSSOP-14 package. All versions are specified for operation from –40°C to +125°C.
Features:
• Low noise: 7.5 nV/√Hz at 1 kHz
• 0.1 Hz to 10 Hz noise: 0.8 µVPP
• Quiescent current: 760 µA (typical)
• Low offset voltage: 5 µV (typ)
• Gain bandwidth product: 5.5 MHz
• Rail-to-rail input and output
• Single-supply operation
• Supply voltage: 2.2 V to 5.5 V
• Space-saving packages: SC70, SOT-23, DSBGA, VSSOP, TSSOP

1.4.9 LMT84
The LMT84 and LMT84-Q1 are precision CMOS integrated-circuit temperature sensors with an analog
output voltage that is linearly and inversely proportional to temperature. The sensor features make it
suitable for many general temperature-sensing applications. The LMT84 can operate down to a 1.5-V
supply with 5.4-µA power consumption, making it ideal for battery-powered devices.
Package options, including the through-hole TO-92 package, allows the LMT84 to be mounted onboard,
off-board, to a heat sink, or on multiple locations in the same application. A class-AB output structure
gives the LMT84/LMT84-Q1 strong output source and sink current capability that can directly drive up to
1.1-nF capacitive loads. The LMT84 is well suited to drive an ADC sample-and-hold input with its transient
load requirements. The device has accuracy specified in the operating range of −50°C to 150°C. The
accuracy, three-lead package options, and other features also make the LMT84/LMT84-Q1 an alternative
to thermistors.
Features:
• LMT84-Q1 is AEC-Q100 Grade 0 qualified and is manufactured on an automotive grade flow
• Low 1.5-V operation
• Very accurate: ±0.4°C typical
• Wide temperature range of –50°C to 150°C
• Low 5.4-µA quiescent current
• Average sensor gain of –5.5 mV/°C
• Output is short-circuit protected
• Push-pull output with ±50-µA drive capability
• Footprint compatible with the industry-standard LM20/19 and LM35 temperature sensors
• Cost-effective alternative to thermistors

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2 System Design Theory

LEDs require constant current drive, and, in most cases, the current must be adjustable to enable
dimming. Thus the user must have a regulated AC-DC power supply and adjustable current controller to
facilitate dimming. For DC lighting systems, low-voltage DC is directly available, and thus a current
controller with dimming capability is adequate. To enable dimming and control, use a wireless control or
wired control system. Wireless lighting controls are becoming more common because of their ease of use
and availability.
The TIDA-01095 platform uses TPS92641, a synchronous buck controller for a precision dimming LED
drive. The controller requires two external MOSFETs that must be sized based on the power
requirements. The TPS92641 is designed for high-speed capability, including an oscillator frequency
range of up to 1 MHz. The dead time between high-side and low-side gate drivers is optimized to provide
very high efficiency over a wide input operating voltage and output power range. The TPS92641 device
accepts both analog and pulse width modulation (PWM) input signals, resulting in exceptional dimming
control range. Linear response characteristics between the input command and LED current is achieved
with true zero LED current, using a low off-set error amplifier and proprietary PWM dimming logic. For
dimming control and wireless connectivity, the SimpleLink™ technology, multi-standard, 2.4-GHz ultra-
low-power wireless MCU CC2650 is used. Besides 2.4-Ghz RF connectivity, the built-in peripherals such
as an analog-to-digital converter (ADC) and PWMs are useful in this lighting application. The SimpleLink
CC2650 Wireless MCU LaunchPad™ kit generates one PWM for the PWM dimming and another PWM,
followed by a four-stage low-pass filter and low-offset operational amplifier (op amp) as a buffer-generated
variable voltage as IADJ, to enable analog dimming. Dimming through IADJ is more efficient and produces
less electromagnetic interference (EMI). However, at very low currents, there is slight variation in the color
temperature of the LEDs, which may not be desirable. The PWM dimming method avoids this issue and
allows higher resolution dimming. However, both dimming methods can be combined through software to
achieve both high efficiency and wider dimming resolution.
The OPT3001 Digital Ambient Light Sensor (ALS) with high-precision human eye response is interfaced
with the CC2650 MCU and thus features, such as constant lumen output and daylight energy harvesting
by automatic dimming of LEDs with the presence of sun light, can easily be implemented in the software.
The CC2650 SimpleLink multi-standard, 2.4-GHz ultra-low-power wireless MCU lets the user implement
any of the various radio frequency (RF) connectivity standards, such as Bluetooth Smart, ZigBee,
6LoWPAN, and ZigBee RF4CE for remote control applications. The LMT84 1.5 V-capable, 10-µA analog
output temperature sensor in the TO-92 package allows the user to measure the temperature of the LED
heatsink, which enables automatic foldback dimming in the case of overtemperature, and enables LED
string or LED COB protection.

2.1 Design Equations

2.1.1 Undervoltage Lockout (UVLO)

The UDIM pin of the TPS92641 is a dual-function input that features an accurate 1.276-V threshold with
programmable hysteresis. This pin functions as both the PWM dimming input of the LEDs and as an input
UVLO with built-in hysteresis. When the pin voltage rises and exceeds the 1.276-V threshold, 21 μA
(typical) of current is driven out of the UDIM pin into the resistor divider (RUDIM1, RUDIM2) providing
programmable hysteresis. The UVLO turnon threshold, VTURN_ON, is defined in Equation 1:
æ (R + RUDIM2 ) ö
VTURN _ ON = 1.276 ´ çç UDIM1 ÷÷
è RUDIM2 ø (1)
To set an undervoltage lockout threshold of 31.9 V (typical), chose RUDIM1 and RUDIM2 as RUDIM1 = 120 k and
RUDIM2 = 5 k. The minimum and maximum threshold values can be calculated based on the corresponding
values specified in the TPS92641 data sheet, as in Equation 2 and Equation 3.
æ (R + RUDIM2 ) ö
VTURN _ ON _ MIN = 1.21 ´ çç UDIM1 ÷÷ = 30.25 V
è RUDIM2 ø (2)
æ (R + RUDIM2 ) ö
VTURN _ ON _ MAX = 1.342 ´ çç UDIM1 ÷÷ = 33.55 V
è RUDIM2 ø (3)

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2.1.2 Overvoltage Protection (OVP)

The TPS92641 has programmable overvoltage protection by using the resistor divider at the VOUT pin.
The OVP limit, VOVP_ON, is defined as in Equation 4.
æ (R + RUDIM2 ) ö
VOVP _ ON = 3.05 ´ çç UDIM1 ÷÷
è RUDIM2 ø (4)
The values of RVOUT1 and RVOUT2 have been chosen to set the limit at 57.5, that is, RVOUT1 = 100 k and
RVOUT2 = 5.6 k. If the output voltage reaches VOVP_ON, the HG, LG, and SDRV pins are pulled low to prevent
damage to the LEDs or the rest of the circuit. The OVP circuit has a fixed hysteresis of 100 mV before the
driver attempts to switch again.

2.1.3 Switching Frequency

The switching frequency, fSW, can be calculated using the following equation from the TPS92641 data
sheet, as in Equation 5.
æ (R VOUT1 + R VOUT2 ) ö
fSW = ç ÷÷
ç (R
è VOUT2 ´ RON ´ CON ) ø (5)
For setting fSW = 220 kHz, the following values of the resistance and capacitance can be used: RON = 47k,
CON = 1.8 nF.
æ (100 + 5.6 ) ö
fSW = ç MHz = 222.89 kHz
ç (5.6 ´ 47 ´ 1.8 ) ÷÷
è ø (6)

2.1.4 Adjustable LED Current (IADJ)

The average LED current regulation is set using a sense resistor in series with the LEDs. The internal
error amplifier regulates the voltage across the sense resistor (VCS) to the IADJ voltage divided by 10. IADJ
can be set to any value up to 2.54 V, by connecting it to VREF through a resistor divider for static output
current settings. IADJ can also be used to change the regulation point, if connected to a controlled voltage
source or potentiometer, to provide analog dimming. IADJ can also be configured for thermal foldback
functions.
The set LED current depends on RCS and VCS, as shown in Equation 7 and Equation 8.
V
I LED = CS
RCS (7)
VIADJ
VCS =
10 (8)
This controllable analog voltage is generated by a buffered, four-pole, RC low-pass filter, which in turn
takes a variable PWM input from the CC2650 MCU. The maximum analog output voltage is 3.3 V (logic
high) at 100% duty cycle. To match this with 2.54 V, a resistor divider is placed following the output of the
buffer. The chosen values of the resistors are 174 Ω at the buffer side and 542 Ω at the ground side,
giving a full scale output voltage of ≈2.596 V.

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2.1.5 Fourth-Order Passive Low-Pass Filter

To utilize the analog dimming feature of the TPS92641, a variable analog voltage is required. This voltage
is achieved by low-pass filtering the PWM generated by the CC2650 device to obtain its average value.
The passive RC low-pass filter gives an analog voltage output with 12 bits of resolution, as in Equation 9.
V
Re solution = PWM
2n
where
• Resolution is the minimum incremental change in the analog output voltage with a change in PWM
duty cycle
• VPWM is the amplitude of the PWM signal
• n is the resolution in bits for the analog signal (12 in this case) (9)
Re solution V
Minimum Ripple = = PWMn
2 2´2 (10)
VPWM
Ripple =
10Order (11)
VPWM V
= PWMn
10Order 2´2 (12)
Order = (n + 1)l og (2 ) (13)
Therefore, the order = 13 × 0.3 = 3.9.
Round up the order to the next highest integer, if it is fractional, to achieve a higher performance than the
goal. Thus, the order = 4.
The equation that sets the cutoff frequency for a simple first-order, RC low-pass filter is given as
Equation 14.
1 1
R1 = = = 866 W
(
2p ´ f CUTOFF ´ C1 )
(2p ´ 391 Hz ´ 470 nF )
where
f PWM 3.91 kHz
f CUTOFF = = = 391 Hz
• 10 10
• R1 and C1 = first-stage low-pass RC filter
• C1 is selected arbitrarily as a standard value; choose near 1 uF as the capacitance in each subsequent
stage divided by 10 (14)
The RC filter loads the microcontroller. The load current is at a maximum when the PWM signal makes a
logic level transition (such as low-to-high or high-to-low). The transient current can be estimated as shown
in Equation 15.
V 3.3
ITRANSIENT = CC = = 3.8 mA
R1 866 (15)
The transient current is 3.8 mA, which is a reasonable load for the CC2650 device. To obtain a higher-
order filter, additional stages of the filter can be cascaded. However, ensure that subsequent stages do
not load the initial stage. A simple approach to prevent the loading is to increase the impedance of each
subsequent stage by a factor of ten, as shown in Table 2.

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Table 2. Impedance Stages

RESISTOR CAPACITOR
LOW-PASS FILTER STAGE RESISTOR VALUE CAPACITOR VALUE
DESIGNATOR DESIGNATOR
Stage 1 R15 866 Ω C14 0.47 µF
Stage 2 R16 8.66 KΩ C15 0.047 µF
Stage 3 R17 86.6 KΩ C16 0.0047 µF
Stage 4 R18 866 KΩ C17 470 pF

Figure 6. VPWM versus Time (1 of 2)

Figure 7. VPWM versus Time (2 of 2)

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2.1.6 Control Loop Compensation (COMP)

Compensating the TPS92641 is relatively simple for most applications. The only compensation required is
a compensation capacitor, CCOMP across the COMP pin, and a ground to place a low-frequency dominant
pole in the system. The pole must be placed low enough to ensure adequate phase margin at the
crossover frequency. For most applications, a CCOMP of 100 nF to 470 nF is adequate, and a 100-nF
capacitor has been selected for this application. Additionally, TI recommends a high-quality ceramic
capacitor with an X7R dielectric rated for 25 V.

2.1.7 Inductor Selection

Because this is a PWM-dimming application, too much output capacitance is not recommended for faster
current rise and fall times, so the inductor ripple current must be close to the 500-mA peak-to-peak. The
governing equation which relates the inductor value (L), inductor ripple current (IL), fSW, VIN, and VOUT is
shown in Equation 16.
æ (V - VOUT ) ´ D ö
L = ç IN ÷
ç DI L ´ fSW ÷
è ø
where
• In this application, VIN = 48 V
• fSW = 222 kHz
• VOUT = VLED + VCS = 38.5 + 0.15 = 38.65
VOUT 38.65
D= = = 0.81
• VIN 48 (16)

L=
(9.35 ´ 0.81) = 68.23 mH
Thus, solve for L: (0.5 ´ 0.222 )
Choose the standard inductor value 68 µH, which results in an ∆IL of 501 mA.

2.1.8 LED Ripple Current Selection

LED ripple current, ΔILED, in an LED driver is the equivalent of output voltage ripple, ΔVO, in a voltage
regulator. In general, the requirements for ΔILED are not as tight as the output voltage ripple. A ripple of a
few mV to 4% P-P of VO is typical for ΔVO, whereas ripple currents for LED drivers range from 10% to 40%
P-P of the average forward current. Allowing larger ripple current means lower inductance and
capacitance for the output filter, which in turn translates to smaller printed-circuit board (PCB) footprints
and lower bill of material (BOM) costs. For this reason, ΔILED can generally be made as large as the
application permits. This application is designed for an LED peak-to-peak ripple current equal to 1/8th or
12.5% of the maximum forward current (2800 mA = 2.8 A), ΔILED = 350 mA.

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2.1.9 Dynamic Resistance of LED

Load resistance is an important parameter in power supply design, particularly for the control loop. In LED
drivers, load resistance is used to select the output capacitance required to achieve the desired LED ripple
current. When the load is an LED or string of LEDs, however, the load resistance is replaced with the
dynamic resistance, rD, and the current sense resistor. Typical dynamic resistance at a specified forward
current is provided by some manufacturers, but in most cases it must be calculated using I-V curves.
The dynamic resistance calculation for the selected LED load is done as shown in Table 3, based upon
the IV measurements.

Table 3. Forward Current and Voltage

FORWARD CURRENT (A) FORWARD VOLTAGE (V)
2.748 41.80
2.462 40.77
2.169 39.75
1.872 38.74
1.578 37.74
1.284 36.72
0.991 35.68
0.700 34.58
0.409 33.29

A least square trend line can be fit in the above data to calculate the dynamic resistance. The equation of
the trend line is V = 3.5691I + 32.038. Therefore, the dynamic resistance, rD, comes out to 4.34 Ω.

2.1.10 Output Capacitor Selection

The LED manufacturers generally recommend values of current ripple, ΔILED, to achieve optimal optical
efficiency. The peak-to-peak current ripple values typically range from ±10% to ±40% of DC current, ILED. A
capacitor placed in parallel with the LED or array of LEDs can be used to reduce ΔILED while keeping the
same average current through both the inductor and the LED array. With this topology, the inductance can
be lowered, making the magnetics smaller and less expensive.
DI L
DI LED =
æ æ r öö
çç 1 + ç D ÷ ÷÷
è è ZCOUT ø ø (17)
1
ZCOUT =
( 2p ´ f SW ´ COUT ) (18)
Rearranging Equation 17 and Equation 18 shows the relation for the required value of COUT shown in
Equation 19.

COUT =
(DI L - DI LED ) =
(0.501 - 0.35 ) mF = 87 nF
(2p ´ f SW ´ rD ´ DI LED ) (2p ´ 0.222 ´ 3.5691 ´ 0.35 ) (19)
Therefore, choose COUT to be 0.1 uF. The actual value of ΔILED for 0.1 uF turns out to be 334 mA.
For low dimming currents of up to 100 uA, a higher output capacitance is required to reduce the output
voltage ripple, and thus the LED ripple current. For this purpose, a 1-uF capacitor is placed in parallel with
the calculated 0.1-uF capacitor. If low dimming currents are not required, then this component can be left
unpopulated.

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2.1.11 Minimum Input Capacitance

Input capacitance is necessary to provide instantaneous current to the discontinuous portions of the circuit
during the high side NFET on-time. The allowable input voltage ripple (ΔVIN-PP) is specified at
approximately 4 V peak-to-peak of VIN = 48 V. The minimum required capacitance (CIN_MIN) to achieve this
specification is as in Equation 20.

C IN _ MIN =
(I LED ´ D) =
(2.8 - 0.81) = 2.55 mF
(DVIN-PP ´ f SW ) (4 ´ 0.222 ) (20)
TI recommends that a higher capacitance be chosen than the value calculated above, especially for PWM
applications. Thus, two capacitors, one each of 1 uF and 2.2 uF, are placed in parallel to jointly make up
an equivalent capacitance of 3.2 uF.

2.1.12 MOSFET Selection

The TPS92640 and TPS92641 devices require two external NFETs for the switching regulator. The FETs
should have a voltage rating at least 20% higher than the maximum input voltage to ensure safe operation
during the ringing of the switch node. In practice, all switching converters have some ringing at the switch
node, due to the diode parasitic capacitance and the lead inductance. The NFETs should also have a
current rating at least 50% higher than the average transistor current. Once NFETs are chosen, the power
rating is verified by calculating the power loss.
The suggested minimum voltage rating, VT_MAX and current rating, IT_MAX are as in Equation 21 and
Equation 22.
V T _ MAX = 1.2 ´ VIN _ MAX = 1.2 ´ 50 = 60 V (21)
æ 42 ö
I T _ MAX = 1.5 ´ DMAX ´ I LED = 1.5 ´ ç ÷ ´ 2.8 = 3.675 A
è 48 ø (22)
The MOSFETs chosen in this application are CSD18537NQ5A (60-V, 50-A N-Channel NexFET™ Power
MOSFET) for the high side and CSD18563Q5A (60-V, 50-A N-Channel NexFET Power MOSFET, logic
level compatible) for the low side. These pair of MOSFETs are designed to minimize losses for power
conversion applications. Specifically, the CSD18563Q5A was designed to pair with the CSD18537NQ5A
control FET and act as the sync FET for a complete industrial buck converter chipset solution.

NOTE: The TIDA-01095 board is also tested with 100-V MOSFETs to enable DC-DC driver
operating voltage up to 80 V. Texas Instruments CSD19534Q5A 100 V N-Channel NexFET
Power MOSFETs are used for both high-side and low-side switch.

2.1.13 High-Side Gate Resistor

As the performance of power devices has improved, the control FET has the ability to switch voltages at
rates greater than 10 kV/µs. However, the fast switching faces a common challenge of dealing with
switching noise. In particular, when the Control FET turns on and the Sync FET is off, the loop inductor,
the loop resistor, and the output capacitor of the sync FET form a series RLC loop and resonate at a
resonant frequency. This resonance results in voltage overshoot and ringing at the switch node. Using a
resistor in series with the gate of the high-side FET is an effective way to reduce ringing. Similar to the
boot-resistor method, this resistor slows down the turnon of the high-side FET. However, because this
resistor is in series with the gate, it is also in the discharge path, so it slows down the turnoff as well. To
reduce the ringing for this design, a 24-Ω gate resistor is used.
The following waveforms in Figure 8, Figure 9, Figure 10, and Figure 11 show the effect of the gate
resistor on the switch node ringing.

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60-V MOSFETs:

Figure 8. Voltage Waveform Across Drain Source of Low-Side MOSFET Without Any Gate Resistor,
Peaking Around 68 V

Figure 9. Voltage Waveform Across Drain Source of Low-Side MOSFET With 25-Ω Gate Resistor,
Peaking at 57.6 V

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100-V MOSFETs:

Figure 10. Voltage Waveform Across Drain Source of Low-Side MOSFET Without Any Gate Resistor,
Peaking Around 124 V

Figure 11. Voltage Waveform Across Drain Source of Low-Side MOSFET Without 25-Ω Gate Resistor,
Peaking at 82.4 V

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2.2 Dimming Techniques

2.2.1 Analog Dimming Using IADJ

The LED current can be set and controlled dynamically by using the IADJ pin of the TPS92641 device. In
this application, the VIADJ voltage is obtained from a fourth-order passive LPF followed by an OPA376
operational amplifier used as a voltage follower. This low-pass filter converts the digital PWM waveform
(logic high – 3.3 V and logic low – 0 V) from the CC2650 to an analog voltage equal to the average value
of the PWM waveform. The output of the op-amp feeds a resistor divider to scale the maximum possible
input voltage from 3.3 V to 1.57 V as desired, and to not let the LED current exceed 3 A, the set maximum
value. Refer to Section 4 for the set of measurements taken using this feature.
The fourth-order low pass filter is designed for PWM frequencies of 3.91 kHz and above. The operating
frequency of the CC2650 MCU is fMCU = 48 MHz. This corresponds to the cycle time of:
TMCU = 125 / 6 = 20.83 ns. The desired PWM frequency is generated by counting these cycles.
To generate a PWM frequency of fPWM, the MCU cycles must be counted until TPWM / TMCU, where
TPWM = 1 / fPWM.
The following list provides more details:
• For the counts: NPWM = TPWM / TMCU = fMCU / fPWM = 48,000,000 / fPWM.
• For any chosen fPWM, the minimum possible duty cycle (resolution of the PWM) can be obtained by
keeping the corresponding I/O pin high for the duration of only 1 MCU cycle, that is, 1 count.
• The PWM resolution = 1 / NPWM × 100%.
• For fPWM = 4 kHz, the PWM resolution is 0.0083%. Thus, one count results in an incremental output
voltage of 0.0083% of 3.3 V = 0.275 mV at the output of the buffer.

2.2.2 Digital PWM Dimming Using UDIM

The UDIM pin can be driven with a PWM signal, which controls the synchronous NFET operation. The
brightness of the LEDs can be varied by modulating the duty cycle (DDIM) of this signal using a Schottky
diode with an anode connected to the UDIM pin. The resulting dimmed LED current (IDIM_LED) is given as in
Equation 23.
I DIM _ LED = DDIM ´ I LED (23)
This PWM is generated using the CC2650 MCU, and the operation of this feature has been tested for
PWM frequencies of 1 kHz and 5 kHz. Refer to Section 4 for the set of measurements taken using this
feature.

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2.2.3 100-µA Flicker-Free Dimming

The TIDA-01095 can be used to achieve 100-µA flicker-free output LED current by modifying the value of
the current sensing resistor from 50 mΩ to 250 mΩ. This increases the voltage at the CS pin seen by the
TPS92640/1 device. The LED current measurements (with the 250-mΩ CS resistor and 60-V MOSFEts) in
the range 49 µA to 1 mA are as shown in Table 4.

Table 4. LED Current Measurements

VIN (V) IIN (mA) VOUT (V) IOUT (mA) VIADJ (V)
48.50 0.021 28.53 0.049 0.0697
48.50 0.021 28.71 0.073 0.0703
48.50 0.021 28.86 0.103 0.0709
48.50 0.021 28.96 0.125 0.0713
48.50 0.021 29.01 0.251 0.0717
48.50 0.021 29.16 0.360 0.0725
48.50 0.021 29.27 0.457 0.0731
48.50 0.021 29.36 0.561 0.0737
48.50 0.021 29.42 0.640 0.0741
48.50 0.021 29.51 0.758 0.0747
48.50 0.021 29.56 0.846 0.0752
48.50 0.021 29.63 0.975 0.0758
48.50 0.021 29.66 1.020 0.0760

1.2

0.8
IOUT (mA)

0.6

0.4

0.2

0
0.069 0.07 0.071 0.072 0.073 0.074 0.075 0.076 0.077
VIADJ (V) D001

Figure 12. IOUT (mA) versus VIADJ (V)

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3 Getting Started Hardware and Software

3.1 Hardware Connections

Figure 13 shows the hardware interconnections and wireless connections required for the TIDA-01095 to
work as expected. As mentioned earlier, the TIDA-01095 connects as a BoosterPack™ Plug-in Module
upon the CC2650 LaunchPad. Both of these boards should be assembled as shown in Figure 14.
TIDA-01095

DC Input Voltage

PWM DIM

USB Power and

Debug to PC

(PWM Input) for

Analog DIM

Constant current
LED Output

OPT3001
EVM

CC2650 /DXQFK3DGŒ
LED mounted on heat sink to which
LMT84 LMT84 is affixed

CC2541 BLE Device

Monitor (Connected to USB Port)
Control Option 1
Control Option 2

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Figure 13. Hardware Connections

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Figure 14. Board Assembly

Along with the TIDA-01095 and CC2650, either a BLE dongle—the TI CC2540 USB dongle has been
used in this design—or a Bluetooth-enabled phone (with a BLE scanner application) is required to control
the dimming setting of the LED.

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3.2 Firmware

3.2.1 Compiling Project in CCS™ Software

The provided project files require TI's Code Composer Studio™ (CCS) software (verified with v6) and the
BLE software stack (BLE-STACK V2.1.0, which must be downloaded from the BLE software stack archive
at http://www.ti.com/tool/BLE-STACK-ARCHIVE. Any other installed versions of BLE software stack will
require uninstallation). After installing CCS and BLE Stack, the compilation can be done as follows. The
following instructions assume that CCS and BLE Stack are installed in the directory C:\ti\, the default
installation directory.
1. Download the project from <URL>.
2. Open CCS and select (create) an existing (new) workspace.
3. Import the example project SimpleBLEPeripheral from
C:\ti\simplelink\ble_cc26xx_2_01_01_44627\Projects\ble\SimpleBLEPeripheral\CC26xx\CCS\SimpleBL
EPeripheral.
4. Import the example project SimpleBLEPeripheralStack from
C:\ti\simplelink\ble_cc26xx_2_01_01_44627\Projects\ble\SimpleBLEPeripheral\CC26xx\CCS\SimpleBL
EPeripheralStack.
5. Build SimpleBLEPeripheralStack.
6. After the SimpleBLEPeripheralStack builds successfully without any error:
• Click on SimpleBLEPeripheral → Application (under the Project Explorer tab), and right-click on
simpleBLEPeripheral.C to select properties. Select Resource on the left pane, and click on Edit to
edit the Location. Then, click on File and browse to <directory-name>, and select
simpleBLEPeripheral.C.
• Similarly, click on SimpleBLEPeripheral → Startup, and right-click on main.C to select properties.
Select Resource on the left pane, and click on Edit to edit the Location. Then, click on File and
browse to <directory-name>, and select main.C.
7. Click on SimpleBLEPeripheral → Startup and open Board.C. Modify the file by adding the following two
lines at line number 64 below the comment.
64 #if defined(LED_Dimmer_CC2650LP)
65 #include "LED_Dimmer/Board.c"
Change the #if directive in the following line (number 66) to #elif. Save the file.
8. Right-click on SimpleBLEPeripheral to open properties.
(a) Select the General option in the left pane. Under the Main tab, tick Manage the project’s target-
configuration automatically and select Texas Instruments XDS110 USB Debug Probe as
Connection.
(b) Click on Include Options under ARM Compiler in the left pane. Click on the Add icon to add the
directory path. Click on Browse and add the path to the <directory-name>. Similarly, add <directory-
name>\LED_Dimmer.
(c) Select Advanced Options → Predefined Symbols from the left pane in the Properties dialogue box.
Add the symbol TI_DRIVERS_I2C_INCLUDED (if not already present in the list) by clicking on the
Add icon, typing in TI_DRIVERS_I2C_INCLUDED, and clicking OK. Also, in the same list, modify
the TI_DRIVERS_LCD_INCLUDED entry (if present) by clicking on the Edit icon and typing
xTI_DRIVERS_LCD_INCLUDED.
(d) Finally, click OK to close the Properties dialogue box.
9. Right-click on SimpleBLEPeripheral → Drivers and select New → Folder. Click on Advanced, and then
Link to alternate location (Linked Folder). Browse to <directory-name>\i2c and Finish.
10. Right-click on SimpleBLEPeripheral and navigate to Folder under New to create a new folder. Leave
the default parent folder SimpleBLEPeripheral unchanged, type in the Folder Name as LEDService,
and click OK.
11. Right-click on SimpleBLEPeripheral and select Add Files.
12. Navigate to <directory-name>\LED_Dimmer. Select all the .C files except Board.C. Select Copy Files
in the next dialogue box and click OK. Move all the added files to LEDService by first selecting all the
files and then right-clicking to select Move to SimpleBLEPeripheral → LEDService.

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13. Right-click on SimpleBLEPeripheral and Clean Project.

14. Build the project.

3.2.2 Using the BLE Device Monitor

The Bluetooth low energy (BLE) Device Monitor is a Windows® application that displays services,
characteristics, and attributes of any BLE device. The BLE Device Monitor requires a CC2540USB dongle
with a HostTestApplication to work. The BLE Device Monitor has been tested on Windows 7 and Windows
8. BLE Device Monitor is used to connect to the CC2650 LaunchPad to read OPT3001 and LMT84 sensor
values, as well as giving the PWM inputs for controlling the LED dimming level. Refer to the BLE Device
Monitor User Guide [1] to get started with the BLE Device Monitor and using it to connect to other BLE
devices.
Figure 15 is a screenshot of the BLE Device Monitor showing the SimpleBLEPeripheral in the BLE
Network tab.

Figure 15. SimpleBLEPeripheral in BLE Network Tab

From the BLE Device Monitor, the frequency and the duty cycle of the two PWMs can be controlled.
These values are required to be given in the HEX format. For setting the time period (frequency) of the
PWM, the total counts (as mentioned in Section 2.2.1) should be entered; for example, 48000, which is
BB,80 in HEX, for 1 kHz. Similarly, the duty cycle should be in proportion to the counts, such as 24000,
which is 5D,C0 in HEX, for 50% duty cycle at 1 kHz.
The temperature sensor characteristic is a read-only characteristic, and shows the output of the ADC
onboard the CC2650 after converting the analog output of the LMT84 in big-endian format. If the
characteristic shows 06:03, then the output of the ADC is 0306, which evaluates to 774 in decimal. The
CC2650 ADC has a 12-bit ADC with a reference voltage of 4.3 V. Thus, the analog voltage value this
corresponds to is 4.3 × 774 / (212 – 1) = 0.812 V. From the mapping table in the LMT84 data sheet, the
temperature is 41°C.
The OPT3001 is interfaced to the CC2650 using I2C. From the BLE Device Monitor, the sensor can be
enabled or put to sleep. As shown in Figure 15, enabling the OPT3001 notifications causes the sensor
values to appear in the Event Log. The duration after which the OPT3001 value is read can also be
controlled.

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4 Test Data (With 60-V MOSFETs)

In this section, the measurements have been performed with 60-V MOSFETs used for Q1 and Q2, as
mentioned in Section 2.1.12.

4.1 Efficiency and Output Current With IADJ Feature

NOTE: The µC duty is the duty cycle of the PWM input fed into the DAC filter circuit (see Table 5).

Table 5. Efficiency and Output Current

VIN (V) IIN (A) PIN (W) VOUT (V) IOUT (A) POUT (W) EFFICIENCY VIADJ (V)
47.86 2.380 113.907 41.55 2.699 112.143 98.45% 1.508
47.94 2.078 99.619 40.56 2.416 97.993 98.37% 1.358
48.01 1.790 85.938 39.60 2.132 84.427 98.24% 1.208
48.08 1.513 72.745 38.64 1.845 71.291 98.00% 1.057
48.15 1.248 60.091 37.68 1.557 58.668 97.63% 0.907
48.22 0.994 47.931 36.69 1.269 46.560 97.14% 0.756
48.28 0.752 36.307 35.66 0.979 34.911 96.16% 0.605
48.34 0.520 25.137 34.58 0.689 23.826 94.78% 0.454
48.40 0.299 14.472 33.32 0.400 13.328 92.10% 0.302
48.45 0.094 4.554 31.71 0.112 3.552 77.98% 0.15151
48.45 0.074 3.585 31.52 0.083 2.616 72.97% 0.13622
48.46 0.055 2.665 31.24 0.055 1.718 64.47% 0.12113
48.46 0.037 1.793 30.87 0.028 0.864 48.21% 0.10605
48.47 0.028 1.357 30.68 0.015 0.460 33.91% 0.09842
48.47 0.021 1.018 30.16 0.005 0.139 13.63% 0.09099
48.47 0.018 0.872 28.98 0.00022 0.006 0.73% 0.09056
48.47 0.018 0.872 26.96 0.00000 0.000 0.00% 0.07590

2.5

2
IOUT (A)

1.5

0.5

0
0 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6
VIADJ (V) D002

Figure 16. IOUT (A) versus VIADJ (V)

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100%

90%

80%

70%

60%
Efficiency

50%

40%

30%

20%

10%

0
0 0.5 1 1.5 2 2.5 3
IOUT (A) D003

Figure 17. Efficiency versus IOUT (A)

1001

1000.5

1000

999.5
IOUT (mA)

999

998.5

998

997.5
4818 4820 4822 4824 4826 4828 4830 4832
PC PWM Count for IADJ Ref PWM DAC D004

Figure 18. IOUT (A) versus µC Counts

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4.2 Efficiency and Output Current With UDIM Feature

4.2.1 At 1-kHz UDIM Frequency

VOUT = 41.8 V is the forward voltage drop for IOUT = 2.764 A (see Table 6).

Table 6. Efficiency and Output Current at 1-kHz UDIM Frequency

UDIM DUTY
VIN (V) IIN (A) PIN (W) IOUT (A) POUT (W) EFFICIENCY
CYCLE
47.86 2.451 117.305 2.764 115.535 98.49% 100%
47.93 2.196 105.254 2.476 103.497 98.33% 90%
47.99 1.956 93.868 2.200 91.960 97.97% 80%
48.05 1.715 82.406 1.924 80.423 97.59% 70%
48.11 1.472 70.818 1.647 68.845 97.21% 60%
48.18 1.229 59.213 1.371 57.308 96.78% 50%
48.24 0.984 47.468 1.094 45.729 96.34% 40%
48.30 0.738 35.645 0.818 34.192 95.92% 30%
48.37 0.490 23.701 0.540 22.572 95.24% 20%
48.43 0.239 11.575 0.262 10.952 94.62% 10%
48.46 0.114 5.524 0.123 5.141 93.07% 5%
48.47 0.089 4.314 0.096 4.013 93.02% 4%
48.48 0.063 3.054 0.068 2.842 93.06% 3%
48.48 0.035 1.697 0.038 1.588 93.61% 2%
48.49 0.013 0.630 0.013 0.543 86.20% 1%
48.49 0.006 0.291 0.003 0.125 43.10% 0.5%
48.49 0.002 0.097 0.00000 0.000 0.00% 0.1%

2.5

2
IOUT (A)

1.5

0.5

0
0 20% 40% 60% 80% 100%
UDIM Duty Cycle D005

Figure 19. IOUT (A) versus UDIM Duty Cycle

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100%

90%

80%
Efficiency

70%

60%

50%

40%
0 0.5 1 1.5 2 2.5 3
IOUT (A) D006

Figure 20. Efficiency versus IOUT (A)

4.2.2 At 5-kHz UDIM Frequency

VOUT = 41.59 V is the forward voltage drop for IOUT = 2.701 A (see Table 7).

Table 7. Efficiency and Output Current at 5-kHz UDIM Frequency

UDIM DUTY
VIN (V) IIN (A) PIN (W) IOUT (A) POUT (W) EFFICIENCY
CYCLE
47.86 2.377 113.763 2.701 112.335 98.74% 100%
47.93 2.092 100.270 2.375 98.776 98.51% 90%
47.99 1.853 88.925 2.098 87.256 98.12% 80%
48.06 1.616 77.665 1.825 75.902 97.73% 70%
48.12 1.383 66.550 1.560 64.880 97.49% 60%
48.18 1.145 55.166 1.288 53.568 97.10% 50%
48.24 0.903 43.561 1.013 42.131 96.72% 40%
48.31 0.665 32.126 0.747 31.068 96.71% 30%
48.37 0.420 20.315 0.472 19.630 96.63% 20%
48.43 0.165 7.991 0.193 — — 10%
48.44 0.141 6.830 0.164 — — 9%
48.45 0.116 5.620 0.137 — — 8%
48.45 0.094 4.554 0.111 — — 7%
48.46 0.072 3.489 0.087 — — 6%
48.46 0.054 2.617 0.065 — — 5%
48.46 0.038 1.841 0.046 — — 4%
48.47 0.023 1.115 0.028 — — 3%
48.47 0.013 0.630 0.014 — — 2%
48.47 0.006 0.291 0.0041 — — 1%
48.47 0.005 0.242 0.0033 — — 0.9%
48.47 0.002 0.097 0.00000 — — 0.1%

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100%

98%

96%

94%

92%
Efficiency

90%

88%

86%

84%

82%

80%
0 0.5 1 1.5 2 2.5 3
IOUT (A) D007

Figure 21. Efficiency versus IOUT (A)

2.5

2
IOUT (A)

1.5

0.5

0
0 20% 40% 60% 80% 100%
UDIM Duty Cycle D008

Figure 22. IOUT (A) versus UDIM Duty Cycle

NOTE: In the preceding tables, there is a lack of output voltage for various duty cycles. The
efficiency is calculated by multiplying the average value of output current (maximum ≈2.8 A
during ON time and 0 A during the OFF time) with the value of the output voltage during the
ON time to get the average power. This value of output voltage during the ON time is the
same, regardless of the duty cycle.

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4.3 Line Regulation

Table 8. Line Regulation

VIN (V) IIN (A) PIN (W) VOUT (V) IOUT (A) POUT (W) EFFICIENCY
42.31 1.902 80.474 39.43 2.023 79.767 99.12%
43.34 1.862 80.699 39.4 2.026 79.824 98.92%
44.35 1.823 80.85 39.39 2.027 79.844 98.76%
45.32 1.786 80.942 39.38 2.028 79.863 98.67%
46.42 1.745 81.003 39.35 2.03 79.881 98.61%
47.36 1.712 81.08 39.34 2.031 79.9 98.54%
48.4 1.679 81.264 39.33 2.034 79.997 98.44%
49.36 1.648 81.345 39.32 2.036 80.056 98.41%

100%

99%

98%
Efficiency

97%

96%

95%
41 42 43 44 45 46 47 48 49 50
VIN (V) D009

Figure 23. Efficiency versus Input Voltage (V)

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4.4 OPT3001 – LUX Measurements

Table 9. LUX Measurements

MSB IN LSB IN
IOUT (A) DATA IN HEX MSB MULTIPLIER DEC TO HEX LUX
DECIMAL DECIMAL
2.699 BBEA B 11 20.48 48106 3050 62464
2.416 BB48 B 11 20.48 47944 2888 59146
2.132 BA85 B 11 20.48 47749 2693 55153
1.845 B993 B 11 20.48 47507 2451 50196
1.557 B87C B 11 20.48 47228 2172 44483
1.269 AE72 A 10 10.24 44658 3698 37868
0.979 AB9E A 10 10.24 43934 2974 30454
0.689 A87B A 10 10.24 43131 2171 22231
0.400 9A31 9 9 5.12 39473 2609 13358
0.112 7BAF 7 7 1.28 31663 2991 3828
0.083 78A9 7 7 1.28 30889 2217 2838
0.055 6B7C 6 6 0.64 27516 2940 1882
0.028 5BA3 5 5 0.32 23459 2979 953
0.015 4B99 4 4 0.16 19353 2969 475
0.005 2CCE 2 2 0.04 11470 3278 131
0.00022 1Ba2 1 1 0.02 7074 2978 60
0.00000 00D9 0 0 0.01 217 217 2

80000

70000

60000
Illuminance (Lux)

50000

40000

30000

20000

10000

0
0 0.5 1 1.5 2 2.5 3
IOUT (A) D010

Figure 24. Illuminance versus IOUT (A)

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4.4.1 Conversion From OPT3001 Sensor Reading in HEX to LUX

The HEX reading obtained from the OPT3001 sensor can be converted into LUX as follows.
Example, 4CB1:
1. Extract the most significant nibble, four in this example, and calculate LSB_size as:
LSB_size = 0.01 × 24 = 0.16.
This nibble may even be A, B, C, D, E, or F, in which case the exponent should be taken as the
corresponding decimal number; that is, 10, 11, 12, 13, 14, and 15, respectively.
2. Convert the remaining three least significant nibbles into decimals and multiply by LSB_size to get the
LUX value.
CB1h = 3249d
LUX = 3249 × 0.16 = 519.84

4.5 Waveforms at 1-kHz UDIM Frequency

Figure 25. LED Current, LED Voltage, and UDIM Input Waveforms at 99% Duty Cycle

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Figure 26. LED Current, LED Voltage, and UDIM Input Waveforms at 50% Duty Cycle

Figure 27. LED Current, LED Voltage, and UDIM Input Waveforms at 2% Duty Cycle

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4.6 Thermal Image at Full Load

Figure 28 shows the thermal image of the TIDA-01095 board at the full load condition.

Figure 28. Thermal Image at Full Load

Table 10. Temperature Values

TEMPERATURE (°C) DEVICE NAME
56.30 High Side MOSFET
55.20 Current Sensing Resistor
54.30 Low Side MOSFET
49.10 TPS92641 IC
30.60 Inductor
27.40 Surface temperature

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5 Test Data (With 100-V MOSFETs)

In this section, the measurements are done with 100-V MOSFETs used for Q1 and Q2, as mentioned in
Section 2.1.12.

5.1 Efficiency and Output Current With IADJ Feature

Table 11 shows the efficiency and output current results.

Table 11. Efficiency and Output Current

VIN (V) IIN (A) PIN (W) VOUT (V) IOUT (A) POUT (W) EFFICIENCY VIADJ (V)
47.94 2.432 116.590 41.71 2.750 114.703 98.38% 1.545
48.02 2.120 101.802 40.67 2.461 100.089 98.32% 1.391
48.09 1.825 87.764 39.68 2.172 86.185 98.20% 1.237
48.16 1.542 74.263 38.71 1.881 72.814 98.05% 1.082
48.23 1.272 61.349 37.74 1.589 59.969 97.75% 0.928
48.29 1.014 48.966 36.77 1.297 47.691 97.40% 0.773
48.35 0.766 37.036 35.72 1.002 35.791 96.64% 0.619
48.41 0.531 25.706 34.60 0.707 24.462 95.16% 0.464
48.47 0.308 14.929 33.35 0.413 13.774 92.26% 0.309
48.52 0.101 4.901 31.75 0.121 3.842 78.39% 0.154
48.52 0.080 3.882 31.57 0.091 2.873 74.01% 0.140
48.52 0.062 3.008 31.29 0.063 1.971 65.53% 0.125
48.52 0.042 2.038 30.92 0.035 1.082 53.11% 0.109
48.53 0.025 1.213 30.29 0.009 0.273 22.47% 0.093
48.53 0.020 0.971 29.42 0.0007 0.021 2.12% 0.0086
48.53 0.019 0.922 28.94 0.0002 0.006 0.63% 0.00084
48.54 0.020 0.971 27.76 0.0000 0.000 0.00% 0.00770

2.5

2
IOUT (A)

1.5

0.5

0
0 0.5 1 1.5 2
VIADJ (V) D011

Figure 29. IOUT (A) versus VIADJ (V)

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100%

90%

80%

70%

60%
Efficiency

50%

40%

30%

20%

10%

0
0 0.5 1 1.5 2 2.5 3
IOUT (A) D012

Figure 30. Efficiency versus IOUT (A)

5.2 Efficiency and Output Current With UDIM Feature

5.2.1 At 1-kHz UDIM Frequency

VOUT = 41.74 V is the forward voltage drop for IOUT = 2.759 A (see Table 12).

Table 12. Efficiency and Output Current at 1-kHz UDIM Frequency

UDIM DUTY
VIN (V) IIN (A) PIN (W) IOUT (A) POUT (W) EFFICIENCY
CYCLE
47.88 2.442 116.923 2.759 115.161 98.49% 100%
47.93 2.186 104.775 2.469 103.056 98.36% 90%
47.98 1.944 93.273 2.192 91.494 98.09% 80%
48.04 1.704 81.860 1.917 80.016 97.75% 70%
48.09 1.462 70.308 1.641 68.495 97.42% 60%
48.14 1.220 58.731 1.364 56.933 96.94% 50%
48.20 0.976 47.043 1.089 45.455 96.62% 40%
48.25 0.731 35.271 0.813 33.935 96.21% 30%
48.30 0.486 23.474 0.537 22.414 95.49% 20%
48.35 0.236 11.411 0.260 10.852 95.11% 10%
48.39 0.112 5.420 0.122 5.092 93.96% 5%
48.39 0.086 4.162 0.094 3.924 94.28% 4%
48.40 0.061 2.952 0.065 2.713 91.89% 3%
48.40 0.036 1.742 0.038 1.586 91.03% 2%
48.41 0.013 0.629 0.013 0.543 86.22% 1%
48.41 0.006 0.290 0.003 0.125 43.11% 0.5%
48.41 0.005 0.242 0.002 0.083 34.49% 0.4%
48.41 0.004 0.194 0.000 0.000 0.00% 0.3%

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100%

90%

80%

70%

60%
Efficiency

50%

40%

30%

20%

10%

0
0 0.5 1 1.5 2 2.5 3
IOUT (A) D013

Figure 31. Efficiency versus IOUT (A)

2.5

2
IOUT (A)

1.5

0.5

0
0 20% 40% 60% 80% 100%
UDIM Duty Cycle D014

Figure 32. IOUT (A) versus UDIM Duty Cycle

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5.2.2 At 5-kHz UDIM Frequency

VOUT = 41.76 V is the forward voltage drop for IOUT = 2.753 A (see Table 13).

Table 13. Efficiency and Output Current at 5-kHz UDIM Frequency

UDIM DUTY
VIN (V) IIN (A) PIN (W) IOUT (A) POUT (W) EFFICIENCY
CYCLE
47.89 2.442 116.947 2.753 114.965 98.31% 100%
47.98 2.136 102.485 2.413 100.767 98.32% 90%
48.04 1.897 91.132 2.139 89.325 98.02% 80%
48.10 1.652 79.461 1.858 77.590 97.65% 70%
48.17 1.405 67.679 1.578 65.897 97.37% 60%
48.23 1.164 56.140 1.307 54.580 97.22% 50%
48.30 0.916 44.243 1.026 42.846 96.84% 40%
48.36 0.668 32.304 0.748 31.236 96.69% 30%
48.43 0.422 20.437 0.474 19.794 96.85% 20%
48.49 0.160 7.758 0.188 — — 10%
48.50 0.141 6.839 0.165 — — 9%
48.50 0.117 5.675 0.140 — — 8%
48.51 0.095 4.608 0.112 — — 7%
48.51 0.074 3.590 0.088 — — 6%
48.52 0.055 2.669 0.065 — — 5%
48.52 0.039 1.892 0.047 — — 4%
48.53 0.024 1.165 0.028 — — 3%
48.53 0.013 0.631 0.014 — — 2%
48.53 0.006 0.291 0.0043 — — 1%
48.53 0.005 0.243 0.0035 — — 0.9%
48.53 0.005 0.243 0.0028 — — 0.8%
48.53 0.002 0.097 0 — — 0.1%

100%

98%

96%

94%

92%
Efficiency

90%

88%

86%

84%

82%

80%
0 0.5 1 1.5 2 2.5 3
IOUT (A) D015

Figure 33. Efficiency versus IOUT (A)

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2.5

2
IOUT (A)

1.5

0.5

0
0 20% 40% 60% 80% 100%
UDIM Duty Cycle D016

Figure 34. IOUT (A) versus UDIM Duty Cycle

5.3 Line Regulation

Table 14 shows the line regulation results.

Table 14. Line Regulation

VIN (V) IIN (A) PIN (W) VOUT (V) IOUT (A) POUT (W) EFFICIENCY
41.82 0.816 34.125 35.57 0.931 33.116 97.04%
42.84 0.799 34.229 35.56 0.931 33.106 96.72%
43.8 0.783 34.295 35.56 0.932 33.142 96.64%
44.82 0.766 34.332 35.55 0.933 33.168 96.61%
45.84 0.751 34.426 35.55 0.934 33.204 96.45%
46.8 0.737 34.492 35.55 0.935 33.239 96.37%
47.82 0.722 34.526 35.55 0.935 33.239 96.27%
48.78 0.709 34.585 35.55 0.936 33.275 96.21%

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100%

99%

98%
Efficiency

97%

96%

95%
40 42 44 46 48 50 52
VIN (V) D0017

Figure 35. Efficiency versus Input Voltage (V)

5.4 LMT84 – Temperature Measurements

Table 15 shows the temperature measurements of the LED COBs heatsink using the LMT84 device with
different LED currents. The temperature rise with a load is dependent on the size of the heatsink and the
velocity of air flow from the cooling fan.

Table 15. Temperature Measurements

LMT84
ADC OUTPUT HEX TO
VIN (V) IIN (A) VOUT (V) IOUT (A) TEMP (°C) OUPUT
(HEX) DECIMAL
VOLTAGE
48.00 0.061 31.55 0.063 27.62 349 841 0.883
47.91 0.320 33.71 0.420 29.91 33D 829 0.871
47.73 0.802 36.16 1.019 32.96 32D 813 0.854
47.65 1.009 37.03 1.258 34.49 325 805 0.845
47.57 1.225 37.86 1.497 36.02 31D 797 0.837
47.49 1.448 38.68 1.735 37.93 313 787 0.826
47.40 1.679 39.36 1.972 39.45 30B 779 0.818
47.32 1.918 40.26 2.208 40.41 306 774 0.813

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45

41
Temperature (qC)

37

33

29

25
0 0.5 1 1.5 2 2.5
IOUT (A) D018

Figure 36. Temperature (°C) versus IOUT (A)

If the output of the ADC is 0306, then it evaluates to 774 in decimal. The CC2650 has a 12-bit ADC with a
reference voltage of 4.3 V. Thus, the analog voltage value 0306 corresponds to is:
(4.3 × 774) / (212 – 1) = 0.812 V. From the mapping table in the LMT84 data sheet, the temperature is
41°C.

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6 Design Files

6.1 Schematics
To download the schematics, see the design files at TIDA-010905.

6.2 Bill of Materials

To download the bill of materials (BOM), see the design files at TIDA-010905.

6.3 PCB Layout Recommendations

The performance of any switching converter depends as much upon the layout of the PCB as the
component selection. Follow a few simple guidelines to maximize noise rejection and minimize the
generation of EMI within the circuit.
Discontinuous currents are the most likely to generate EMI, therefore take care when routing these paths.
The main path for discontinuous current in the TPS92640 and TPS92641 buck converters contains the
input capacitor (CIN), the low-side MOSFET (QLS), and the high-side MOSFET (QHS). This loop should
be kept as small as possible, and the connections between all three components should be short and thick
to minimize parasitic inductance. In particular, the switch node (where L, QLS, and QHS connect) should
be just large enough to connect the components without excessive heating from the current it carries. The
current sense trace (CS pin) should be run along with a ground plane or have differential traces run for CS
and ground.
In some applications, the LED or LED array can be far away (several inches or more) from the circuit, or
on a separate PCB connected by a wiring harness. When an output capacitor is used and the LED array
is large or separated from the rest of the converter, the output capacitor should be placed close to the
LEDs to reduce the effects of parasitic inductance on the AC impedance of the capacitor.

6.3.1 Layout Prints

To download the layer plots, see the design files at TIDA-010905.

6.4 Altium Project

To download the Altium project files, see the design files at TIDA-010905.

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6.5 Layout Guidelines

The ground plane of the CC2650 LaunchPad and the LED driver section are separated clearly and
connected through a 0-Ω resistor.

Figure 37. Layout Guidelines 1

Figure 38. Layout Guidelines 2

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6.6 Gerber Files

To download the Gerber files, see the design files at TIDA-010905.

6.7 Assembly Drawings

To download the assembly drawings, see the design files at TIDA-010905.

7 Software Files
To download the software files, see the design files at TIDA-010905.

8 References
1. Texas Instruments, BLE Device Monitor User Guide, TI Wiki
(http://processors.wiki.ti.com/index.php/BLE_Device_Monitor_User_Guide)
2. Texas Instruments, Dimming Techniques for Switched-Mode LED Drivers, LM3406/LM3409 Application
Report (SNVA605)
3. Texas Instruments, Ringing Reduction Techniques for NexFET High Performance MOSFETs,
Application Report (SLPA010)
4. Texas Instruments, Microcontroller PWM to 12-Bit Analog Out, TIPD127 User's Guide (TIDU027)
5. Texas Instruments, WEBENCH® Design Center, (http://www.ti.com/webench)

9 About the Authors

SEETHARAMAN DEVENDRAN is a Systems Architect at Texas Instruments, where he is responsible for
developing reference design solutions for the industrial segment. Seetharaman brings to this role his
extensive experience in analog and mixed signal system-level design expertise. Seetharaman earned his
Bachelor’s degree in Electrical Engineering (BE, EEE) from Thiagarajar College of Engineering, Madurai,
India.
MUSTAFA LOKHANDWALA is an undergraduate student at the Indian Institute of Technology Bombay
(IITB), where he is pursuing a Bachelor of Technology (BTech) in Electrical Engineering. His areas of
interest include design and debug of circuits and systems, as well as hardware product development.
VENKATADRI SHANTARAM is a Field Applications Engineer at Texas Instruments, where he is
responsible for supporting customers across various end equipments; primarily supporting customers on
Texas Instruments Embedded processors, microcontrollers, and wireless SoCs. Venkatadri earned his
Bachelor’s degree in Electrical Engineering (BE, EEE) from The National Institute of Engineering, Mysore,
India.

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Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.

Changes from Original (June 2016) to A Revision ......................................................................................................... Page

• Changed the design guide from preview to active and added remaining material ............................................... 1

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