IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 39, NO.

4, APRIL 1991
643
Tunable Microwave and Millimeter-Wave
Band-Pass Filters
J aroslaw Uher, Member, IEEE, and Wolfgang J . R. Hoefer, Fellow, IEEE
general design principles are described. Recent progress in the perfor-
mance of various tunable filters is reported. The paper surveys magneti-
cally tunable filters (ferrimagnetic resonance filters, MSW filters,
evanescent waveguide filters, E-plane printed circuit filters), electroni-
cally tunable filters, and mechanically tunable filters. The typical perfor-
mance parameters are summarized and compared in terms of suitability
for different applications.
Abstract -This paper presents an overview of tunable microwave and
millimeter-wave hand-pass filters realized in different technologies. Some
I. INTRODUCTION
UNABLE band-pass filters can be realized in many
T techniques, but whatever the method of tuning may be,
they must conserve as much as possible their transmission
and reflection characteristics over a given tuning range. The
tuning can be accomplished by varying either the length or
the inductive or capacitive loading of the resonators. Re-
search on tunable filters has spanned more than three
decades, resulting in a large number of attractive realizations
and applications.
Fig. 1 shows a block diagram of the proposed common
spare payload for European DBS systems. The two output
multiplexers can be tuned via telecommand, thus allowing
broadcasts in any frequency band allocated for DBS operat-
ing countries. The manifold multiplexers incorporate five
four-pole elliptic filters which are mechanically tunable, with
a tuning range of +/ - 40 MHz [Sl]. Fig. 2 demonstrates
another typical application of tunable filters in switched
tracking preselectors-mixers [151. This circuit contains a
three-sphere YIG-tuned preselector and a single YI G sphere
which acts as a discriminator generating an error signal to
lock the preselector frequency. Further common applications
for tunable band-pass filters include frequency hopped re-
ceivers, Doppler radar, and troposcatters.
The above examples demonstrate that the spectrum of
applications for tunable filters includes all major areas of
microwave engineering. Obviously, different requirements
for microwave systems have led to the development of vari-
ous types of tunable filters with a performance matched to
the system demands. Most tunable filters described in the
literature fall into three basic types: mechanically tunable,
H
Rx / Tx
Ant enna
LHCP
-
Broadband Dl pl exer
TUMUX
channel i zed
sect i on
Fig. I . Block diagramof proposed common spare payload for Euro-
pean DBS containing tunable output multiplexers (after [Sl]).
magnetically tunable, and electronically tunable filters. Since
the number of problems associated with the theory and
design of tunable filters is extremely large, it is not possible
to deal with all of them in a single paper. Moreover, some of
the tuning methods described in the literature are technically
extremely complex (e.g. plasma-dielectric multilayer struc-
tures in continuously varied electric field [54]) and therefore
they have never found practical application. The intention of
this paper, therefore, is to highlight only the most important
innovations in tunable filters, emphasizing design theory and
the resulting improvement in performance.
Manuscript received February 26, 1990; revised December 10, 1990.
J . Uher was with the Electrical Engineering Department, University
of Ottawa, Ottawa, Ontario, Canada. He is now with Spar Aerospace
Limited, 21025 Trans-Canada Highway, Ste-Anne-de-Bellevue, Quebec,
Canada H9X 3R2.
W. J , R, Hoefer is wi th the Electrical Engi neeri ng Department,
University of Ottawa, Ottawa, Ontario, Canada KI N 6NS.
I EEE Log Number 9042495.
11. THEORY OF TUNABLE MICROWAVE FILTERS
A. Definitions
described by the same criteria as used for high-quality fixed
components (low insertion loss, high selectivity, high dynamic
range) and additionally by certain parameters relevant only
The performance of tunable band-pass filters may be
001 8-9480/Y 1 /0400-0643$01 .OO 0 1 YO1 IEEE
644 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 39, NO. 4, APRIL 1991
TO RF FIRST CONVERTOR 321.4 MHz IF OFFSET COIL
U
-
1)ISCRIMINATOR
OUTPUT
RF INPUT I I I
DC-22 GHZ SYTF MD
A m
GHz LO
x1, x2, x4
3- 6. 7
GHz LO
Fig. 2. Magnetically tunable YIG filter in switched-tracking preselec-
tor mixer.
to tunable filters. The most important of them are:
Tuning range-defined as the difference between the
lowest and the highest midband frequency which is achiev-
able within acceptable limits for insertion loss, bandwidth,
and response distortion.
Tuning speed-defined as the time which is necessary to
change the filter response to another steady state by the
unit frequency shift.
Tuning linearity-the maximum deviation of the center
frequency versus a parameter which enforces the tuning
(coil current, voltage, static magnetic field, or resonator
length variation) from a best fit straight line over the
specified operating frequency range.
Tuning sensitivity or tuning efficiency-this can be written
as
where f o( X2) and f o ( X, ) are center frequencies corre-
sponding to X2 and X I ; Xi is the variable enforcing the
tuning.
B. Design Principles and Example
Even though each type of tunable filter has its specific
theoretical description, some universal design principles can
be formulated. The basic requirements which all tunable
filters must satisfy are a constant filter response shape and
constant bandwidth over the tuning range. For narrow-
band-pass filters, with direct inductive coupling, two inde-
pendent conditions for constant bandwidth and constant
response shape have been derived [l]. I f the conditions for
constant response and bandwidth are derived from the ex-
pressions for the external Q factor of the end resonators,
they take the following form:
where (x,), and ( x n ) , are the resonator slope parameter
values at the mean tuning frequency (f,),.
However, if the same condition is calculated from the
expressions for coupling coefficients between resonators, a
different formulation is obtained:
(3)
The contradiction in these two equations is due to different
reference points in calculating the slope parameters. Equa-
tion (2) refers to the slope parameters of resonators l and n
"seen" from the end-coupling reactances Xol and X, , , , ,,
while (3) expresses the slope parameters which are "seen"
from the interresonator reactances X, , and Xn- l , n. In order
to reconcile conditions ( 2) and (3) and thus minimize band-
width and change of response shape across the tuning range,
coupling reactances X, , and Xn, n+l must be different from
Good examples of tunable band-pass filters where the
number of simplifying assumptions in the design theory is
minimized are magnetically tunable printed circuit filters.
The features of various E-plane filter types are discussed in
another part of this paper. In this section, the design of a
ferrite-loaded single metal insert filter (Fig. 3(a)) will be
considered. A fixed metal insert band-pass filter belongs to
the class of filters with direct inductively coupled cavities and
has been analyzed extensively in the literature [41-[61. To
design tunable filters with resonators partially filled with
E-plane ferrite slabs, one must consider some additional
aspects. Although the basic synthesis from a lumped/distrib-
uted element low-pass prototype followed by computer-aided
optimization remains unchanged, the optimization goal will
be specific to tunable filters. In this step it will be required
that the tuning range become maximum and that other
performance parameters (insertion loss, 3 dB bandwidth
variation, 40 dB bandwidth variation) not exceed acceptable
limits.
The filter is synthesized using the formalism derived by
Cohn [7] and generalized by Young [8], Levy [91, and Rhodes
[lo]. The equivalent circuit of an E-plane inductive septum
(Fig. 3(b)) was introduced in [4] and will be used to realize
the required impedance inverter. As design specifications we
assume a lower and upper band-edge frequency for the mid
tuning band (f / )m, (f , , ), n, out-of-band rejection L (dB), and
ripple ( E ) , . Then the design procedure can be summarized
as follows:
at the
XI2 and Xn-1.n.
1) Compute the midband guide wavelength
mid tuning frequency by solving
where ( A, / , ) l , , and ( A Xl / ) , p , are the guide wavelengths in
the resonators at ( f / - ) , ?, and ( f H) m. The above equation
can be solved numerically after finding roots of the
complex transcendental equation associated with the
five-layer geometry shown in Fig. 3(c). The method for
solving such an equation has been discussed in [36]. At
UHER AND HOEFER: TUNABLE MICROWAVE AND MILLIMETER-WAVE BAND-PASS FILTERS 645
n I n
W
t I I
Fig. 3. Magnetically tunable E-plane metal insert filter: (a) general
topology; (b) equivalent circuit of impedance inverter; (c) ferrite loaded
resonator section.
this point a suitable ferrite material and slab geometry
must be determined. For simplicity, the air gap be-
tween the ferrite slab and the narrow waveguide wall
(due to the nonzero radius of the waveguide corner)
should be kept constant while the ferrite slab thickness
(wf), as well as the material data ( MS, e r ) , should be
varied. For appropriate selection of the resonator load-
ing, both the real and the imaginary part of the cavity
eigenvectors must be examined, and the correct solu-
tion must satisfy the following conditions:
The maximum value of the attenuation constant of
the dominant mode corresponds to a power loss factor
of about 0.1 dB/mm (which comes to approximately 1
dB of loss per resonator). The Hdc field at which this
attenuation occurs is the maximum biasing field, and
the corresponding resonant frequency is the upper tun-
ing range limit.
The phase constant variation checked at zero and
maximum bias must be maximized in order to make the
tuning range as wide as possible. The midband guide
wavelength at the center tuning frequency (eq. (4)) can
then be computed.
Calculate the scaling parameter (Y as
=( Ago) , / [ (An),sin ( 4 A n O ) , / ( A , L ) , ) ] . (5)
Determine the number of resonators N.
If T, is the first-kind Chebyshev polynominal of
degree N, and A, is the guide wavelength at the
designated stopband frequency f,, then the number of
resonators can be computed by finding the minimum
value of N for which the most severe constraints on the
rejection L satisfy
L,=lOlog I +(),
i
If an another type of the filter response is required
(e.g. Butterworth, Bessel), (6) must be adequately mod-
ified.
4) Calculate the impedances of the distributed elements
and the normalized impedance of the inverter. Suitable
expressions can be found in [6].
5) Determine the geometry of the coupling sections. US-
ing the equivalent circuit of the impedance inverter
(Fig. 3(b)), we obtain
Kj - =tan - Qj +tan-' xsj) 1,
I i:
Qj = -tan-'(2XPj + ~,,)-tan-'X,~. (7)
After converting the scattering matrix for the dominant
mode of the two-port E-plane septum into an
impedance matrix and equating it to the impedance
matrix of the equivalent T circuit, one obtains
The scattering coefficients are a function of the septum
geometry and have been computed using the mode
matching technique [37], [38]. Therefore, in order to
realize the required inverters, the computation of the S
parameters with variable septum lengths must be re-
peated until these values have been obtained.
6) Compute the lengths of the resonators:
This combined analytical and numerical synthesis procedure
yields excellent starting values for the final optimization of
the filter. For fixed E-plane filters, a similar synthesis proce-
dure leads to a filter response that satisfies all design specifi-
cations [ 6] . However, for tunable filters some discrepancy
between theoretical and experimental curves can be ex-
pectcd. This is mainly due to uncertainty in characterization
of thc ferrite, whose theoretically predicted permeability
646 I EEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 39, NO. 4, APRI L 1991
4 0
3 0
2 0
1 0
1 2 13 1 4 1 5 1 6 1 7
f / GHz
-
Fig. 4. Computer-optimized Ku-band magnetically tunable E-plane
metal insert filter. Frequency response.
tensor parameters do not always reflect the real frequency
response of the material. Some improvement in filter perfor-
mance can be achieved by slightly varying the dimensions of
the resonators and coupling sections. Therefore, an adequate
optimization routine involving also the widening of the tun-
ing range should be implemented in the design procedure.
A wide class of tunable waveguide filters can be designed
in a similar way. The most important steps in the
analytical/numerical procedure are solution of the eigen-
value problem in the resonator and modal S-matrix compu-
tation of the key building blocks of a filter. With this analysis
a number of components can be designed. Such devices
include metal insert filters, multiple insert filters, large-gap
finline filters, inductive/resonant iris filters, evanescent
waveguide filters (with E-plane ferrimagnetic slabs in res-
onators), and mechanically tunable filters. In all other practi-
cal cases (YIG filters, varactor-diode filters) a rigorous field
theory design is extremely complex and has not been pre-
sented so far. However, the filter synthesis from the low-pass
prototype including all mentioned preclusions and limita-
tions also yields satisfactory results as will be shown in some
examples.
Ill. MAGNETICALLY TUNABLE FILTERS
A. YIG Filters
A filter circuit which make use of gyromagnetic coupling
was first developed by deGrasse [12] and was reported in
1958. Since these filter types usually contain single-crystal
YIG spheres in their resonators, the ferrimagnetic resonance
filters are commonly termed YIG filters. However other
low-loss ferrites ((LiFe),,Fe,O, or barium ferrites) can also
be used. YIG filters are perhaps the most popular tunable
microwave filters because of their multioctave tuning range,
very high selectivity, spurious-free response, and compact
size. The most remarkable fact in the operating principle of
YIG filters is that, if the anisotropy effects can be neglected,
the center frequency for a given shape of the resonator does
not depend on its size but only on the biasing magnetic field.
The resonant frequency of a ferrimagnetic resonator is given
by the Kittel equation [13]:
AS the frame algorithm for optimization, an evolution strat-
egy method has been applied 1401. The frequency response of
the filter is computed using a modal S-matrix approach with
exact expressions given in [37] and [38].
Fig. 4 shows the computer-optimized frequency response
of a Ku-band filter design. The design is based on the
following specifications: and Fierstad [161.
where N,' and N,' are the effective demagnetization factors
[14], considered as the measure of magnetic anisotropy; N,,
Ny , and N, are the demagnetization factors in the x , y , and
z directions, respectively. The design principles for multi-
stage YIG filters operating from 0.5 to 40 GHz are described
in monographs [l], [2] and also in the papers by Carter [201
Tuned center frequency: 15 GHz; minimum tuning range:
+/ - 600 MHz; passband: 260 MHz; lower rejection: 40
dB at 14.4 GHz; upper rejection: 40 dB at 15.6 GHz;
ripple: 0.05 dB, maximum 3 dB bandwidth variation at the
tuning range edges: +/ - 5%; maximum 20 dB bandwidth
variation across tuning range: +/ - 7%.
These requirements have been realized in the following
design: waveguide housing: R140; number of resonators: 3;
ferrite material ( M, =2.74. lo5 A/m; E , =12.8; slab thick-
ness wf =1.1 mm); narrow wall spacing =0.15 mm; septum
thickness =0.19 mm; I,, =I,, =2.219, I,, =I,, =8.855, I,., =
I,, =10.478, I,, =10.513. The filter tuning range has been
optimized and goes from 13.85 to 16.20 GHz with the inser-
tion loss 0.8-2.2 dB. The tuned center frequency corre-
sponds to a biasing field of 1.9.105A/m.
Recently, significant progress in millimeter-wave magnetic
resonance filters has been reported [ 151, [19]. Instead of
using YIG spheres, the millimeter-wave filters employ highly
anisotropic hexagonal ferrites, thus reducing dramatically the
value of the external magnetic field, which is necessary for
resonance. The filter reported in [15] has been tuned from 50
to 75 GHz with an insertion loss of about 6 dB and an
off-resonance isolation (ORI ) of 35-40 dB. A W-band filter
(75-110 GHz) showed an insertion loss of about 8 dB and an
A further development in YI G filter technology is the
extension of the operating range toward lower frequen-
cies. If the anisotropy effect is neglected, the lowest res-
onant frequency for the spherical YIG resonator (M, =
1.4.1OS A/m) is 1700 MHz, and for Ga-doped YI G it is
about 1000 MHz. By using a disk-shaped resonator, one
OR1 of about 25-30 dB.
647
UHER AND HOEFER: TUNABLE MI CROWAVE AND MI LLI METER-WAVE BAND-PASS FILTERS
MHr at 18 GHz
Fig. 5. Frequency response of six-stage YIG filter (after [23]).
can reduce this frequency theoretically to almost zero. How-
ever, until recently, the lowest resonant frequency achieved
with planar resonators was 500 MHz. This discrepancy be-
tween theory and experiment has been explained [26] by
taking the effective demagnetization factors (eq. (10)) into
account. If a single-crystal planar resonator with rotational
symmetry is configured so that the axis of rotation is ori-
ented, with respect to the crystalline lattice of the ferrimag-
netic material, parallel to a (100) plane for a material with a
negative anisotropy and parallel to a (110) plane inside the
acute angle formed by two [ l l l ] axes for a material with a
positive anisotropy, then the lowest resonant frequency of
such resonator is as low as 50 MHz.
Another problem associated with YIG filters is their mod-
erate tuning speed, which is usually not below some millisec-
onds/GHz. The reason for low tuning speed is the induction
of eddy currents in the magnetic circuit during the tuning
process. More recently, a novel fast tunable type of YI G
filter has been reported [24]. A typical biasing circuit con-
taining an electromagnet has been replaced by two orthogo-
nal Helmholtz coils. The YIG resonator is planar in shape
and the coupling is accomplished by two orthogonal striplines
short-circuited just behind the resonators. The dc magnetic
field can be changed either in amplitude or in direction by
varying the current intensity in the coils. These changes have
no eddy-current delay effect; thus the tuning can be accom-
plished significantly faster.
Further progress in YIG filter technology has been
achieved by introducing six-stage structures. Until recently
the maximum number of spheres used in YIG filters was 4.
The problems associated with alignment of the magnetic
field, which must be uniform for all resonators, were too
severe to obtain six-stage filters with overall good perfor-
mance. This difficulty, however, has recently been overcome
Strip Transducer Mai l Disc
\
\ I
I
GGG Substrate YIG - Film
(a)
1 2 3 4 5
f I [GHz]
v
(b)
Fig. 6. Magnetostatic wave tunable filter. (a) Cross-sectional view of
parallel transducer MSW filter. (b) Frequency response. Geometrical
dimensions: YIG film: 20 x 5 X 0.043 mm; transducer: 5 X 0.2 X 0.05 mm,
spacing: 0.3 mm; disk: I#J =3 mm (after [351).
[23]. By adding two stages, the passband is widened while the
selectivity rises to 36 dB/octave (Fig. 5) . The suppression of
undesirable out-of-band responses is also significantly im-
proved.
B. Magnetostatic Wave Filters
Magnetostatic wave (MSW) devices have been developed
as an extension of SAW components at microwave frequen-
cies [30]-[34]. MSW filters are made of thin YIG films grown
by liquid phase epitaxy on gadolinium gallium garnet sub-
strates. Three modes of magnetostatic wave propagation
(surface wave, forward volume, backward volume) are possi-
ble, depending on the direction of the static magnetic field
which is biasing the YIG films to the resonance. The cou-
pling of the microwave energy to magnetostatic modes is
accomplished by means of microstrip transducers in meander
line or grating configuration. The design of MSW filters is
usually based on the computation of the radiation impedance
for a given transducer geometry. The thickness of the YIG
.. .
648 IEEE TRANSACTIONS ON MI CROWAVE THEORY AND TECHNIQUES, VOL. 39, NO. 4, APRI L 1791
I
I
I
stepped terrl te sl ab
waveguide below cutoff
Fig. 7. Evanescent waveguide magnetically tunable band-pass filter.
film determines the bandwidth of the filter, which is usually
very narrow (0.1%-1%.) The typical insertion loss of such
filters is, however, very high (20-30 dB.1
Recently, a low-insertion-loss MSW band-pass filter (for-
ward volume) has been proposed [35]. The filter configura-
tion (Fig. 6(a)) consists of parallel microstrip transducers
(three to seven) and three MSW disk resonators positioned
on YIG-GGG substrate. By carefully optimizing the radia-
tion resistance, an average insertion loss of 6 dB within a
0.7-5.2 GHz tuning range (Fig. 6(b)) has been achieved.
C. Evanescent Waveguide Filters
The major drawback of ferrimagnetic resonance filters is
their low power handling capability. Even though YIG res-
onators are coupled by sections of a rectangular waveguide,
the power handling capability of such a filter is usually lower
than 200 mW. High-power signals excite spurious modes in
the nonlinear ferrite and degrade the filter response. The
intensity of this undesirable phenomenon is significantly
reduced in off-resonance magnetically tunable filters. Two
types of filters operating in the off-resonance region have
been proposed so far. Historically, the first comprises
evanescent waveguide filters [3], [41], [42]. Fig. 7 shows an
evanescent waveguide filter with stepped ferrite loading [42].
This filter is a modification of a fixed tuned evanescent filter,
and it features a very wide stopband, high stopband attenua-
tion, and compact size. By symmetrically loading the evanes-
cent section with ferrite slabs, the waveguide cutoff fre-
quency is varied. Thus the filter center frequency can be
tuned. For the filter design, a regular lumped-element equiv-
alent network synthesis has been applied. Although the
synthesis procedure described in [421 made an attempt to
compensate for the bandwidth variation in the tuning range,
the experimental results were rather unsatisfactory. The
bandwidth was strongly reduced within a moderate tuning
range. The stopband was narrow, with a relatively low isola-
tion.
D. Printed E-Plane Filters
The drawbacks in the performance of evanescent wave-
guide filters have been overcome to a large extent with the
recently introduced printed E-plane filters [36]-[39]. The
operating principle of these filters is based on the well-known
fact that the resonant frequency of a resonator partially
loaded with ferrite can be tuned by varying the biasing
magnetic field. The following two types of microwave struc-
Fig. 8. Magnetically tunable E-plane printed circuit filters. (a) Sym-
metrically ferrite-loaded metal insert filter. (b) Nonsymmetrically loaded
metal insert filter. (c) Large-gap finline filter on a ferrite substrate. (d)
Large-gap finline filter on a dielectric substrate.
tures are represented in this filter class: metal insert filters
with symmetrical (Fig. 8(a)) or nonsymmetrical (Fig. Nb))
ferrite loading, and large-gap finline filters using printed
ferrite substrate (Fig. 8(c)) or printed dielectric substrate
with symmetrically attached ferrite slabs (Fig. 8(d)). Both
types feature low insertion loss and high power handling
capability, and they can be designed very accurately using
previously described methods. The potential applications
range from communication systems (troposcatters, variable
output multiplexers) to radar and navigation systems. The
most critical aspects of design of these devices are as follows.
There is an absence of ferrite loading in the coupling
sections (ferrite parts in the coupling region cause stop-
band narrowing and large bandwidth variations).
All air gaps between ferrite and waveguide narrow wall
(caused by finite corner radius) must be taken into
account while evaluating the eigenvalue problem. If the
air gaps are neglected, a serious error in the computa-
tion of resonator eigenvectors can occur.
High power handling capability requires some criteria to
be considered for a suitable choice of a corresponding
ferrite material and biasing dc field. Ferrites with rela-
tively constant thermal characteristics and low losses are
recommended to avoid midband frequency shift when a
part of the microwave energy is dissipated in the wave-
guide housing and ferrite. Subsidiary resonance losses
may be reduced if a ferrite material with a suitable spin
wave line width combined with appropriate biasing field
range is selected. For a device with variable bias, opera-
tion below subsidiary resonance is recommended. Some
UHER AND HOEFER: TUNABLE MICROWAVE AND MILLIMETER-WAVE BAND-PASS FILTERS
649
1 1 1 2 1 3
f / GHz
-
40
30
20
10
f/GHz
-
Fig. 9. Ku-band magnetically tunable large-gap finline filter. Corn- Fig. 10. Ku-band magnetically tunable metal insert filter. Computed
puted and measured frequency response. Design data: Ferrite TTVG- and measured frequency response. Design data: Ferrite: lTI-2800,
1200, a =2b =15.799 mm, t =15 wm, I , =I , =1.96 mm, I , =1, =2.136 a =2b =15.799 mm, t =0.19 mm, I =SO mm, w =1 mm, I , = I , =3.59
mm, / , =I , =6.157 mm, I , =I , =7.262 mm, I , =6.204 mm, HI =0, mm, I, =l h =8.916 mm, I, =1, =9.417 mm, 1, =8.921 mm, HI =0,
H, =2.3, 10sA/m, w =0.7 mm. H, =1.72.10sA/m.
examples of maximum values for the magnetic field Hdc
for high-power tunable filters are given in [37].
The millimeter-wave filter design calls for very high
accuracy in the manufacturing of the filter circuit. The
ferrite slabs for frequencies above 40 GHz are too thin
and brittle to be handled safely. Therefore, thin-layer
deposition techniques rather than bulk material must be
considered for millimeter-wave structures.
Since all E-plane filters can be treated analytically with
high accuracy, the resulting designs show very good perfor-
mance and excellent agreement with the theoretically pre-
dicted response. Fig. 9 shows the calculated and measured
responses of a magnetically tunable finline filter for the Ku
band. This filter type was manufactured by etching a finline
circuit from a Cu-sputtered TTVG-1200 (Transtech Inc.)
ferrite substrate. The thickness of the ferrite slab was 1 mm.
The measured insertion losses ranged from 1.3 to 2.3 dB.
Since the electrical lengths of both the copper-clad coupling
sections and the resonator regions depend equally on the
biasing magnetic field, the bandwidth decreases relatively
quickly as the filter is tuned. Fig. 10 shows the computed and
measured responses of a tunable metal insert filter for the
Ku band. The resonators are symmetrically filled with lateral
TTI-2800 (Transtech Inc.) ferrite slabs. The measured inser-
tion loss was about 1.3 dB in the center of the tuning range.
The insertion loss can be further reduced by loading only
one side of the resonator with ferrite. The measured inser-
tion loss was about 0.8 dB for this filter type. The trade-off is
mostly in reduced tuning efficiency and narrower stopband.
The tuning speed of all the filters presented is of the order
of milliseconds and is comparable to the tuning speed of
common YIG filters. Much faster tuning can be achieved if
the homogeneous ferrite slabs are replaced by small ferrite
toroids (Fig. ll(a)). For this filter type only a single wire loop
is necessary for biasing. The typical switching time for the
phase shifters utilizing similar toroids is about 1-3 ps. Thus,
the tuning speed will be improved by two to three orders of
magnitude for such filters. Fig. l l (b) shows the calculated
Ku-band response of this filter type. In this design a ferrite
material with a large built-in magnetic field (2.8. lo5 A/m)
has been assumed. This was necessary to obtain a sufficient
level of premagnetization since the biasing field generated by
a single wire is usually too low. The midband of the filter can
be tuned between 12.3 and 13.1 GHz with an average inser-
tion loss of 1.4 dB. The design theory, as well as experimen-
tal verification of such a filter, was described
in [53].
IV. ELECTRONICALLY TUNABLE BAND-PASS FILTERS
Electronically tunable filters can be tuned very fast (about
1 GHz/ps) over a wide (an octave) tuning range, and they
offer compact size. They have found wide application in
ESM receivers [43]. In most electronically tunable filters,
GaAs varactor diodes are used. The varactor diode capaci-
tance varies with reverse (negative) voltage. When a varactor
is in series with a resonator circuit or element, this capaci-
tance change alters the resonant frequency. A typical varac-
tor-tuned filter utilizes a combline circuit (Fig. 12(a)) con-
structed in suspended stripline technique [44], [46]. The
small-signal equivalent circuit usually assumes resonators
consisting of distributed inductors in parallel with lumped
capacitors, while the coupling is represented by distributed
650 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 39, NO. 4, APRIL 1991
12. 13. 14.
f / GHz
-
(b)
Fig. 11. Fast tunable E-plane passband filter. (a) Positioning of ferrite
toroids in a waveguide housing. (b) Frequency response of Ku-band
design. Design data: a =2b =15.799 mm, a , =19.29 mm, wf =0.6 mm,
wd=0. 4mm, wa=0.15mm, Hdc,=O, HdcZ=2. 75. 105A/ m, Ms=1. 96.
lo5 A/m.
series inductors. The frequency dependence of internal cou-
pling can be compensated by a matching network which also
takes into account the varactor Q effect. The Q factor of a
varactor GaAs diode is usually low ( Q =75-125 at 5 GHz);
therefore the insertion loss of the varactor-tuned filter de-
pends mainly on the Q factor of the diode. The maximum
reported insertion loss of two-resonator filters tuned from 3
to 5 GHz is 5.5 dB [44]. Recently, a similar filter has been
designed using a more rigorous theory, resulting in a reduced
insertion loss: maximum 1.5 dB [46]. Fig. 12(b) shows the
computed and measured filter responses for different bias
values. The variations of bandwidth within the tuning range
have also been reduced and do not exceed 10%.
The main advantage of varactor-tuned filters resides in
their superior tuning speed. However, they also have some
matching
net wor k
f ...
mr t chl ng
net wor k
0
s* 1
d 0
- 1 0
- 2 0
2.8 4. i l GHz 5.2
(b)
filter circuit. (b) Frequency response, after (461.
Fig. 12. Varactor diode tuned combline filter. (a) Topology of the
serious disadvantages. Two-resonator filters are character-
ized by a relatively low selectivity (12 dB/oct) and low
stopband isolation (30 dB). By adding more resonators, an
improved selectivity can be achieved, but only at the expense
of larger insertion loss. Another problem associated with
varactor-tuned filters is their low power handling capability.
Since a varactor diode is a nonlinear device, larger signals
generate harmonics and subharmonics. Large-signal testing
of such filters indicated a third-order intercept point of 7
dBm at zero bias and 37 dBm at 30 V bias [44].
Varactor-tuned filters can also be realized in rectangular
waveguide technique. Examples of such filters are discussed
in [45]. Contrary to common tunable band-pass filters, they
feature variable bandwidth while maintaining a constant
center frequency. This feature has been obtained by coupling
the resonators through variable capacitance. A varactor-
tuned filter with similar passband behavior realized in planar
technique has been described in [49]. Electronically tuned
band-pass filters using ceramic resonators have also been
reported [47], [48].
V. MECHANICALLY TUNABLE FILTERS
Mechanically tunable band-pass filters still draw consider-
able attention among filter designers. Their large power
handling capability and low insertion loss are often decisive
factors if a tunable filter for long-distance communication
(satellite or troposcatter) or radar systems is required. Me-
UHER AND HOEFER: TUNABLE MICROWAVE AND MILLIMETER-WAVE BAND-PASS FILTERS
~
65 1
I I I I I
H M k
(a)
slidln w Ils
A
apertures
(b)
Fig. 13. Mechanically tunable filters. (a) Mechanically tunable band-
pass filter with coaxial resonators. (b) Waveguide tunable band-pass
filter.
chanically tunable filters are usually realized using either
coaxial or waveguide resonators [l]. A systematical design
theory of tunable waveguide filters was presented in [50].
Fig. 13(a) shows a four-resonator coaxial band-pass filter.
The resonators operate in the TEM mode and are AJ4
long at resonance. The input and output ports are coupled to
the resonators by means of a magnetic loop. All resonators
are approximately coaxial in the cross section with a some-
what flattened coupling region to keep the irises as thin as
possible. The coaxial filters are manufactured with between
two and six resonators and cover the frequency range from
400 MHz to 12 GHz with a typical tuning range between
20% and 60% of a standard waveguide band.
Fig. 13(b) shows a waveguide counterpart of a coaxial
filter. The filter is tuned by a sliding wall on one side of each
resonator cavity. The resonators are coupled by apertures
which are shifted from the positions of the aperture coupling
at the input and output. This reduces the variations of
bandwidth and response shape as the filter is tuned. The
cavities of mechanically tunable filters (usually between two
and six) can be either rectangular or cylindrical. In [52] a
TE,,,,-mode filter with four cylindrical cavities was reported.
The tuning can be accomplished by a movable plunger of a
diameter smaller than or equal to that of the cavity. The
TE,, ,-mode field distribution allows for a nontouching pis-
ton because the wall currents are purely circumferential.
Therefore the Q factor of the cavity remains high
dB
5 0
4 0
30
2 0
1 0
11.8 12. 12. 2
f / GHz
L
Fig. 14. Mechanically tunable TE, , 3 dual-mode band-pass filter-
frequency response (after (511).
(Q =9000-11000). Such filters possess a very high power
handling capability (500 W CW at 12 GHz).
The main drawbacks of the TEol l filters are spurious
resonances, which are relatively close to the main resonance
and must be suppressed to an acceptable level, and the large
size and mass of such filters. This size can be reduced by
using dual-mode techniques. In [51] a TE 113 dual-mode tun-
able filter was presented. The two cavities are arranged side
by side in split block technique. The filter is tuned by only
two plungers which are located opposite to the coupling side.
Since for this particular filter only a moderate tuning range
is required (+ / - 50 MHz at 12 GHz), only a small part of
the end plate is used as a plunger. Fig. 14 shows the
measured frequency response of the filter [51]. The return
loss of this filter is better than 25 dB, and the equiripple
bandwidth varies by about 10%.
The biggest disadvantage of the mechanically tunable fil-
ters is their low tuning speed. The filters can be tuned
manually or via telecommand if the filter is combined with a
remotely controlled motor.
VI. CONCLUSIONS
I n Table I, the performance of all the filters presented is
summarized. From this comparison it is obvious that none of
these devices can simultaneously satisfy all requirements for
perfect tunable filters. For microwave systems where multi-
octave tuning is essential, the YIG filter is an obvious choice.
In systems where the requirement of high power handling
capability combined with low insertion loss predominates,
mechanically tunable filters and magnetically tunable E-
plane filters are recommended. I f the tuning speed is a
crucial requirement, varactor-tuned filters or E-plane filters
with ferrite toroids are devices of choice. For millimeter-wave
design, the most promising structures are ferrimagnetic reso-
nance filters utilizing hexagonal ferrite resonators or, up to
65 2 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 39, NO. 4, APRIL 1991
TABLE I
TYPICAL PERFORMANCE PARAMETERS OF MICROWAVE TUNABLE BAND-PASS FILTERS
Performance
Parameter
Bandwidth
[%I
Insertion loss
IdBl
Selectivity
[dB/oct]
Rejection att.
[dBl
Power handling
capability [W]
Tuning range
[% of WG-band]
Tuning speed
[GHz/msl
BW variation
[%I
Tuning linearity
[MHzl
Temp. Stability
Millimeter-wave
capability
[MHzl
Mechanically Tunable YIG MSW E-Plane
Filter Filter Filter Filter
0.3-3 0.2-3 0.2-0.5 1-10
0.5-2.5 3-8 6-10 0.7-2.5
12-24 12-36 24 12-24
>50 40-60 >45 >SO
100-500 0.1-1 0.05 50-200
5-20 multi- multi- 60-70
octave octave
very low 0.5-2 0.5-2 0.5-103
10-20 10-40 10-40 5-10
+/ - E +/ - l o +/ - l o +/ - E
20 15 15 25
no yes no yes
Varactor
Tuned Filter
2-20
0.3-2.5
12-24
>30
0.05-0.1
- octave
103
10-20
+/-35
25
no
60 GHz, magnetically tunable E-plane printed circuit filters.
Further developments in the field of microwave tunable
filters will depend strongly on the progress in the relevant
microwave technologies. The most important fields for re-
search seem to be in MMIC integration, high-Q varactors,
and low-cost manufacturing methods for single-crystal ferrite
layers. Superconductive cavities will also have a strong im-
pact on the future development of high-performance tunable
filters.
REFERENCES
[l ] G. L. Matthaei, L. Young, and E. M. T. J ones, Microwace
Filters, Impedance Matching Networks, and Coupling Structures.
New York: McGraw-Hill, 1964, ch. 17.
[2] J . Helszajn, YIG Resonators and Filters. New York: Wiley,
1985.
[3] G. Craven, Evanescent Mode Waueguide Components. Boston:
Artech House, 1987.
[4] Y. Konishi and K. Uenakada, The design of a bandpass filter
with inductive strip-planar circuit mounted in waveguide,
IEEE Trans. Microwace Theory Tech., vol. MTT-22, pp.
151 R. Vahldieck, J . Bornemann, F. Arndt, and D. Grauerholtz,
Optimized waveguide E-plane metal insert filters for millime-
ter-wave applications, IEEE Trans. Microwace Theory Tech .,
vol. MTT-31, pp. 65-69, J an. 1983.
[61 L. Q. Bui, D. Ball, and T. Itoh, Broad-band millimeter-wave
E-plane bandpass filters, IEEE Trans. Microwace Theory
Tech., vol. Ml T-32, pp. 1655-1659, Dec. 1984.
[7] S. Cohn, Direct-coupled-resonator filters, Proc. IRE, vol. 45,
pp. 187-196, Feb. 1957.
[8] L. Young, Direct-coupled cavity filters for wide and narrow
bandwidths, IEEE Trans. Microware Theory Tech., vol. MTT-
[91 R. Levy, Theory of direct-coupled cavity filters, IEEE Truns.
Microwave Theory Tech., vol. MTT-15, pp. 340-348, 1967.
1101J . D. Rhodes, Theory of Electrical Filters. New York: Wiley,
ch. 4.
[l l ] I. Wolff, Felder und Wellen in Gyrofropen Mikrowellenstruk-
turen. Braunschweig: Verlag Vieweg and Sohn, 1973.
[12] R. W. deGrasse, Low-loss gyromagnetic coupling through
single crystal garnets, J. Appl. Phys., vol. 30, pp. 1555-1559.
[131 C. Kittel, On the theory of ferromagnetic resonance absorp-
tion, Phys. Rec., vol. 77, pp. 155-161, 1Y48.
1209-1216, 1974.
11, pp. 162-178, 1963.
[14] R. Soohoo, Microwace Magnetics. New York: Harper and
Row, 1985.
[15] H. Tanbakuchi, D. Nicholson, B. Kunz and W. Ishak, Magnet-
ically tunable oscillators and filters, IEEE Trans. Magn., vol.
25, pp. 3248-3253, Sept. 1989.
[16] R. F. Fjerstad, Some design considerations and realizations of
iris-coupled YIG-tuned filters in the 12-40 GHz region, IEEE
Trans. Microwaue Theory Tech., vol. MTT-18, pp. 205-212,
Apr. 1970.
[17] G. H. Thiess, Theory and design of tunable YIG-filters,
Microwaces, Sept. 1964.
[ 181 K. D. Gilbert, Dynamic tuning characteristics of YIG-devices,
Micrewace J. , no. 6, pp. 36-40, J une 1970.
[19] D. Nicholson, Ferrite tuned millimeter wave bandpass filters
with high off-resonance isolation, in IEEE 2988 MTT-S Int.
Microwar>e Symp. Dig., May 1988, pp. 867-870.
[20] P. S. Carter, Equivalent circuit of orthogonal-loop-coupled
magnetic resonance filters and bandwidth narrowing due to
coupling resonance, IEEE Trans. Microwace Theory Tech.,
vol. MTT-18, pp. 100-105, Feb. 1970.
[21] P. S. Carter, Side-wall coupled, strip-transmission line tunable
filters employing ferrimagnetic Y I G resonators, IEEE Trans.
Microwace Theory Tech., vol. MTT-13, pp. 306-315, May 1965.
[22] L. Rhymes, Two-sphere Y I G multiplier/filter ensures purity,
Microwaies/HF, pp. 109-116, Apr. 1988.
[23] M. Korber, Six-stage filters, Watkins-Johnson Company Tech-
nical Notes, vol. 15, no. 5, Sept./Oct. 1988.
[24] C. Vittoria, High speed frequency tunable microwave filter,
U.S. Patent 4 197 517, Apr. 8, 1980.
[25] C. Vittoria, Tunable microwave filters utilizing a slotted line
circuit, U.S. Patent 4 590 448, May 20, 1986.
[2h] G. U. Sorger and D. Raicu, Magnetically tunable resonators
and tunable devices such as filters and resonant circuits for
oscillators using magnetically tuned resonators, U.S. Patent 4
555 683, Nov. 26, 1985.
[27] L. Young and D. B. Weller, A 500-1000 MHz magnetically
tunable bandpass filter using two YIG-disc resonators, IEEE
Trans. Microwuic Theory Tech., vol. MTT-15, pp. 72-86, Feb.
1967.
[28] Improving Microwave Measurement with Ferre-trac- Electr.
Tuned Filters 0.5-26 GHz, Technical Note Ferretec-FT3,
Ferretec Inc., Fremont, CA.
[29] YIG-Filters, Technical Note,- Siversima, Culver City, CA.
[30] Y. Murakami, T. Ohgihara, and T. Okamoto, A 0.5-4.0 GHz
tunable bandpass filter using YIG-film grown by LPE, IEEE
Trans. Microwai:e Theory Tech., vol. MTT-35, pp. 1192-1197,
Dec. 1987.
UHER AND HOEFER: TUNABLE MICROWAVE AND MILLIMETER-WAVE BAND-PASS FILTERS
653
[311 J . P. Castera and P. Hartemann, A multipole magnetostatic
volume wave resonator filter, IEEE Trans. Magn., vol. MAG-
18, pp. 1601-1603, 1982.
[321 E. Huijer and W. Ishak, MSSW resonators with straight edge
reflectors, IEEE Trans. Magn., vol. MAG-20, pp. 1232-1234,
1984.
[331 J .D. Adam, A MSW tunable bandpass filter, in Proc. IEEE
Ultrasonics Symp., 1985.
[341 W. S. Ishak and K. W. Chang, Tunable microwave resonators
using magnetostatic wave in YIG-films, IEEE Trans. Mi-
crowae Theory Tech., vol. MTT-34. m. 1383-1393. Dec. 1986.
pean Microwace Conf. (London), Sept. 1989.
[53] J . Uher and W. J . R. Hoefer, Fast tunable bandpass filter with
high power handling capability, in MIOP 1990 ConJ Dig.
[54] G. C. Tai, C. H. Chen, and Y.-W. Kiang, Plasma-dielectric
sandwich structure used as a tunable bandpass microwave
filter, IEEE Trans. Microwace Theory Tech., vol. MTT-32, pp.
111-113, J an. 1984.
T. Nishikawa et al., A low-loss magnetostatic wave filter using
parallelstrip transducer, in 1989 IEEE MTT-S Int. Microwace
Symp. Dig., J une 1989, pp. 153-158.
J . Uher, J . Bornemann, and F. Arndt, Magnetically tunable
rectangular waveguide E-plane integrated circuit filters, IEEE
Trans. Microwace Theory Tech., vol. 36, pp. 1014-1022, J une
1988.
J . Uher, J . Bornemann, and F. Arndt, Computer-aided design
and improved performance of tunable ferrite-loaded E-plane
integrated circuit filters, IEEE Trans. Microwace Theory Tech.,
vol. 36, pp. 1841-1858, Dec. 1988.
J . Uher, J . Bornemann, and F. Arndt, Ferrite tunable millime-
ter wave printed circuit filters, in IEEE I988 MTT-S Int.
Microwace Symp. Dig., May 1988, pp. 871-874.
J . Uher and W. J . R. Hoefer, Analysis and tuning efficiency
optimization of magnetically tuned printed E-plane circuit
filters, in IEEE 1989 MTT-S Int. Microwace Symp. Dig., J une
1989, pp. 1273-1276.
H. Schmiedel, Anwendung der Evolutionsoptimierung bei
Mikrowellenschaltunden, Frequenz, vol. 35, pp. 306-310, Nov.
1981.
R. F. Skedd and G. Craven, Magnetically tunable multisection
bandpass filters in ferrite-loaded evanescent waveguide, Elec-
tron. Lett., vol. 3, pp. 62-63, Feb. 1967.
R. Snyder, Stepped-ferrite tunable evanescent filters, IEEE
Trans. Microwave Theory Tech., vol. MTT-29, pp. 364-371,
Apr. 1981.
C. B. Hofmann and A. R. Baron, Wideband ESM receiving
systems, Microwace J ., no. 9, pp. 24-34, Sept. 1980.
I. C. Hunter and J . D. Rhodes, Electronically tunable mi-
crowave bandpass filters, IEEE Trans. Microwace Theory
Tech., vol. MTT-30, pp. 1354-1360, Apr. 1982.
S. Toyoda, Variable bandpass filters using varactor diodes,
IEEE Trans. Microwace Theory Tech., vol. MTT-29, pp.
356-362, Apr. 1981.
S. Kumar and Y. Liang, Varactor tuned suspended substrate
combline filter, in Proc. Can. Conf. Elect. and Comp. Eng.,
Sept. 1989.
M. A. Harris, High-power tunable filter, U.S. Patent 4 692
724, Sept. 8, 1987.
J . B. West, Ceramic TEM resonator bandpass filters with
varactor tuning, U.S. Patent 4 721 932, J an. 26, 1988.
A. Schwarzmann, Adjustable passband filter, U.S. Patent
4 250 457, Feb. 10, 1981.
B. Rawat, Tunable waveguide filters-A practical design pro-
cedure, Microwares and RF, no. 9, pp. 97-102, Sept. 1983.
U. Rosenberg, D. Rosowski, W. Rummer, and D. Wolk,
Tunable manifold multiplexers-A new possibility for satellite
redundancy philosophy, in Proc. MIOP Int. Conf. (W. Ger-
many), Feb. 1989.
M. A. Kunes and G. G. Connor, A digitally controlled tun-
able, high output filter for space applications, in Proc. Euro-
Jaroslaw Uher (M88) received the M.Sc. de-
gree in electronic engineering from the Tech-
nical University of Wroclaw, Poland, in 1978
and the Dr.-Ing. degree in microwave engi-
neering from the University of Bremen, Bre-
men, West Germany, in 1987.
From 1978 to 1982 his work dealt with
planar ferrite technology at the Institute of
Electronic Technology, Technical University
of Wroclaw. From 1983 to 1988 he was with
the Microwave Department of the University
of Bremen, where his research activities involved field problems of
ferrimagnetic slab discontinuities in waveguides structures and the
design of ferrite control components and tunable filters. From
September 1988 to August 1990 he was a Research Engineer at the
University of Ottawa, Ottawa, Canada, where he was involved in
research on quasi-planar components and numerical techniques. I n
September 1990 he joined SPAR Aerospace Ltd., where he is
presently working as a Senior Member of the Technical Staff,
developing numerical tools for microwave components design for
satellite hardware.
Wolfgang J. R. Hoefer (M71-SM78-F91)
received the diploma in electrical engineering
from the Technische Hochschule Aachen,
Aachen, Germany, in 1964 and the D.Ing.
degree from the University of Grenoble,
France, in 1968.
After one year of teaching and research at
the Institut Universitaire de Technologic,
Grenoble, France, he joined the Department
of Electrical Engineering, University of Ot-
tawa, Ottawa, Ont., Canada, where he is
currently a Professor. His sabbatical activities have included SIX
months with the Space Division of AEG-Telefunken in Backnang,
Germany, six months with the Electromagnetics Laboratory of the
Institut National Polytechnique de Grenoble, France, and one year
with the Space Electronics Directorate of the Communication Re-
search Centre in Ottawa, Canada. His research interests include
microwave measurement techniques, millimeter-wave circuit design,
and numerical techniques for solving electromagnetic problems.
Dr. Hoefer is a registered Professional Engineer in the province
of Ontario, Canada.