Tessent Scan and ATPG User’s Manual

Software Version 2014.3

September 2014

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 Table of Contents

Chapter 1
Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
What is Design-for-Test?. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
DFT Strategies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
Top-Down Design Flow with DFT . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18

Chapter 2
Understanding Scan and ATPG Basics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
Understanding Scan Design. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
Internal Scan Circuitry . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
Scan Design Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
Understanding Scan . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
Understanding Wrapper Chains . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
Understanding Test Points . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
Test Structure Insertion with Tessent Scan. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
Understanding ATPG. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
The ATPG Process . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
Mentor Graphics ATPG Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
Scan Sequential ATPG with the ATPG Tool . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
Understanding Test Types and Fault Models . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32
Test Types. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33
Fault Modeling . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
User-Defined Fault Modeling . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 42
Multiple Detect . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 51
Embedded Multiple Detect . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 52
Fault Detection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 54
Fault Classes . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 55
Testability Calculations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 65

Chapter 3
Understanding Common Tool Terminology and Concepts . . . . . . . . . . . . . . . . . . . . . . . . 67
Scan Terminology . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 67
Scan Cells . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 67
Master Element . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 69
Slave Element . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 69
Shadow Element . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 69
Copy Element . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 70
Extra Element . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 71
Scan Chains. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 71
Scan Groups . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 72
Scan Clocks. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 73
Scan Architectures . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 73

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Mux-DFF . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 74
Test Procedure Files. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 74
Model Flattening . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 75
Understanding Design Object Naming . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 75
The Flattening Process . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 75
Simulation Primitives of the Flattened Model . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 77
Learning Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 80
Equivalence Relationships . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 80
Logic Behavior . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 81
Implied Relationships . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 81
Forbidden Relationships . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 82
Dominance Relationships . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 82
ATPG Design Rules Checking . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 83
General Rules Checking . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 83
Procedure Rules Checking . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 83
Bus Mutual Exclusivity Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 84
Scan Chain Tracing. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 85
Shadow Latch Identification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 85
Data Rules Checking. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 86
Transparent Latch Identification. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 86
Clock Rules Checking. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 86
RAM Rules Checking . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 86
Bus Keeper Analysis. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 87
Extra Rules Checking . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 87
Scanability Rules Checking . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 88
Constrained/Forbidden/Block Value Calculations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 88

Chapter 4
Understanding Testability Issues . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 91
Synchronous Circuitry . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 92
Synchronous Design Techniques . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 92
Asynchronous Circuitry . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 93
Scannability Checking . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 93
Scannability Checking of Latches . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 94
Support for Special Testability Cases . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 94
Feedback Loops . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 94
Structural Combinational Loops and Loop-Cutting Methods . . . . . . . . . . . . . . . . . . . . . . 94
Structural Sequential Loops and Handling . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 100
Redundant Logic. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 101
Asynchronous Sets and Resets . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 101
Gated Clocks. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 102
Tri-State Devices . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 103
Non-Scan Cell Handling . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 104
Clock Dividers . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 108
Pulse Generators . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 109
JTAG-Based Circuits . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 110
Testing RAM and ROM . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 111
Incomplete Designs. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 117

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Chapter 5
Inserting Internal Scan
and Test Circuitry. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 119
The Tessent Scan Process Flow. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 119
Tessent Scan Inputs and Outputs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 121
Test Structures Supported by Tessent Scan . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 122
Invoking Tessent Scan . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 123
Preparing for Test Structure Insertion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 123
Defining Scan Cell and Scan Output Mapping. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 123
Enabling Test Logic Insertion. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 124
Specifying Clock Signals . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 127
Specifying Existing Scan Information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 128
Handling Existing Boundary Scan Circuitry . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 130
Changing the System Mode (Running Rules Checking) . . . . . . . . . . . . . . . . . . . . . . . . . . 131
Setting Up a Basic Scan Insertion Run . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 131
Setting Up for Wrapper Chain Identification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 131
Manually Specifying Control and Observe Points . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 134
Manually Including and Excluding Cells for Scan . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 134
Reporting Scannability Information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 136
Automatic Recognition of Existing Shift Registers . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 138
The Identification Process. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 141
Reporting Identification Information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 141
Inserting Test Structures . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 141
Setting Up for Internal Scan Insertion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 141
Naming Scan Input and Output Ports . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 142
Running the Insertion Process . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 144
Merging Scan Chains with Different Shift Clocks . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 145
Saving the New Design and ATPG Setup . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 147
Writing the Netlist. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 147
Writing the Test Procedure File and Dofile for ATPG. . . . . . . . . . . . . . . . . . . . . . . . . . . . 148
Running Rules Checking on the New Design. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 148
Exiting Tessent Scan. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 148
Inserting Scan Block-by-Block . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 148
Verilog Flow Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 149

Chapter 6
Core Mapping for Top-Level ATPG. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 153
Core Mapping for ATPG Process Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 154
Core Mapping Examples . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 162
Limitations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 166

Chapter 7
Generating Test Patterns . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 167
ATPG Basic Tool Flow . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 168
ATPG Tool Inputs and Outputs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 170
Understanding the ATPG Method . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 171
Performing Basic Operations. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 176
Invoking the Application . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 177

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Setting the System Mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 177

Setting Up Design and Tool Behavior . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 177
Setting Up the Circuit Behavior . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 177
Setting Up Tool Behavior . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 182
Defining the Scan Data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 187
Checking Rules and Debugging Rules Violations. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 189
Running Good/Fault Simulation on Existing Patterns. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 189
Fault Simulation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 190
Good-Machine Simulation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 192
Running Random Pattern Simulation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 193
Changing to the Fault System Mode. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 193
Adding the Faults List. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 193
Running the Simulation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 193
Setting Up the Fault Information for ATPG . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 193
Changing to the Analysis System Mode. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 194
Setting the Fault Type. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 194
Creating the Faults List. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 194
Adding Faults to an Existing List . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 195
Loading Faults from an External List . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 195
Writing Faults to an External File. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 195
Setting the Fault Sampling Percentage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 196
Setting the Fault Mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 196
Setting the Possible-Detect Credit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 196
Performing ATPG . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 197
Setting Up for ATPG . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 197
Creating Patterns with Default Settings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 205
Approaches for Improving ATPG Efficiency. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 205
Saving the Test Patterns . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 208
Low-Power ATPG . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 208
Low-Power Capture . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 208
Low-Power Shift. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 209
Setting up Low-Power ATPG. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 209
Creating an IDDQ Test Set . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 211
Generating an IDDQ Test Set . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 212
Specifying Leakage Current Checks. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 213
Net Pair Identification with Calibre for Bridge Fault Test Patterns . . . . . . . . . . . . . . . . . . . 214
Four-Way Dominant Fault Model . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 214
Top-level Bridging ATPG . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 214
Incremental Multi-Fault ATPG. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 215
The Bridge Parameters File . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 217
Creating a Delay Test Set . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 219
Creating a Transition Delay Test Set . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 220
Transition Fault Detection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 221
Basic Procedure for Generating a Transition Test Set . . . . . . . . . . . . . . . . . . . . . . . . . . . . 225
Timing for Transition Delay Tests . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 225
Preventing Pattern Failures Due to Timing Exception Paths . . . . . . . . . . . . . . . . . . . . . . . 228
Creating a Path Delay Test Set . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 234
At-Speed Test Using Named Capture Procedures . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 245
Mux-DFF Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 253

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Support for Internal Clock Control . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 259

Generating Test Patterns for Different Fault Models and Fault Grading . . . . . . . . . . . . . . 263
Using Timing-Aware ATPG. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 265
Generating Patterns for a Boundary Scan Circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 276
Dofile and Explanation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 276
TAP Controller State Machine . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 277
Test Procedure File and Explanation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 278
Using the MacroTest Capability . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 282
The MacroTest Process Flow . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 284
Qualifying Macros for MacroTest . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 286
When to Use MacroTest . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 287
Defining the Macro Boundary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 289
Defining Test Values . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 293
Recommendations for Using MacroTest . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 295
MacroTest Examples . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 297
Verifying Test Patterns . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 302
Simulating the Design with Timing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 302
Potential Causes of Simulation Mismatches. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 305
When, Where, and How Many Mismatches? . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 306
DRC Issues . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 307
Shadow Cells . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 307
Library Problems . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 307
Timing Violations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 307
Analyzing the Simulation Data. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 308
Analyzing Simulation Mismatches . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 310

Chapter 8
Multiprocessing for ATPG and Simulation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 321
Definition of Multiprocessing Terms . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 321
Using Multiprocessing to Reduce Runtime . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 322
Multiprocessing Requirements . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 322
Procedures for Multiprocessing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 324
Using Multiprocessing Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 324
Troubleshooting SSH Environment and Passphrase Errors . . . . . . . . . . . . . . . . . . . . . . . . 325
Disabling Multithreading Functionality . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 327
Adding Threads to the Master. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 327
Adding Processors in Manual Mode. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 328
Adding Processors in Grid Mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 330
Adding Processors to the LSF Grid . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 331
Deleting Processors Added in Manual Mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 332
Deleting Processors Added in Grid Mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 332

Chapter 9
Scan Pattern Retargeting . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 333
Scan Pattern Retargeting Process Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 334
Core-Level Pattern Generation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 338
Scan Pattern Retargeting of Test Patterns . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 340
Test Procedure Retargeting For Scan Pattern Retargeting . . . . . . . . . . . . . . . . . . . . . . . . . 341

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Scan Pattern Retargeting Without a Netlist . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 343

Generating Patterns at the Core Level . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 348
Retargeting Patterns in Internal Mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 349
Extracting Connectivity Between the Top Level and the Core Boundary. . . . . . . . . . . . . . . 351
Retargeting Patterns Without a Top-Level Netlist. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 352
Generating Patterns in External Mode. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 353
Retargeting Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 354
Core-in-Core Pattern Retargeting . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 358
Overview. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 359
Generation of Child Core Internal Mode Patterns . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 361
Retargeting of Child Core Internal Mode Patterns . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 361
Generation of Parent Internal Mode Patterns . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 362
Retargeting of Parent Internal Mode Patterns. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 363
Limitations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 364

Chapter 10
Test Pattern Formatting and Timing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 365
Test Pattern Timing Overview. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 366
Timing Terminology . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 366
General Timing Issues . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 366
Generating a Procedure File. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 367
Defining and Modifying Timeplates . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 368
Saving Timing Patterns . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 369
Features of the Formatter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 369
Pattern Formatting Issues . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 370
Saving Patterns in Basic Test Data Formats . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 372
Saving in ASIC Vendor Data Formats . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 377

Chapter 11
Test Pattern File Formats . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 379
ASCII File Format . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 379
Header_Data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 379
Setup_Data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 379
Functional_Chain_Test . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 382
Scan_Test . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 383
Scan_Cell . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 385
BIST Pattern File Format. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 387
Setup_Data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 387
Scan_Test . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 388

Chapter 12
Power-Aware DRC and ATPG . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 391
Power-Aware Overview. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 391
Assumptions and Limitations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 392
Multiple Power Mode Test Flow . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 392
Power-Aware ATPG for Traditional Fault Models . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 393
CPF and UPF Parser . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 394
Power-Aware ATPG Procedure. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 396

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Power-Aware Flow Examples . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 397

Chapter 13
Testing Low-Power Designs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 401
Low-Power Testing Overview. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 402
Assumptions and Limitations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 402
Low-Power CPF/UPF Parameters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 402
Test Insertion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 404
Low-Power Test Flow. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 404
Scan Insertion with Tessent Scan . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 405
Power-Aware Design Rule Checks . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 409
Low-Power DRCs. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 409
Low-Power DRC Troubleshooting . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 409
Power State-Aware ATPG. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 411
Power Domain Testing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 411
Low-Power Cell Testing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 413

Chapter 14
Using MTFI Files . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 415
MTFI File Format . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 416
MTFI Features . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 418
Support of Fault Classes and Sub-Classes . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 419
Support of Stuck and Transition Fault Information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 421
Support of N-Detect Values . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 421
Support of Different Fault Types in the Same File. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 423
Support for Hierarchical Fault Accounting. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 423
Commands that Support MTFI . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 425

Chapter 15
Graybox Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 427
What is a Graybox? . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 427
Graybox Process Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 429
Example dofile for Creating a Graybox Netlist . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 431
Generating Graybox Netlist for EDT Logic Inserted Blocks . . . . . . . . . . . . . . . . . . . . . . . 431

Appendix A
Clock Gaters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 435
Basic Clock Gater Cell. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 435
Two Types of Embedding . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 435
Ideal Case (Type-A) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 436
Potential DRC Violator (Type-B). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 437
Cascaded Clock Gaters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 439
Understanding a Level-2 Clock Gater . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 439
Example Combinations of Cascaded Clock Gaters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 440
Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 441

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Appendix B
Debugging State Stability . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 443
Displaying the State Stability Data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 443
State Stability Data Format. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 444
State Stability Examples . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 445

Appendix C
Running Tessent Shell as a Batch Job . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 467
Commands and Variables for the dofile . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 467
Command Line Options. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 468
Scheduling a Batch Job for Execution Later. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 469

Appendix D
Getting Help . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 471
Documentation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 471
Mentor Graphics Support. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 472
Index
Third-Party Information
End-User License Agreement

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 List of Figures

Figure 1-1. Top-Down Design Flow Tasks and Products . . . . . . . . . . . . . . . . . . . . . . . . . . . 20

Figure 2-1. Test Concepts . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
Figure 2-2. Design Before and After Adding Scan . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
Figure 2-3. Scan Representation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
Figure 2-4. Example of Partitioned Design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
Figure 2-5. Wrapper Chains Added to Partition A . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
Figure 2-6. Uncontrollable and Unobservable Circuitry . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28
Figure 2-7. Testability Benefits from Test Point Circuitry . . . . . . . . . . . . . . . . . . . . . . . . . . 28
Figure 2-8. Manufacturing Defect Space for a Design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32
Figure 2-9. Internal Faulting Example. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
Figure 2-10. Single Stuck-At Faults for AND Gate . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37
Figure 2-11. IDDQ Fault Testing. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39
Figure 2-12. Transition Fault Detection Process . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40
Figure 2-13. Fault Detection Process. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 54
Figure 2-14. Path Sensitization Example. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 55
Figure 2-15. Example of “Unused” Fault in Circuitry. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 56
Figure 2-16. Example of “Tied” Fault in Circuitry . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 57
Figure 2-17. Example of “Blocked” Fault in Circuitry . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 58
Figure 2-18. Example of “Redundant” Fault in Circuitry . . . . . . . . . . . . . . . . . . . . . . . . . . . 58
Figure 2-19. Fault Class Hierarchy . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 61
Figure 3-1. Common Tool Concepts . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 67
Figure 3-2. Generic Scan Cell . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 68
Figure 3-3. Generic Mux-DFF Scan Cell Implementation . . . . . . . . . . . . . . . . . . . . . . . . . . 68
Figure 3-4. LSSD Master/Slave Element Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 69
Figure 3-5. Dependently-clocked Mux-DFF/Shadow Element Example . . . . . . . . . . . . . . . 70
Figure 3-6. Independently-clocked Mux-DFF/Shadow Element Example . . . . . . . . . . . . . . 70
Figure 3-7. Mux-DFF/Copy Element Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 71
Figure 3-8. Generic Scan Chain. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 72
Figure 3-9. Generic Scan Group . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 72
Figure 3-10. Scan Clocks Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 73
Figure 3-11. Mux-DFF Replacement. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 74
Figure 3-12. Design Before Flattening . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 76
Figure 3-13. Design After Flattening. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 76
Figure 3-14. 2x1 MUX Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 78
Figure 3-15. LA, DFF Example. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 78
Figure 3-16. TSD, TSH Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 79
Figure 3-17. PBUS, SWBUS Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 79
Figure 3-18. Equivalence Relationship Example. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 80
Figure 3-19. Example of Learned Logic Behavior . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 81
Figure 3-20. Example of Implied Relationship Learning . . . . . . . . . . . . . . . . . . . . . . . . . . . 81

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 List of Figures

Figure 3-21. Forbidden Relationship Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 82

Figure 3-22. Dominance Relationship Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 82
Figure 3-23. Bus Contention Example. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 84
Figure 3-24. Bus Contention Analysis. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 84
Figure 3-25. Simulation Model with Bus Keeper . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 87
Figure 3-26. Constrained Values in Circuitry . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 88
Figure 3-27. Forbidden Values in Circuitry. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 89
Figure 3-28. Blocked Values in Circuitry . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 89
Figure 4-1. Testability Issues. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 91
Figure 4-2. Structural Combinational Loop Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 95
Figure 4-3. Loop Naturally-Blocked by Constant Value. . . . . . . . . . . . . . . . . . . . . . . . . . . . 95
Figure 4-4. Cutting Constant Value Loops . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 96
Figure 4-5. Cutting Single Multiple-Fanout Loops . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 96
Figure 4-6. Loop Candidate for Duplication . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 97
Figure 4-7. TIE-X Insertion Simulation Pessimism . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 97
Figure 4-8. Cutting Loops by Gate Duplication . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 98
Figure 4-9. Cutting Coupling Loops . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 99
Figure 4-10. Sequential Feedback Loop . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 100
Figure 4-11. Test Logic Added to Control Asynchronous Reset. . . . . . . . . . . . . . . . . . . . . . 102
Figure 4-12. Test Logic Added to Control Gated Clock . . . . . . . . . . . . . . . . . . . . . . . . . . . . 103
Figure 4-13. Tri-state Bus Contention . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 104
Figure 4-14. Requirement for Combinationally Transparent Latches . . . . . . . . . . . . . . . . . . 105
Figure 4-15. Example of Sequential Transparency . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 106
Figure 4-16. Clocked Sequential Scan Pattern Events. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 107
Figure 4-17. Clock Divider . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 109
Figure 4-18. Example Pulse Generator Circuitry . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 109
Figure 4-19. Long Path Input Gate Must Go to Gates of the Same Type . . . . . . . . . . . . . . . 110
Figure 4-20. Design with Embedded RAM . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 111
Figure 4-21. RAM Sequential Example. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 113
Figure 5-1. Internal Scan Insertion Procedure . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 119
Figure 5-2. Basic Scan Insertion Flow with Tessent Scan. . . . . . . . . . . . . . . . . . . . . . . . . . . 120
Figure 5-3. The Inputs and Outputs of Tessent Scan. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 121
Figure 5-4. Test Logic Insertion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 125
Figure 5-5. Example Report from report_dft_check Command . . . . . . . . . . . . . . . . . . . . . . 137
Figure 5-6. Lockup Cell Insertion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 146
Figure 5-7. Hierarchical Design Prior to Scan. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 149
Figure 5-8. Final Scan-Inserted Design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 151
Figure 6-1. Generation of Core Description Files . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 157
Figure 6-2. Core Mapping to Top Level . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 158
Figure 6-3. Generation of Top-Level TCD File for Top-Level Logic. . . . . . . . . . . . . . . . . . 160
Figure 6-4. Alternate Process for Core Mapping to Top Level . . . . . . . . . . . . . . . . . . . . . . 161
Figure 6-5. Mapping of Cores to Chip Level. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 162
Figure 7-1. Test Generation Procedure . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 167
Figure 7-2. Overview of ATPG Tool Usage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 168
Figure 7-3. ATPG Tool Inputs and Outputs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 170

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List of Figures

Figure 7-4. Clock-PO Circuitry . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 173

Figure 7-5. Data Capture Handling Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 185
Figure 7-6. Efficient ATPG Flow . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 197
Figure 7-7. Circuitry with Natural “Select” Functionality. . . . . . . . . . . . . . . . . . . . . . . . . . . 199
Figure 7-8. Simulation Frames . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 201
Figure 7-9. Waveform Modeling for DFFs and Latches . . . . . . . . . . . . . . . . . . . . . . . . . . . . 202
Figure 7-10. Flow for Creating a Delay Test Set. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 220
Figure 7-11. Transition Delay . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 221
Figure 7-12. Transition Launch and Capture Events . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 222
Figure 7-13. Events in a Broadside Pattern . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 222
Figure 7-14. Basic Broadside Timing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 223
Figure 7-15. Events in a Launch Off Shift Pattern . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 223
Figure 7-16. Basic Launch Off Shift Timing. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 223
Figure 7-17. Broadside Timing Example. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 226
Figure 7-18. Launch Off Shift (Skewed) Timing Example . . . . . . . . . . . . . . . . . . . . . . . . . . 227
Figure 7-19. Multicycle Path Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 228
Figure 7-20. Setup Time and Hold Time Violations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 229
Figure 7-21. Across Clock Domain Hold Time Violation. . . . . . . . . . . . . . . . . . . . . . . . . . . 230
Figure 7-22. Effect Cone of a Non-specific False Path Definition . . . . . . . . . . . . . . . . . . . . 234
Figure 7-23. Path Delay Launch and Capture Events . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 235
Figure 7-24. Robust Detection Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 237
Figure 7-25. Non-robust Detection Example. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 238
Figure 7-26. Functional Detection Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 239
Figure 7-27. Example Use of Transition_condition Statement. . . . . . . . . . . . . . . . . . . . . . . 241
Figure 7-28. Example of Ambiguous Path Definition . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 243
Figure 7-29. Example of Ambiguous Path Edges . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 243
Figure 7-30. On-chip Clock Generation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 247
Figure 7-31. PLL-Generated Clock and Control Signals. . . . . . . . . . . . . . . . . . . . . . . . . . . . 248
Figure 7-32. Cycles Merged for ATPG . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 250
Figure 7-33. Cycles Expanded for ATPG . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 251
Figure 7-34. Mux-DFF Example Design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 254
Figure 7-35. Mux-DFF Broadside Timing, Cell to Cell . . . . . . . . . . . . . . . . . . . . . . . . . . . . 254
Figure 7-36. Broadside Timing, Clock Pulses in Non-adjacent cycles . . . . . . . . . . . . . . . . . 256
Figure 7-37. Mux-DFF Cell to PO Timing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 257
Figure 7-38. Mux-DFF PI to Cell Timing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 258
Figure 7-39. Simplified Per-Cycle Clock Control Model . . . . . . . . . . . . . . . . . . . . . . . . . . . 259
Figure 7-40. Simplified Sequence Clock Control Model . . . . . . . . . . . . . . . . . . . . . . . . . . . 260
Figure 7-41. Clock Control Capture Cycles . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 261
Figure 7-42. Timing Slack . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 265
Figure 7-43. Testcase 1 Logic . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 270
Figure 7-44. Dofile. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 270
Figure 7-45. Reports . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 271
Figure 7-46. Glitch Detection Case . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 275
Figure 7-47. State Diagram of TAP Controller Circuitry . . . . . . . . . . . . . . . . . . . . . . . . . . . 277
Figure 7-48. Conceptual View of MacroTest . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 283

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 List of Figures

Figure 7-49. Basic Scan Pattern Creation Flow with MacroTest . . . . . . . . . . . . . . . . . . . . . 285
Figure 7-50. Mismatch Diagnosis Guidelines . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 305
Figure 7-51. Simulation Transcript . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 309
Figure 7-52. Automatic Simulation Mismatch Analysis Flow . . . . . . . . . . . . . . . . . . . . . . . 312
Figure 7-53. Manual Simulation Mismatch Analysis Flow. . . . . . . . . . . . . . . . . . . . . . . . . . 315
Figure 7-54. Clock-Skew Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 319
Figure 9-1. Core-level Pattern Generation Process . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 335
Figure 9-2. Scan Pattern Retargeting - Internal Mode. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 336
Figure 9-3. Pattern Generation - External Mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 337
Figure 9-4. Merging Patterns for Multiple Views of a Core at the Top Level . . . . . . . . . . . 346
Figure 9-5. Merging Patterns for Top Level and Core at the Top Level . . . . . . . . . . . . . . . . 347
Figure 9-6. Retargeting of Core-level Patterns . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 354
Figure 9-7. Three Levels of Wrapped Cores . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 359
Figure 9-8. Core-in-Core Retargeting Process Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . 360
Figure 10-1. Defining Basic Timing Process Flow . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 365
Figure 11-1. Example Scan Circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 386
Figure 13-1. Scan Chain Insertion by Power Domain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 404
Figure 13-2. Input Wrapper Cells and Power Domains. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 407
Figure 13-3. Output Wrapper Cells and Power Domains . . . . . . . . . . . . . . . . . . . . . . . . . . . 408
Figure 14-1. Using MTFI with Hierarchical Designs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 425
Figure 15-1. Full Hierarchical Block Netlist . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 428
Figure 15-2. Graybox Version of Block Netlist. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 429
Figure A-1. Basic Clock Gater Cell. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 435
Figure A-2. Two Types of Embedding for the Basic Clock Gater . . . . . . . . . . . . . . . . . . . . 436
Figure A-3. Type-B Clock Gater Causes Tracing Failure . . . . . . . . . . . . . . . . . . . . . . . . . . . 438
Figure A-4. Sample EDT Test Procedure Waveforms . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 438
Figure A-5. Two-level Clock Gating . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 439
Figure B-1. Design Used in State Stability Examples . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 447
Figure B-2. Typical Initialization Problem . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 449
Figure B-3. Three-bit Shift Register (Excerpted from Figure D-1). . . . . . . . . . . . . . . . . . . . 451
Figure B-4. Initialization with a Non-Shift Clock . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 452
Figure B-5. Clocking ff20 with a Pulse Generator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 454

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List of Tables

List of Tables

Table 2-1. Test Type/Fault Model Relationship . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35

Table 2-2. UDFM Keywords . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 43
Table 2-3. Fault Sub-classes . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 62
Table 4-1. RAM/ROM Commands . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 115
Table 5-1. Scan Direction and Active Values . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 126
Table 7-1. ATPG Constraint Conditions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 200
Table 7-2. Bridge Definition File Keywords . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 217
Table 7-3. Testcase 2 Data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 266
Table 12-1. Power Data Commands Directly Related to ATPG and DRC . . . . . . . . . . . . . 394
Table 12-2. Example Design With Four Power domains and Power Modes . . . . . . . . . . . . 397
Table 13-1. Power Data Commands Directly Related to ATPG . . . . . . . . . . . . . . . . . . . . . 403
Table 14-1. MTFI Command Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 425
Table 15-1. Graybox Command Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 427
Table A-1. Clock Gater Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 441

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 List of Tables

16 Tessent Scan and ATPG User’s Manual, v2014.3

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 Chapter 1
Overview

This manual gives an overview of ASIC/IC Design-for-Test (DFT) strategies and shows how to
use Mentor Graphics ASIC/IC DFT products as part of typical DFT design processes. This
manual discusses the Tessent products that use Scan and ATPG technology:
• Tessent Scan, which is Tessent Shell operating in “dft -scan” context
• Tessent FastScan, which is Tessent Shell operating in “patterns -scan” context
• Tessent TestKompress (with EDT off), which is Tessent Shell operating in “patterns
-scan” context
This manual uses the term “ATPG tool” to refer to any of the following products: Tessent
FastScan, Tessent TestKompress (with EDT off), or Tessent Shell operating in the “patterns
-scan” context.

For information about contexts in Tessent Shell, refer to “Contexts and System Modes” in the
Tessent Shell User’s Manual. For information about any of the commands mentioned in this
manual, refer to the Tessent Shell Reference Manual.

What is Design-for-Test?
Testability is a design attribute that measures how easy it is to create a program to
comprehensively test a manufactured design’s quality. Traditionally, design and test processes
were kept separate, with test considered only at the end of the design cycle. But in contemporary
design flows, test merges with design much earlier in the process, creating what is called a
design-for-test (DFT) process flow.
Testable circuitry is both controllable and observable. In a testable design, setting specific
values on the primary inputs results in values on the primary outputs that indicate whether or not
the internal circuitry works properly. To ensure maximum design testability, designers must
employ special DFT techniques at specific stages in the development process.

DFT Strategies
At the highest level, there are two main approaches to DFT: ad hoc and structured. The
following subsections discuss these DFT strategies.

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Overview
Top-Down Design Flow with DFT

Ad Hoc DFT
Ad hoc DFT implies using good design practices to enhance a design's testability without
making major changes to the design style.
Some ad hoc techniques include:

• Minimizing redundant logic

• Minimizing asynchronous logic
• Isolating clocks from the logic
• Adding internal control and observation points
Using these practices throughout the design process improves the overall testability of your
design. However, using structured DFT techniques with the Mentor Graphics DFT tools yields
far greater improvement. Thus, the remainder of this document concentrates on structured DFT
techniques.

Structured DFT
Structured DFT provides a more systematic and automatic approach to enhancing design
testability.
Structured DFT’s goal is to increase the controllability and observability of a circuit. Various
methods exist for accomplishing this. The most common is the scan design technique, which
modifies the internal sequential circuitry of the design. You can also use the Built-in Self-Test
(BIST) method, which inserts a device’s testing function within the device itself. Another
method is boundary scan, which increases board testability by adding circuitry to a chip.
“Understanding Scan and ATPG Basics” describes these methods in detail.

Top-Down Design Flow with DFT

The following text and figure shows the basic steps and the Mentor Graphics tools you would
use during a typical ASIC top-down design flow.
This document discusses those steps shown in grey; it also mentions certain aspects of other
design steps, where applicable. This flow is just a general description of a top-down design
process flow using a structured DFT strategy.

As Figure 1-1 shows, the first task in any design flow is creating the initial RTL-level design,
through whatever means you choose. In the Mentor Graphics environment, you may choose to
create a high-level Verilog description using ModelSim® or a schematic using Design
Architect®. You then verify the design’s functionality by performing a functional simulation,
using ModelSim or another vendor's Verilog simulator.

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 Overview
Top-Down Design Flow with DFT

At this point in the flow you are ready to insert internal scan circuitry into your design using
Tessent Scan.. You may then want to re-verify the timing because you added scan circuitry.
Once you are sure the design is functioning as desired, you can generate test patterns. You can
use the ATPG tool to generate a test pattern set in the appropriate format.

Now you should verify that the design and patterns still function correctly with the proper
timing information applied. You can use ModelSim or some other simulator to achieve this
goal. You may then have to perform a few additional steps required by your ASIC vendor
before handing the design off for manufacture and testing.

Note
It is important for you to check with your vendor early on in your design process for
specific requirements and restrictions that may affect your DFT strategies. For example,
the vendor's test equipment may only be able to handle single scan chains (see page 22),
have memory limitations, or have special timing requirements that affect the way you
generate scan circuitry and test patterns.

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Overview
Top-Down Design Flow with DFT

Figure 1-1. Top-Down Design Flow Tasks and Products

ModelSim
Create Initial
Text Editor
Design
Design Architect®

Verify a < =b+c; ModelSim

Functionality

1011
Insert/Verify
Built-in Self Test
Circuitry
P/F
Insert/Verify
Boundary Scan
Circuitry

Design Compiler® Synthesize/Optimize

BuildGates® Design
& Other Synthesis Tools
Insert/Verify PrimeTime®
Boundary Scan
Verify Timing & Other Timing
Circuitry Analysis Tools

Tessent Scan Insert Internal

Scan Circuitry

Re-verify Timing
(optional)

0 Tessent ATPG tool

Generate/Verify 1 ModelSim
Test Patterns 1
0

Hand off
to Vendor

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 Chapter 2
Understanding Scan and ATPG Basics

Before you begin the testing process, you must first have an understanding of certain testing
concepts. Once you understand these concepts, you can determine the best test strategy for your
particular design.
Figure 2-1 shows the concepts this section discusses.

Figure 2-1. Test Concepts

1. Understanding Scan Design

Understand
DFT Basics 2. Understanding ATPG
3. Understanding Test Types and Fault Models
Understand
Tool Concepts

Scan circuitry facilitates test generation and can reduce external tester usage. There are two
main types of scan circuitry: internal scan and boundary scan. Internal scan (also referred to as
scan design) is the internal modification of your design’s circuitry to increase its testability. A
detailed discussion of internal scan begins in “Internal Scan Circuitry.”

While scan design modifies circuitry within the original design, boundary scan adds scan
circuitry around the periphery of the design to make internal circuitry on a chip accessible via a
standard board interface. The added circuitry enhances board testability of the chip, the chip I/O
pads, and the interconnections of the chip to other board circuitry.

Understanding Scan Design

This section gives you an overview of scan design and how it works. For more detailed
information on the concepts presented in this section, refer to the documentation references
cited earlier.

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Understanding Scan and ATPG Basics
Understanding Scan Design

Internal Scan Circuitry

As previously discussed, internal scan (or scan design) is the internal modification of your
design’s circuitry to increase its testability.
Scan techniques are discussed on page 24.

Scan Design Overview

The goal of scan design is to make a difficult-to-test sequential circuit behave (during the testing
process) like an easier-to-test combinational circuit. Achieving this goal involves replacing
sequential elements with scannable sequential elements (scan cells) and then stitching the scan
cells together into scan registers, or scan chains. You can then use these serially-connected scan
cells to shift data in and out when the design is in scan mode.
The design shown in Figure 2-2 contains both combinational and sequential portions. Before
adding scan, the design had three inputs, A, B, and C, and two outputs, OUT1 and OUT2. This
“Before Scan” version is difficult to initialize to a known state, making it difficult to both
control the internal circuitry and observe its behavior using the primary inputs and outputs of
the design.

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 Understanding Scan and ATPG Basics
Understanding Scan Design

Figure 2-2. Design Before and After Adding Scan

Before Scan

A
Combinational Logic OUT1
B

D Q D Q D Q

CLK

C Combinational Logic OUT2

After Scan

A
Combinational Logic OUT1
B
sc_out
D Q D Q D Q
sc_in
sci sci sci
sen sen sen

CLK
sc_en

C Combinational Logic OUT2

After adding scan circuitry, the design has two additional inputs, sc_in and sc_en, and one
additional output, sc_out. Scan memory elements replace the original memory elements so that
when shifting is enabled (the sc_en line is active), scan data is read in from the sc_in line.

The operating procedure of the scan circuitry is as follows:

1. Enable the scan operation to allow shifting (to initialize scan cells).
2. After loading the scan cells, hold the scan clocks off and then apply stimulus to the
primary inputs.

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Understanding Scan and ATPG Basics
Understanding Scan Design

3. Measure the outputs.

4. Pulse the clock to capture new values into scan cells.
5. Enable the scan operation to unload and measure the captured values while
simultaneously loading in new values via the shifting procedure (as in
step 1).

Understanding Scan
Scan is a scan design methodology that replaces all memory elements in the design with their
scannable equivalents and then stitches (connects) them into scan chains. The idea is to control
and observe the values in all the design’s storage elements so you can make the sequential
circuit’s test generation and fault simulation tasks as simple as those of a combinational circuit.
Figure 2-3 gives a symbolic representation of a scan design.

Figure 2-3. Scan Representation

Scan Output

Scan Input

The black rectangles in Figure 2-3 represent scan elements. The line connecting them is the
scan path. Because this is a scan design, all storage elements were converted and connected in
the scan path. The rounded boxes represent combinational portions of the circuit.

For information on implementing a scan strategy for your design, refer to “Test Structures
Supported by Tessent Scan” on page 122.

Scan Benefits
The following are benefits of employing a scan strategy:
• Highly automated process.
Using scan insertion tools, the process for inserting scan circuitry into a design is highly-
automated, thus requiring very little manual effort.
• Highly-effective, predictable method.
Scan design is a highly-effective, well-understood, and well-accepted method for
generating high test coverage for your design.

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 Understanding Scan and ATPG Basics
Understanding Scan Design

• Ease of use.
Using scan methodology, you can insert both scan circuitry and run ATPG without the
aid of a test engineer.
• Assured quality.
Scan assures quality because parts containing such circuitry can be tested thoroughly
during chip manufacture. If your end products are going to be used in market segments
that demand high quality, such as aircraft or medical electronics—and you can afford
the added circuitry—then you should take advantage of the scan methodology.

Understanding Wrapper Chains

The ATPG process on very large, complex designs can often be unpredictable. This problem is
especially true for large sequential or partial scan designs. To reduce this unpredictability, a
number of hierarchical techniques for test structure insertion and test generation are beginning
to emerge. Creating wrapper chains is one of these techniques. Large designs that are split into
a number of design blocks benefit most from wrapper chains.
Wrapper chains add controllability and observability to the design via a hierarchical wrapper
scan chain. A wrapper chain is a series of scan cells connected around the boundary of a design
partition that is accessible at the design level. The wrapper chain improves both test coverage
and run time by converting sequential elements to scan cells at inputs (outputs) that have low
controllability (observability) from outside the block.

The architecture of wrapper chains is illustrated in the following two figures. Figure 2-4 shows
a design with three partitions, A, B, and C.

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Understanding Scan and ATPG Basics
Understanding Scan Design

Figure 2-4. Example of Partitioned Design

Design

Partition B

Design Design
Primary Primary
Inputs Partition A
Outputs

Partition C

The bold lines in Figure 2-4 indicate inputs and outputs of partition A that are not directly
controllable or observable from the design level. Because these lines are not directly accessible
at the design level, the circuitry controlled by these pins can cause testability problems for the
design.

Figure 2-5 shows how adding wrapper chain structures to partition A increases the
controllability and observability (testability) of partition A from the design level.

Note
Only the first elements that are directly connected to the uncontrollable (unobservable)
primary inputs (primary outputs) become part of the wrapper chain.

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 Understanding Scan and ATPG Basics
Understanding Scan Design

Figure 2-5. Wrapper Chains Added to Partition A

Design-Level
Partition A Scan Out
Design-Level Pin Added
Scan In
Pin Added

Uncontrollable
Inputs
Unobservable
Outputs

The wrapper chain consists of two types of elements: sequential elements connected directly to
uncontrolled primary inputs of the partition, and sequential elements connected directly to
unobservable (or masked) outputs of the partition. The partition also acquires two design-level
pins, scan in and scan out, to give direct access to the previously uncontrollable or unobservable
circuitry.

You can also use wrapper chains in conjunction with scan structures. Sequential elements not
eligible for wrapper chains become candidates for internal scan.

For information on implementing a scan strategy for your design, refer to “Setting Up for
Wrapper Chain Identification” on page 131.

Understanding Test Points

A design can contain a number of points that are difficult to control or observe. Sometimes this
is true even in designs containing scan. By adding special circuitry at certain locations called
test points, you can increase the testability of the design.
For example, Figure 2-6 shows a portion of circuitry with a controllability and observability
problem.

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Understanding Scan and ATPG Basics
Understanding Scan Design

Figure 2-6. Uncontrollable and Unobservable Circuitry

VCC
Fault Effects 1
Blocked From 1
Observation Uncontrollable
Circuitry

Test Point Test Point

(for Observation) (for Controllability)

In this example, one input of an OR gate is tied to a 1. This blocks the ability to propagate
through this path any fault effects in circuitry feeding the other input. Thus, the other input must
become a test point to improve observation. The tied input also causes a constant 1 at the output
of the OR gate. This means any circuitry downstream from that output is uncontrollable. The
pin at the output of the gate becomes a test point to improve controllability. Once identification
of these points occurs, added circuitry can improve the controllability and observability
problems.

Figure 2-7 shows circuitry added at these test points.

Figure 2-7. Testability Benefits from Test Point Circuitry

PO
VCC
Fault Effects 1
can now be 1 Circuitry can
Observed MUX now be
controlled
PI

Test_Mode

At the observability test point, an added primary output provides direct observation of the signal
value. At the controllability test point, an added MUX controlled by a test_mode signal and
primary input controls the value fed to the associated circuitry.

This is just one example of how test point circuitry can increase design testability. Refer to
“Manually Specifying Control and Observe Points” on page 134 for information on identifying
test points and inserting test point circuitry.

Test point circuitry is similar to test logic circuitry. For more information on test logic, refer to
“Enabling Test Logic Insertion” on page 124.

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 Understanding Scan and ATPG Basics
Understanding Scan Design

Test Structure Insertion with Tessent Scan

Tessent Scan, the Mentor Graphics internal scan synthesis tool, can identify sequential elements
for conversion to scan cells and then stitch those scan cells into scan chains.
Tessent Scan contains the following features:

• Verilog format.
Reads and writes a Verilog gate-level netlist.
• Multiple scan types.
Supports insertion of three different scan types, or methodologies: mux-DFF, clocked-
scan, and LSSD.
• Multiple test structures.
Supports identification and insertion of scan (both sequential ATPG-based and scan
sequential procedure-based), wrapper chains, and test points.
• Scannability checking.
Provides powerful scannability checking/reporting capabilities for sequential elements
in the design.
• Design rules checking.
Performs design rules checking to ensure scan setup and operation are correct—before
scan is actually inserted. This rules checking also guarantees that the scan insertion done
by Tessent Scan produces results that function properly in an ATPG tool.
• Interface to ATPG tools.
Automatically generates information for the ATPG tools on how to operate the scan
circuitry Tessent Scan creates.
• Optimal partial scan selection.
Provides optimal partial scan analysis and insertion capabilities.
• Flexible scan configurations.
Allows flexibility in the scan stitching process, such as stitching scan cells in fixed or
random order, creating either single- or multiple-scan chains, and using multiple clocks
on a single-scan chain.
• Test logic.
Provides capabilities for inserting test logic circuitry on uncontrollable set, reset, clock,
tri-state enable, and RAM read/write control lines.
• User specified pins.
Allows user-specified pin names for test and other I/O pins.
• Multiple model levels.
Handles gate-level, as well as gate/transistor-level models.
• Online help.
Provides online help for every command along with online manuals.

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Understanding Scan and ATPG Basics
Understanding ATPG

For information about using Tessent Scan to insert scan circuitry into your design, refer to
“Inserting Internal Scan and Test Circuitry” on page 119.

Understanding ATPG
ATPG stands for Automatic Test Pattern Generation. Test patterns, sometimes called test
vectors, are sets of 1s and 0s placed on primary input pins during the manufacturing test process
to determine if the chip is functioning properly. When the test pattern is applied, the Automatic
Test Equipment (ATE) determines if the circuit is free from manufacturing defects by
comparing the fault-free output—which is also contained in the test pattern—with the actual
output measured by the ATE.

The ATPG Process

The goal of ATPG is to create a set of patterns that achieves a given test coverage, where test
coverage is the total percentage of testable faults the pattern set actually detects.
For a more precise definition of test coverage, see page 65.

ATPG consists of two main steps: 1) generating patterns and, 2) performing fault simulation to
determine which faults the patterns detect. Mentor Graphics ATPG tools automate these two
steps into a single operation or ATPG process. This ATPG process results in patterns you can
then save with added tester-specific formatting that enables a tester to load the pattern data into
a chip’s scan cells and otherwise apply the patterns correctly.

This section discusses only the generation of test patterns. “Fault Classes” on page 55 discusses
the fault simulation process. The two most typical methods for pattern generation are random
and deterministic. Additionally, the ATPG tools can fault simulate patterns from an external set
and place those patterns detecting faults in a test set. The following subsections discuss each of
these methods.

Random Pattern Test Generation

An ATPG tool uses random pattern test generation when it produces a number of random
patterns and identifies only those patterns that detect faults. It then stores only those patterns in
the test pattern set. The type of fault simulation used in random pattern test generation cannot
replace deterministic test generation because it can never identify redundant faults. Nor can it
create test patterns for faults that have a very low probability of detection. However, it can be
useful on testable faults terminated by deterministic test generation. As an initial step, using a
small number of random patterns can improve ATPG performance.

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Deterministic Test Pattern Generation

An ATPG tool uses deterministic test pattern generation when it creates a test pattern intended
to detect a given fault. The procedure is to pick a fault from the fault list, create a pattern to
detect the fault, fault simulate the pattern, and check to make sure the pattern detects the fault.
More specifically, the tool assigns a set of values to control points that force the fault site to the
state opposite the fault-free state, so there is a detectable difference between the fault value and
the fault-free value. The tool must then find a way to propagate this difference to a point where
it can observe the fault effect. To satisfy the conditions necessary to create a test pattern, the test
generation process makes intelligent decisions on how best to place a desired value on a gate. If
a conflict prevents the placing of those values on the gate, the tool refines those decisions as it
attempts to find a successful test pattern.

If the tool exhausts all possible choices without finding a successful test pattern, it must perform
further analysis before classifying the fault. Faults requiring this analysis include redundant,
ATPG-untestable, and possible-detected-untestable categories (see page 55 for more
information on fault classes). Identifying these fault types is an important by-product of
deterministic test generation and is critical to achieving high test coverage. For example, if a
fault is proven redundant, the tool may safely mark it as untestable. Otherwise, it is classified as
a potentially detectable fault and counts as an untested fault when calculating test coverage.

External Pattern Test Generation

An ATPG tool uses external pattern test generation when the preliminary source of ATPG is a
pre-existing set of external patterns. The tool analyzes this external pattern set to determine
which patterns detect faults from the active fault list. It then places these effective patterns into
an internal test pattern set. The “generated patterns”, in this case, include the patterns (selected
from the external set) that can efficiently obtain the highest test coverage for the design.

Mentor Graphics ATPG Applications

Tessent FastScan and Tessent TestKompress (EDT off) are the Mentor Graphics scan sequential
ATPG products, and are the same thing as Tessent Shell operating in “patterns -scan” context.
This manual refers to all three of these products as the “ATPG tool.”
The following subsections introduce the features of Tessent Shell operating in “patterns -scan”
context. “Generating Test Patterns” on page 167” discusses the ATPG products in greater
detail.

Scan Sequential ATPG with the ATPG Tool

Mentor Graphics ATPG products include the following features:
• Deliver high performance ATPG for designs with structured scan
• Reduce run time with no effect on coverage or pattern count using distributed ATPG

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• Maximize test coverage by minimizing the impact of X’s caused by false and multicycle
paths
• Identify testability problems early using comprehensive design rule checking
• Reduce test validation time with automatic simulation mismatch debugging
• Ensure shorter time to market with integration into all design flows and foundry support
• Have extensive fault model support, including stuck-at, IDDQ, transition, path delay and
bridge
• Have on-chip PLL support for accurate at-speed test
• Automate testing small embedded memories and cores with scan
• Supported in the Tessent SoCScan hierarchical silicon test environment
For more information about ATPG functionality, refer to the Tessent Shell Reference Manual.

Understanding Test Types and Fault Models

A manufacturing defect is a physical problem that occurs during the manufacturing process,
causing device malfunctions of some kind. The purpose of test generation is to create a set of
test patterns that detect as many manufacturing defects as possible.
Figure 2-8 gives an example of possible device defect types.

Figure 2-8. Manufacturing Defect Space for a Design

Functional
Defects circuitry opens
circuitry shorts

At-Speed
IDDQ Defects
Defects bridges
CMOS stuck-on slow transistors
CMOS stuck-open

Each of these defects has an associated detection strategy. The following subsection discusses
the three main types of test strategies.

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Test Types
The previous figure shows three main categories of defects and their associated test types:
functional, IDDQ, and at-speed. Functional testing checks the logic levels of output pins for a
“0” and “1” response. IDDQ testing measures the current going through the circuit devices. At-
speed testing checks the amount of time it takes for a device to change logic states. The
following subsections discuss each of these test types in more detail.

Functional Test
Functional test continues to be the most widely-accepted test type. Functional test typically
consists of user-generated test patterns, simulation patterns, and ATPG patterns.
Functional testing uses logic levels at the device input pins to detect the most common
manufacturing process-caused problem, static defects (for example, open, short, stuck-on, and
stuck-open conditions). Functional testing applies a pattern of 1s and 0s to the input pins of a
circuit and then measures the logical results at the output pins. In general, a defect produces a
logical value at the outputs different from the expected output value.

IDDQ Test
IDDQ testing measures quiescent power supply current rather than pin voltage, detecting device
failures not easily detected by functional testing—such as CMOS transistor stuck-on faults or
adjacent bridging faults. IDDQ testing equipment applies a set of patterns to the design, lets the
current settle, then measures for excessive current draw. Devices that draw excessive current
may have internal manufacturing defects.
Because IDDQ tests do not have to propagate values to output pins, the set of test vectors for
detecting and measuring a high percentage of faults may be very compact. The ATPG tool
efficiently creates this compact test vector set.

In addition, IDDQ testing detects some static faults, tests reliability, and reduces the number of
required burn-in tests. You can increase your overall test coverage by augmenting functional
testing with IDDQ testing.

IDDQ test generation methodologies break down into these categories:

• Every-vector
This methodology monitors the power-supply current for every vector in a functional or
stuck-at fault test set. Unfortunately, this method is relatively slow—on the order of 10-
100 milliseconds per measurement—making it impractical in a manufacturing
environment.
• Supplemental
This methodology bypasses the timing limitation by using a smaller set of IDDQ
measurement test vectors (typically generated automatically) to augment the existing
test set.

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Three test vector types serve to further classify IDDQ test methodologies:

• Ideal
Ideal IDDQ test vectors produce a nearly zero quiescent power supply current during
testing of a good device. Most methodologies expect such a result.
• Non-ideal
Non-ideal IDDQ test vectors produce a small, deterministic quiescent power supply
current in a good circuit.
• Illegal
If the test vector cannot produce an accurate current component estimate for a good
device, it is an illegal IDDQ test vector. You should never perform IDDQ testing with
illegal IDDQ test vectors.
IDDQ testing classifies CMOS circuits based on the quiescent-current-producing circuitry
contained inside as follows:

• Fully static
Fully static CMOS circuits consume close to zero IDDQ current for all circuit states.
Such circuits do not have pullup or pull-down resistors, and there can be one and only
one active driver at a time in tri-state buses. For such circuits, you can use any vector for
ideal IDDQ current measurement.
• Resistive
Resistive CMOS circuits can have pullup/pull-down resistors and tristate buses that
generate high IDDQ current in a good circuit.
• Dynamic
Dynamic CMOS circuits have macros (library cells or library primitives) that generate
high IDDQ current in some states. Diffused RAM macros belong to this category.
Some designs have a low current mode, which makes the circuit behave like a fully static
circuit. This behavior makes it easier to generate ideal IDDQ tests for these circuits.

The ATPG tool currently supports only the ideal IDDQ test methodology for fully static,
resistive, and some dynamic CMOS circuits. The tools can also perform IDDQ checks during
ATPG to ensure the vectors they produce meet the ideal requirements. For information on
creating IDDQ test sets, refer to “Creating an IDDQ Test Set”.

At-Speed Test
Timing failures can occur when a circuit operates correctly at a slow clock rate, and then fails
when run at the normal system speed. Delay variations exist in the chip due to statistical
variations in the manufacturing process, resulting in defects such as partially conducting
transistors and resistive bridges.
The purpose of at-speed testing is to detect these types of problems. At-speed testing runs the
test patterns through the circuit at the normal system clock speed.

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Fault Modeling
Fault models are a means of abstractly representing manufacturing defects in the logical model
of your design. Each type of testing—functional, IDDQ, and at-speed—targets a different set of
defects.

Test Types and Associated Fault Models

The following table associates test types, fault models, and the types of manufacturing defects
targeted for detection.

Table 2-1. Test Type/Fault Model Relationship

Test Type Fault Model Examples of Mfg. Defects Detected
Functional Stuck-at, toggle Some opens/shorts in circuit interconnections
IDDQ Pseudo stuck-at CMOS transistor stuck-on/some stuck-open
conditions, resistive bridging faults, partially
conducting transistors
At-speed Transition, path Partially conducting transistors, resistive bridges
delay

Fault Locations
By default, faults reside at the inputs and outputs of library models. However, faults can instead
reside at the inputs and outputs of gates within library models if you turn internal faulting on.
Figure 2-9 shows the fault sites for both cases.

Figure 2-9. Internal Faulting Example

Library Model Library Model

a n0 a n0
b b
z z
c c
d n1 d n1
set_internal_fault On set_internal_fault Off
(default)
= fault sites

To locate a fault site, you need a unique, hierarchical instance pathname plus the pin name.

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You can also use Verilog `celldefine statements to extend cell boundaries beyond library
models. Using this technique has several implications:

• The default fault population changes. By default, all fault locations are at library
boundary pins. However, when the library boundary moves from the ATPG library level
up to the `celldefine level, the fault locations and fault population change as a result.
• The flattened model can be different because the logic inside `celldefine module might
be optimized to reduce the flattened model size.
• Hierarchical instance/pin names inside `celldefine module are not treated as legal
instance/pin names.

Fault Collapsing
A circuit can contain a significant number of faults that behave identically to other faults. That
is, the test may identify a fault, but may not be able to distinguish it from another fault. In this
case, the faults are said to be equivalent, and the fault identification process reduces the faults to
one equivalent fault in a process known as fault collapsing. For performance reasons, early in
the fault identification process the ATPG tool singles out a member of the set of equivalent
faults and use this “representative” fault in subsequent algorithms. Also for performance
reasons, these applications only evaluate the one equivalent fault, or collapsed fault, during
fault simulation and test pattern generation. The tools retain information on both collapsed and
uncollapsed faults, however, so they can still make fault reports and test coverage calculations.

Supported Fault Model Types

The ATPG tool supports stuck-at, pseudo stuck-at, toggle, and transition fault models. In
addition to these, the tool supports static bridge and path delay fault models. The following
subsections discuss these supported fault models, along with their fault collapsing rules.
In addition to the above standard fault models, the ATPG tool supports User-Defined Fault
Models (UDFM). For more information, refer to “User-Defined Fault Modeling” on page 42.

Functional Testing and the Stuck-At Fault Model

Functional testing uses the single stuck-at model, the most common fault model used in fault
simulation, because of its effectiveness in finding many common defect types. The stuck-at
fault models the behavior that occurs if the terminals of a gate are stuck at either a high (stuck-
at-1) or low (stuck-at-0) voltage. The fault sites for this fault model include the pins of primitive
instances.
Figure 2-10 shows the possible stuck-at faults that could occur on a single AND gate.

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Figure 2-10. Single Stuck-At Faults for AND Gate

a
c
b

Possible Errors: 6
“a” s-a-1, “a” s-a-0
“b” s-a-1, “b” s-a-0
“c” s-a-1, “c” s-a-0

For a single-output, n-input gate, there are 2(n+1) possible stuck-at errors. In this case, with
n=2, six stuck-at errors are possible.

The ATPG tool uses the following fault collapsing rules for the single stuck-at model:

• Buffer - input stuck-at-0 is equivalent to output stuck-at-0. Input stuck-at-1 is equivalent

to output stuck-at-1.
• Inverter - input stuck-at-0 is equivalent to output stuck-at-1. Input stuck-at-1 is
equivalent to output stuck-at-0.
• AND - output stuck-at-0 is equivalent to any input stuck-at-0.
• NAND - output stuck-at-1 is equivalent to any input stuck-at-0.
• OR - output stuck-at-1 is equivalent to any input stuck-at-1.
• NOR - output stuck-at-0 is equivalent to any input stuck-at-1.
• Net between single output pin and single input pin - output pin stuck-at-0 is
equivalent to input pin stuck-at-0. Output pin stuck-at-1 is equivalent to input pin stuck-
at-1.

Functional Testing and the Toggle Fault Model

Toggle fault testing ensures that a node can be driven to both a logical 0 and a logical 1 voltage.
This type of test indicates the extent of your control over circuit nodes. Because the toggle fault
model is faster and requires less overhead to run than stuck-at fault testing, you can experiment
with different circuit configurations and get a quick indication of how much control you have
over your circuit nodes.
The ATPG tool uses the following fault collapsing rules for the toggle fault model:

• Buffer - a fault on the input is equivalent to the same fault value at the output.
• Inverter - a fault on the input is equivalent to the opposite fault value at the output.

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• Net between single output pin and multiple input pin - all faults of the same value are
equivalent.

IDDQ Testing and the Pseudo Stuck-At Fault Model

IDDQ testing, in general, can use several different types of fault models, including node toggle,
pseudo stuck-at, transistor leakage, transistor stuck, and general node shorts.
The ATPG tool supports the pseudo stuck-at fault model for IDDQ testing. Testing detects a
pseudo stuck-at model at a node if the fault is excited and propagated to the output of a cell
(library model instance or primitive). Because library models can be hierarchical, fault
modeling occurs at different levels of detail.

The pseudo stuck-at fault model detects all defects found by transistor-based fault models—if
used at a sufficiently low level. The pseudo stuck-at fault model also detects several other types
of defects that the traditional stuck-at fault model cannot detect, such as some adjacent bridging
defects and CMOS transistor stuck-on conditions.

The benefit of using the pseudo stuck-at fault model is that it lets you obtain high defect
coverage using IDDQ testing, without having to generate accurate transistor-level models for all
library components.

The transistor leakage fault model is another fault model commonly used for IDDQ testing.
This fault model models each transistor as a four terminal device, with six associated faults. The
six faults for an NMOS transistor include G-S, G-D, D-S, G-SS, D-SS, and S-SS (where G, D,
S, and SS are the gate, drain, source, and substrate, respectively).

You can only use the transistor level fault model on gate-level designs if each of the library
models contains detailed transistor level information. Pseudo stuck-at faults on gate-level
models equate to the corresponding transistor leakage faults for all primitive gates and fanout-
free combinational primitives. Thus, without the detailed transistor-level information, you
should use the pseudo stuck-at fault model as a convenient and accurate way to model faults in
a gate-level design for IDDQ testing.

Figure 2-11 shows the IDDQ testing process using the pseudo stuck-at fault model.

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Figure 2-11. IDDQ Fault Testing

IDD

VDD 2) Measure IDDQ

1) Apply Input Patterns

VSS

The pseudo stuck-at model detects internal transistor shorts, as well as “hard” stuck-ats (a node
actually shorted to VDD or GND), using the principle that current flows when you try to drive
two connected nodes to different values. While stuck-at fault models require propagation of the
fault effects to a primary output, pseudo stuck-at fault models allow fault detection at the output
of primitive gates or library cells.

IDDQ testing detects output pseudo stuck-at faults if the primitive or library cell output pin goes
to the opposite value. Likewise, IDDQ testing detects input pseudo stuck-at faults when the
input pin has the opposite value of the fault and the fault effect propagates to the output of the
primitive or library cell.

By combining IDDQ testing with traditional stuck-at fault testing, you can greatly improve the
overall test coverage of your design. However, because it is costly and impractical to monitor
current for every vector in the test set, you can supplement an existing stuck-at test set with a
compact set of test vectors for measuring IDDQ. This set of IDDQ vectors can either be
generated automatically or intelligently chosen from an existing set of test vectors. Refer to
section “Creating an IDDQ Test Set” on page 211 for information.

The fault collapsing rule for the pseudo stuck-at fault model is as follows: for faults associated
with a single cell, pseudo stuck-at faults are considered equivalent if the corresponding stuck-at
faults are equivalent.

set_transition_holdpi — Freezes all primary inputs values other than clocks and RAM
controls during multiple cycles of pattern generation.

The Static Bridge Fault Model

Certain pairs of nets in a design have specific characteristics that make them vulnerable to
bridging. The static bridge fault model is used by the ATPG tool to test against potential bridge

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sites (net pairs) extracted from the design. You can load the bridge sites from a bridge definition
file or from the Calibre query server output file.
For more information, see “Net Pair Identification with Calibre for Bridge Fault Test Patterns”
on page 214.

This model uses a 4-Way Dominant fault model that works by driving one net (dominant) to a
logic value and ensuring that the other net (follower) can be driven to the opposite value.

Let sig_A and sig_B be two nets in the design. If sig_A and sig_B are bridged together, the
following faulty relationships exist:

• sig_A is dominant with a value of 0 (sig_A=0; sig_B=1/0)

• sig_A is dominant with a value of 1 (sig_A=1; sig_B=0/1)
• sig_B is dominant with a value of 0 (sig_B=0; sig_A=1/0)
• sig_B is dominant with a value of 1 (sig_B=1; sig_A=0/1)
By default, the tool creates test patterns that test each net pair against all four faulty
relationships.

At-Speed Testing and the Transition Fault Model

Transition faults model large delay defects at gate terminals in the circuit under test. The
transition fault model behaves as a stuck-at fault for a temporary period of time. The slow-to-
rise transition fault models a device pin that is defective because its value is slow to change
from a 0 to a 1. The slow-to-fall transition fault models a device pin that is defective because its
value is slow to change from a 1 to a 0.
Figure 2-12 demonstrates the at-speed testing process using the transition fault model. In this
example, the process could be testing for a slow-to-rise or slow-to-fall fault on any of the pins of
the AND gate.

Figure 2-12. Transition Fault Detection Process

1) Apply Initialization Vector

3) Wait Allotted Time

4) Measure Primary
Output Value
2) Apply Transition
Propagation Vector

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A transition fault requires two test vectors for detection: an initialization vector and a transition
propagation vector. The initialization vector propagates the initial transition value to the fault
site. The transition vector, which is identical to the stuck-at fault pattern, propagates the final
transition value to the fault site. To detect the fault, the tool applies proper at-speed timing
relative to the second vector, and measures the propagated effect at an external observation
point.

The tool uses the following fault collapsing rules for the transition fault model:

• Buffer - a fault on the input is equivalent to the same fault value at the output.
• Inverter - a fault on the input is equivalent to the opposite fault value at the output.
• Net between single output pin and single input pin - all faults of the same value are
equivalent.
set_fault_type — Specifies the fault model for which the tool develops or selects ATPG
patterns. The transition option for this command specifies the tool to develop or select ATPG
patterns for the transition fault model.

For more information on generating transition test sets, refer to “Creating a Transition Delay
Test Set” on page 220.

At-Speed Testing and the Path Delay Fault Model

Path delay faults model defects in circuit paths. Unlike the other fault types, path delay faults do
not have localized fault sites. Rather, they are associated with testing the combined delay
through all gates of specific paths (typically critical paths).
Path topology and edge type identify path delay faults. The path topology describes a user-
specified path from beginning, or launch point, through a combinational path to the end, or
capture point. The launch point is either a primary input or a state element. The capture point is
either a primary output or a state element. State elements used for launch or capture points are
either scan elements or non-scan elements that qualify for clock-sequential handling. A path
definition file defines the paths for which you want patterns generated. The add_faults
command specifies the edge type for path delay faults.

The edge type defines the type of transition placed on the launch point that you want to detect at
the capture point. A “0” indicates a rising edge type, which is consistent with the slow-to-rise
transition fault and is similar to a temporary stuck-at-0 fault. A “1” indicates a falling edge type,
which is consistent with the slow-to-fall transition fault and is similar to a temporary stuck-at-1
fault.

The ATPG tool targets multiple path delay faults for each pattern it generates. Within the
(ASCII) test pattern set, patterns that detect path delay faults include comments after the pattern
statement identifying the path fault, type of detection, time and point of launch event, time and
point of capture event, and the observation point. Information about which paths were detected
by each pattern is also included.

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For more information on generating path delay test sets, refer to “Creating a Path Delay Test
Set” on page 234.

User-Defined Fault Modeling

The standard fault models supported by the tool are typically sufficient to create a highly
efficient pattern set that detects the majority of potential defects. However, a number of defects
are not covered by these models and are detected only by accident; these defects require specific
conditions that cannot be defined for the existing fault models.
You can use user-defined fault models (UDFMs) to define custom fault models. UDFMs extend
the natively-supported fault models (primarily stuck-at and transition) by adding combinational
or sequential constraints on other pins/nets. These custom models enable you to generate
specific test patterns for process-related defects like intra-cell bridges or any other kind of
defect that requires additional constraints. You can also use UDFMs to define required
conditions in an additional fault model to reduce the need for functional test patterns.

Specifically, UDFMs provide the following capabilities:

• Support for generating high compact pattern sets

• Close integration with the existing ATPG flow
• Support of static (stuck-at) and delay (transition) fault models
• Definition of test alternatives
• Definition of additional single or multiple-cycle conditions
• Definition on library or hierarchy levels
• Definition on specific instances
• Write and load of fault status information using the MTFI format. For more information,
refer to “Using MTFI Files” on page 415.

Restrictions
The following restrictions apply to UDFMs:
• No support for any kind of fault collapsing.
• No support for fault protection (set_fault_protection command).
• No support for multiple detection during pattern generation (set_multiple_detection
command).
• No support for the analyze fault functionality (analyze_fault command).

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• Separation of static and delay fault models. Both fault types cannot be handled together
in one pattern generation step. The definitions must be handled in different pattern
generation runs; therefore, this leads to multiple pattern sets.

Note
UDFM requires additional memory. The amount of additional memory needed depends
on the number of UDFM definitions and their complexity.

UDFM File Format

Input for: read_fault_sites
The User-Defined Fault Model (UDFM) file is an ASCII file that models custom fault sites for
internal library cells. The UDFM file is created manually or is created using Tessent
CellModelGen. UDFM provides a method for expanding native ATPG faults such as stuck-at to
exhaustively test library cell models.
Format
A UDFM file must conform to the following syntax rules:

• Precede each line of comment text with a pair of backslashes (//).

• Keywords are not case-sensitive.
• Use a colon sign to define a value for a keyword.
• Enclose all string values in double quotation marks (" ").
• Enclose all UDFM declarations in braces ({}).
• Separate each entry within the UDFM declaration with a semicolon (;).
Parameters
In a UDFM file, you use keywords to define the fault models you are creating and their
behavior. Table 2-2 specifies version 1.0 of the UDFM syntax.

Table 2-2. UDFM Keywords

Keyword Description
UDFM Required. Specifies version 1.0 of the UDFM syntax. The syntax version
number must precede the UDFM declaration.
UdfmType Required. Specifies a user-defined identifier for all faults declared within the
UDFM file. This identifier allows you to target the group of faults declared in
a UDFM file for an ATPG run or other test pattern manipulations. The fol-
lowing commands allow you to use the UdfmType identifier as an argument:
add_faults, delete_faults, report_fault_sites, report_faults, and write_faults.

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Table 2-2. UDFM Keywords

Keyword Description
Cell Optional. Maps a set of defined faults to a specified library cell. When the
UDFM is loaded, the defined faults are applied to all instances of the speci-
fied library cell. The cell definition supports only input and output ports.
Module Optional. Maps a set of defined faults to a specified module. When the
UDFM is loaded, the defined faults are applied to all instances of the speci-
fied module. The specified value can be a library cell or a module at any hier-
archy in the design.
Instance Optional. Maps a set of defined faults to a specified instance. When the
UDFM is loaded, the defined faults are applied to the specified instance.
Complete pathname must be specified either the full hierarchical path (start-
ing with “/”) or relative to the instance path.
Fault Required. Defines one or more faults. The definition contains two parts:
• Fault identification string — name used to reference the fault.
• List of test alternatives — When more than one test is listed, the fault is
tested if any test alternatives are satisfied.
Test Required. Specifies the fault conditions to test for. You must minimally spec-
ify the defect type with the faulty value and, optionally, a list of conditions.
For more information, see the StaticFault or Delay Fault keyword.
Conditions Optional. Defines the nodes and values that represent all fault conditions. To
fulfill a condition, the specified node must be set to the specified value:
• 0: state low
• 1: state high
• —: no condition
StaticFault or One of these keywords is required, except when you also use the
DelayFault “Observation” keyword set to “false.” Specifies the faulty value at the speci-
fied location. If this statement applies to a cell, you can specify only cell
ports; if it applies to module definitions, you can specify only ports and mod-
ule internal elements.
Observation Optional. Allows you to disable fault propagation as required by faults mod-
els like toggle bridge. By default, fault propagation is enabled. Valid argu-
ments are “true” or “false.”
Note that when you disable observation (observation:false), the definition of
the UDFM faulty value (StaticFault or DelayFault) becomes optional. This
attribute is supported only on the “UDFMType” level and disables the fault
propagation for all fault models that follow.
Properties Optional. Stores information for future uses or by external user tools.
EncryptedTest Optional. Automatically used by Tessent CellModelGen to protect the IP. For
more information, refer to the Tessent CellModelGen Tool Reference.

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Example UDFM Files

Intra-cell Bridge Fault Example
The following is an example that tests for a cell internal bridge fault of a 2-to-1 multiplexer
(MUX21) from the CMOS123_std library. This example specifies two tests with the required
activation conditions on the input pins for observing on pin Z the effects of the bridge fault.

MUX21

D0 IN0
OUT Z
D1 IN1
CNT

UDFM {
version : 1;
Properties {
"library-name" : "CMOS123_std";
"create-by" : "my tool 1.0";
}
UdfmType ("intra-cell-bridges") {
Cell ("MUX21") {
Fault ("myFlt-N1-Z") {
Test {
StaticFault {"Z" : 1;}
Conditions {"D0" : 0; "D1" : 0; "S" : 0;}
}
Test {
StaticFault {"Z" : 0;}
Conditions {"D0" : 0; "D1" : 1; "S" : 0;}
}
}
}
}
}

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Delay Fault Example

The following is an example that tests for delay faults on the input pins of a 2-to-1 multiplexer
(MUX21, shown in the previous example). This example specifies a test where the D0 input
transitions from 1 to 0 while observing that the Z output is faulty if it remains at 1.

UDFM {
version : 1;
UdfmType ("Special-delay") {
Cell ("MUX21") {
Fault ("myFlt-D0") {
Test {
DelayFault {"Z" : 1;}
Conditions {"D0" : 10; "D1" : 00; "S" : 00;}
}
}
}
}
}

Instance-based Fault Example

The following is an example that tests for a bridge fault on the input pins of a single generic
2-input instance. This example specifies two tests for observing on pin Z the effects of a bridge
fault on the input pins.
/top/mod1/inst4
A IN0
OUT Z
B IN1

UDFM {
version : 1;
UdfmType ("intra-cell-bridges") {
Instance ("/top/mod1/inst4") {
Fault ("f001") {
Test{ StaticFault{"Z":0;} Conditions{"A":0;"B":1;} }
}
Fault ("f002") {
Test{ StaticFault{"Z":1;} Conditions{"A":1;"B":1;} }
}
}
}
}

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Toggle Bridges Example

The following is an example that tests for two specific toggle bridge faults within an entire
design. This example tests for bridge faults between the following two sets of nets:

Bridge1: /top/mod2/i3/y /top/mod2/i5/a

Bridge2: /top/mod1/i47/z /top/mod1/iop/c
This example specifies two tests to detect each bridge fault by ensuring that each net pair can be
set to different values simultaneously.

UDFM {
Version:2;
UdfmType("toggle-bridges") {
Observation:false;
Instance("/") {
Fault("Bridge1") {
Test{Conditions{"top/mod2/i3/y":1; "top/mod2/i5/a":0;}}
Test{Conditions{"top/mod2/i3/y":0; "top/mod2/i5/a":1;}}
}
Fault("Bridge2") {
Test{Conditions{"top/mod1/i47/z":1; "top/mod1/iop/c":0;}}
Test{Conditions{"top/mod1/i47/z":0; "top/mod1/iop/c":1;}}
}
}
}
}

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Creating a User-Defined Fault Model

The following procedure describes how you can manually define your own fault models.

Prerequisites
• Text editor

Procedure
Note
In this procedure, entries added in the current step are shown in bold.

1. In a text editor, enter the UDFM statement as shown here:

UDFM { }

All of the additional statements you use to define all fault models will be contained
within the curly brackets of this statement.
2. Add the UDFM version number statement next.
UDFM {
version : 1;
}

3. Add the UdfmType keyword to the ASCII file under the version number statement to
create a fault model name.
UDFM {
version : 1; // Syntax version ensures future compatibility
UdfmType ("udfm_fault_model") {
}
}

4. Specify the type of object you want to attach the fault to using the Cell, Module, and
Instance keywords. For more information on these keywords, see “UDFM Keywords.”
UDFM {
version : 1;
UdfmType ("udfm_fault_model") {
Cell ("MUX21") {
}
}
}

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5. Define a unique fault model name using the Fault keyword. You can define multiple
faults.
UDFM {
version : 1;
UdfmType ("udfm_fault_model") {
Cell ("MUX21") {
Fault ("myFlt-D0") {
}
}
}
}

6. Define how the defect can be tested using the Test, StaticFault or Delay Fault, and
Conditions keywords. Notice that the following example also shows how to use the
conditions statement for a single assignment or a list of assignments. See “UDFM
Keywords” for the complete list of keywords.
UDFM {
version : 1;
UdfmType ("udfm_fault_model") {
Cell ("MUX21") {
Fault ("myFlt-N1-Z") {
Test {
StaticFault {"Z" : 1;}
Conditions {"D0" : 0; "D1" : 0; "S" : 0;}
}
Test {
StaticFault {"Z" : 0;}
Conditions {"D0" : 0; "D1" : 1; "S" : 0;}
}
}
}
}
}

7. Save the file using a .udfm extension.

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Generating UDFM Test Patterns

UDFM test pattern generation is similar to pattern generation for bridge logic faults. In both
cases, fault definitions are accessed from an external file and the internal fault list is built using
that data. This procedure describes the differences needed for UDFM pattern generation.
UDFM test pattern generation for static fault models is similar to test pattern generation for
stuck-at faults. As a result, most ATPG settings for stuck-at faults can be used for UDFM faults.
However, because UDFM fault definitions must be imported, the following commands will
work slightly differently than they do for stuck-at faults.

add_faults delete_fault_sites report_faults set_fault_type

delete_faults read_fault_sites report_udfm_statistics write_faults

Note
The process for generating cell-aware test patterns uses the same flow and commands
described in this section.

Prerequisites
• UDFM file that contains fault definitions.

Procedure
1. Set the fault type to UDFM using the set_fault_type command with the UDFM option:
set_fault_type udfm

Note
When you change the fault type, the current fault list and internal test pattern set are
deleted.

2. Load the fault definitions from a specified UDFM file into your current tool session
using the read_fault_sites command:
read_fault_sites <filename>.udfm

3. Create the internal fault list using all of the fault definitions from the UDFM file with the
add_faults command:
add_faults -All

4. Generate test patterns using the create_patterns command:

create_patterns

For more information, refer to the Tessent Shell Reference Manual.

5. Print out the resulting statistics using the report_statistics command.

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report_statistics

For more information, refer to the Tessent Shell Reference Manual.

Related Topics
set_fault_type

Multiple Detect
The basic idea of multiple detect (n-detect) is to randomly target each fault multiple times. By
changing the way the fault is targeted and the other values in the pattern set, the potential to
detect a bridge increases. This approach starts with a standard stuck-at or transition pattern set.
Each fault is graded for multiple detect. Additional ATPG is then performed and patterns
created targeting the faults that have lower than the multiple detect target threshold.

Bridge Coverage Estimate

Bridge coverage estimate (BCE) is a metric for reporting the ability of multiple detect to
statistically detect a bridge defect. If a bridge fault exists between the target fault site and
another net then there is a 50% change of detecting the fault with one pattern. If a second
randomly different pattern targets the same fault then the probability of detecting the bridge is
1 - 0.52. BCE performs this type of calculation for all faults in the target list and will always be
lower than the test coverage value.
The following shows the statistics report that is automatically produced when multiple detection
or embedded multiple detect is enabled:

Statistics Report
Transition Faults
---------------------------------------------------------------
Fault Classes #faults
(total)
---------------------------- ---------------------------------
FU (full) 1114
-------------------------- ---------------------------------
DS (det_simulation) 1039 (93.27%)
UU (unused) 30 ( 2.69%)
TI (tied) 4 ( 0.36%)
RE (redundant) 3 ( 0.27%)
AU (atpg_untestable) 38 ( 3.41%)
---------------------------------------------------------------
Coverage
--------------------------
test_coverage 96.47%
fault_coverage 93.27%
atpg_effectiveness 100.00%
---------------------------------------------------------------
#test_patterns 130
#clock_sequential_patterns 130
#simulated_patterns 256

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CPU_time (secs) 0.5

---------------------------------------------------------------
Multiple Detection Statistics
---------------------------------------------------------------
Detections DS Faults Test Coverage
(N) (Detection == N) (Detection >= N)
---------------------------- ---------------- ----------------
1 0 ( 0.00%) 1039 ( 96.47%)
2 0 ( 0.00%) 1039 ( 96.47%)
3 93 ( 8.35%) 1039 ( 96.47%)
4 56 ( 5.03%) 946 ( 87.84%)
5 54 ( 4.85%) 890 ( 82.64%)
6 56 ( 5.03%) 836 ( 77.62%)
7 50 ( 4.49%) 780 ( 72.42%)
8 46 ( 4.13%) 730 ( 67.78%)
9 54 ( 4.85%) 684 ( 63.51%)
10+ 630 ( 56.55%) 630 ( 58.50%)
---------------------------------------------------------------
bridge_coverage_estimate 94.71%
---------------------------------------------------------------

The report includes the BCE value and the number of faults with various detects less than the
target detection value.

Embedded Multiple Detect

Embedded multiple detect (EMD) is a method of improving the multiple detect of a pattern set
without increasing the number of patterns within that pattern set. Essentially, EMD produces the
same quality patterns as a standard pattern set but in addition adds improved multiple detection
for free. The only cost for the extra multiple detection is a longer run time during pattern
creation of about 30% to 50%. As a result, EMD is often considered a no-cost additional value
and used for ATPG.
When performing ATPG, the tool tries to detect as many previously undetected faults in parallel
within the same pattern as possible. However, even though ATPG maximizes the number of
previously undetected faults detected per pattern, only a small percentage of scan cells will have
specific values necessary for the detection. These specified bits need to be loaded in scan cells
for that pattern are referred to as the test cube. The remaining scan cells that are not filled with
test cube values are randomly filled for fortuitous detection of untargeted faults. EMD uses the
same ATPG starting point to produce a test cube but then determines if there are some faults
that previously had a low number of detections. For these faults, EMD will put additional scan
cell values added to the test cube to improve multiple detection on top of the new detection
pattern.

EMD will have a multiple detection that is better than normal ATPG but might not be as high a
BCE as n-detect with additional patterns could produce. In a design containing EDT circuitry,
the amount of detection will be dependent on the how aggressive the compression is. The more
aggressive (higher) compression, the lower the encoding capacity and the fewer test cube bits
can be specified per pattern. If a design is targeting 200x compression then the available test
cube bits might be mostly filled up for many of the patterns with values for the undetected fault

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detection. As a result, the additional EMD multiple detection might not be significantly higher
than BCE for the standard pattern set.

Standard multiple detect will have a cost of additional patterns but will also have a higher
multiple detection than EMD. How much difference between EMD and multiple detect will be
dependent on the particular design’s pattern set and the level of compression used1.

Enabling Multiple Detect for EMD

You can enable either EMD or multiple detect using the set_multiple_detection command.
Either approach is supported with stuck-at and transition patterns. You can use the following
arguments with the set_multiple_detection command:
• -Guaranteed_atpg_detection — Sets the multiple detect target for each fault. ATPG
will try to target each fault the specified number of times. ATPG does not guarantee that
the fault will be detected in a completely different path but randomly changes the way it
excites and propagates the fault. In addition, the random fill is different so values around
the target fault are likely to be randomly different than previous detections.
• -Desired_atpg_detections — Sets the EMD target. Users often set this target to 5 or a
value in that range.
• -Simulation_drop_limit — This is the accuracy of the BCE calculation. In general,
there is no reason to change this value from the default of 10. This means that the BCE
simulations stop once a fault is learned to be detected 10 times. A fault multiple detected
10 times will have a BCE and statistical chance of detecting a defect of 1 – 1/2e10 or
.99902. Which is only a 0.0009% inaccuracy which is slightly conservative but
insignificant.

Logic BIST
Logic BIST has a natural very high multiple detection. The faults that are detected with logic
BIST would often have multiple detection well above 10. This is in part due to the very large
number of patterns typically used for logic BIST. In addition, many of the hard to detect areas of
a circuit are made randomly testable and easier to produce high multiple detect coverage with
test logic inserted during logic BIST.

Multiple Detect and AU Faults

During multiple detect ATPG, the AU (ATPG_untestable) fault count changes only at the
beginning of each ATPG loop rather than during the loop. This is normal behavior when using
NCPs (named capture procedures) for ATPG. The tool updates AU faults after going through all
the NCPs at the end of the loop.

1. J. Geuzebroek, et al., “Embedded Multi-Detect ATPG and Its Effect on the Detection of Unmodeled
Defects”, Proceedings IEEE Int. Test Conference, 2007

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Fault Detection
Faults detection works by comparing the response of a known-good version of the circuit to that
of the actual circuit, for a given stimulus set. A fault exists if there is any difference in the
responses. You then repeat the process for each stimulus set.
Figure 2-13 shows the basic fault detection process.

Figure 2-13. Fault Detection Process

Apply Stimulus

Actual Good
Circuit Circuit

Compare
Response

N
Difference? Repeat for
Next Stimulus
Y
Fault
Detected

Path Sensitization and Fault Detection

One common fault detection approach is path sensitization. The path sensitization method,
which is used by the tool to detect stuck-at faults, starts at the fault site and tries to construct a
vector to propagate the fault effect to a primary output. When successful, the tools create a
stimulus set (a test pattern) to detect the fault. They attempt to do this for each fault in the
circuit's fault universe.
Figure 2-14 shows an example circuit for which path sensitization is appropriate.

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Figure 2-14. Path Sensitization Example

x1 s-a-0

x2
y1
y2
x3

Figure 2-14 has a stuck-at-0 on line y1 as the target fault. The x1, x2, and x3 signals are the
primary inputs, and y2 is the primary output. The path sensitization procedure for this example
follows:

1. Find an input value that sets the fault site to the opposite of the desired value. In this
case, the process needs to determine the input values necessary at x1 and/or x2 that set
y1 to a 1, since the target fault is s-a-0. Setting x1 (or x2) to a 0 properly sets y1 to a 1.
2. Select a path to propagate the response of the fault site to a primary output. In this case,
the fault response propagates to primary output y2.
3. Specify the input values (in addition to those specified in step 1) to enable detection at
the primary output. In this case, in order to detect the fault at y1, the x3 input must be set
to a 1.

Fault Classes
The tool categorizes faults into fault classes, based on how the faults were detected or why they
could not be detected. Each fault class has a unique name and two character class code. When
reporting faults, the tool uses either the class name or the class code to identify the fault class to
which the fault belongs.
Note
The tools may classify a fault in different categories, depending on the selected fault type.

Untestable (UT)
Untestable (UT) faults are faults for which no pattern can exist to either detect or possible-detect
them. Untestable faults cannot cause functional failures, so the tools exclude them when
calculating test coverage. Because the tools acquire some knowledge of faults prior to ATPG,
they classify certain unused, tied, or blocked faults before ATPG runs. When ATPG runs, it
immediately places these faults in the appropriate categories. However, redundant fault
detection requires further analysis.

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The following list discusses each of the untestable fault classes.

• Unused (UU)
The unused fault class includes all faults on circuitry unconnected to any circuit
observation point and faults on floating primary outputs. For information about UU fault
sub-classes, refer to Table 2-3 on page 62. Figure 2-15 shows the site of an unused fault.

Figure 2-15. Example of “Unused” Fault in Circuitry

Site of “Unused” Fault
D Q

Master
CLK Latch QB
s-a-1/s-a-0

• Tied (TI)
The tied fault class includes faults on gates where the point of the fault is tied to a value
identical to the fault stuck value. The tied circuitry could be due to:
o Tied signals
o AND and OR gates with complementary inputs
o Exclusive-OR gates with common inputs
o Line holds due to primary input pins held at a constant logic value during test by
CT0 or CT1 pin constraints you applied with the add_input_constraints command

Note
The tools do not use line holds set by the “add_input_constraints -C0” (or C1) command
to determine tied circuitry. C0 and C1 pin constraints (as distinct from CT0 and CT1
constraints) result in ATPG_untestable (AU) faults, not tied faults. For more information,
refer to the add_input_constraints command.

Figure 2-16 shows the site of a tied fault.

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Figure 2-16. Example of “Tied” Fault in Circuitry

Sites of “Tied” Faults

A B C D
s-a-0

GND

Because tied values propagate, the tied circuitry at A causes tied faults at A, B, C, and D.
• Blocked (BL)
The blocked fault class includes faults on circuitry for which tied logic blocks all paths
to an observable point. The tied circuitry could be due to:
o Tied signals
o AND and OR gates with complementary inputs
o Exclusive-OR gates with common inputs
o Line holds due to primary input pins held at a constant logic value during test by
CT0 or CT1 pin constraints you applied with the add_input_constraints command.

Note
The tools do not use line holds set by the “add_input_constraints -C0” (or C1) command
to determine tied circuitry. C0 and C1 pin constraints (as distinct from CT0 and CT1
constraints) result in ATPG_untestable (AU) faults, not blocked faults. For more
information, refer to the add_input_constraints command.

This class also includes faults on selector lines of multiplexers that have identical data
lines. Figure 2-17 shows the site of a blocked fault.

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Figure 2-17. Example of “Blocked” Fault in Circuitry

Site of “Blocked” Fault

s-a-0

GND

Note
Tied faults and blocked faults can be equivalent faults.

• Redundant (RE)
The redundant fault class includes faults the test generator considers undetectable. After
the test pattern generator exhausts all patterns, it performs a special analysis to verify
that the fault is undetectable under any conditions. Figure 2-18 shows the site of a
redundant fault.

Figure 2-18. Example of “Redundant” Fault in Circuitry

Site of “Redundant” Fault

VCC
E
A
D G
B
C s-a-1

F
GND

In this circuit, signal G always has the value of 1, no matter what the values of A, B, and
C. If D is stuck at 1, this fault is undetectable because the value of G can never change,
regardless of the value at D.

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Testable (TE)
Testable (TE) faults are all those faults that cannot be proven untestable. The testable fault
classes include:
• Detected (DT)
The detected fault class includes all faults that the ATPG process identifies as detected.
The detected fault class contains two groups:
o det_simulation (DS) - faults detected when the tool performs fault simulation.
o det_implication (DI) - faults detected when the tool performs learning analysis.
The det_implication group normally includes faults in the scan path circuitry, as well
as faults that propagate ungated to the shift clock input of scan cells. The scan chain
functional test, which detects a binary difference at an observation point, guarantees
detection of these faults. the tool also classifies scan enable stuck-in-system-mode
faults on the multiplexer select line of mux-DFFs as DI.
The tool provides the update_implication_detections command, which lets you
specify additional types of faults for this category. Refer to the
update_implication_detections command description in the Tessent Shell Reference
Manual.
For path delay testing, the detected fault class includes two other groups:
o det_robust (DR) - robust detected faults.
o det_functional (DF) - functionally detected faults.
For detailed information on the path delay groups, refer to “Path Delay Fault Detection”
on page 235.
• Posdet (PD)
The posdet, or possible-detected fault class includes all faults that fault simulation
identifies as possible-detected but not hard detected. A possible-detected fault results
from a good-machine simulation observing 0 or 1 and the faulty machine observing X.
A hard-detected fault results from binary (not X) differences between the good and
faulty machine simulations. The posdet class contains two groups:
o posdet_testable (PT) - potentially detectable posdet faults. PT faults result when the
tool cannot prove the 0/X or 1/X difference is the only possible outcome. A higher
abort limit may reduce the number of these faults.
o posdet_untestable (PU) - proven ATPG_untestable during pattern generation and
hard undetectable posdet faults. Typically, faults may be classified as PU during
ATPG or when the “compress_patterns -reset_au” command is used.
By default, the calculations give 50% credit for posdet faults. You can adjust the credit
percentage with the set_possible_credit command.

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• ATPG_untestable (AU)
The ATPG_untestable fault class includes all faults for which the test generator is
unable to find a pattern to create a test, and yet cannot prove the fault redundant.
Testable faults become ATPG_untestable faults because of constraints, or limitations,
placed on the ATPG tool (such as a pin constraint or an insufficient sequential depth).
These faults may be possible-detectable, or detectable, if you remove some constraint,
or change some limitation, on the test generator (such as removing a pin constraint or
changing the sequential depth). You cannot detect them by increasing the test generator
abort limit.
The tools place faults in the AU category based on the type of deterministic test
generation method used. That is, different test methods create different AU fault sets.
Likewise, the tool can create different AU fault sets even using the same test method.
Thus, if you switch test methods (that is, change the fault type) or tools, you should reset
the AU fault list using the reset_au_faults command.

Note
During multiple detect ATPG, the AU fault count changes only at the beginning of each
ATPG loop rather than during the loop. This is normal behavior when using NCPs
(named capture procedures) for ATPG. The tool updates AU faults after going through all
the NCPs at the end of the loop.

AU faults are categorized into several predefined fault sub-classes, as listed in Table 2-3
on page 62.
• Undetected (UD)
The undetected fault class includes undetected faults that cannot be proven untestable or
ATPG_untestable. The undetected class contains groups:
o uncontrolled (UC) - undetected faults, which during pattern simulation, never
achieve the value at the point of the fault required for fault detection—that is, they
are uncontrollable.
o unobserved (UO) - faults whose effects do not propagate to an observable point.
All testable faults prior to ATPG are put in the UC category. Faults that remain UC or
UO after ATPG are aborted, which means that a higher abort limit may reduce the
number of UC or UO faults.

Note
Uncontrolled and unobserved faults can be equivalent faults. If a fault is both
uncontrolled and unobserved, it is categorized as UC.

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Fault Class Hierarchy

Fault classes are hierarchical. The highest level, Full, includes all faults in the fault list. Within
Full, faults are classified into untestable and testable fault classes, and so on, as shown in the
following figure:.

Figure 2-19. Fault Class Hierarchy

1. Full (FU)
1.1 TEstable (TE)
a. DETEcted (DT)
i. DET_Simulation (DS)
ii. DET_Implication (DI)
iii. DET_Robust (DR)—Path Delay Testing Only
iv. DET_Functional (DF)—Path Delay Testing Only
b. POSDET (PD)
i. POSDET_Untestable (PU)
ii. POSDET_Testable (PT)
c. Atpg_untestable (AU)
d. UNDetected (UD)
i. UNControlled (UC)
ii. UNObserved (UO)
1.2 UNTestable (UT)
a. UNUsed (UU)
b. TIed (TI)
c. Blocked (BL)
d. Redundant (RE)

For any given level of the hierarchy, the tool assigns a fault to one—and only one—class. If the
tool can place a fault in more than one class of the same level, the tool places the fault in the
class that occurs first in the list of fault classes.

Fault Sub-classes
The DI, AU, UD, and UU fault classes are further categorized into fault sub-classes.
Table 2-3 lists the fault sub-classes. For more information about each AU and UD fault sub-
classes, click on the hyperlinks in the table.

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The DI and AU fault classes can also contain user-defined sub-classes, which you create by
grouping a set of faults into one sub-class and assigning a name to the group.

Table 2-3. Fault Sub-classes

Fault Fault Sub-class Code Description
Class
DI EDT_LOGIC EDT Faults in EDT logic detected by implication when
EDT Finder is On. Supported for stuck and
transition fault types.
• Includes faults implicitly tested when applying
the EDT patterns.
• Excludes redundant faults in EDT logic, faults in
bypass logic or other dual configurations.
• Excludes faults already identified as implicitly
tested.
AU BLACK_BOXES BB Fault untestable due to black box
EDT_BLOCKS EDT AU faults in EDT block
PIN_CONSTRAINTS PC Tied or blocked by input constraint
TIED_CELLS TC Tied or blocked by tied non-scan cell
CELL_CONSTRAINTS CC Tied or blocked by cell constraint
FALSE_PATHS FP Fault untestable due to false path
MULTICYCLE_PATHS MCP Fault untestable due to multicycle path
SEQUENTIAL_DEPTH SEQ Untestable due to insufficient sequential depth
UD ATPG_ABORT AAB Fault aborted by ATPG tool
UNSUCCESS UNS Fault undetected due to unsuccessful test
EDT_ABORT EAB Fault aborted due to insufficient EDT encoding
capacity
UU CONNECTED CON Fault in the logic cone but with no observation point
FLOATING FL Fault on the unconnected output of a cell

AU.BB — BLACK_BOXES
These are faults that are untestable due to a black box, which includes faults that need to be
propagated through a blackbox to reach an observation point, as well as faults whose control or
observation requires values from the output(s) of a blackbox. You can use the
report_black_boxes command to identify the black boxes in your design. Your only potential
option for resolving AU.BB faults is to create a model for each black box, although this may not
fix the entire problem.

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Understanding Test Types and Fault Models

AU.EDT — EDT_BLOCKS
These are faults inside an instance identified as an EDT instance using the set_edt_instances
command. If EDT Finder is turned on, most of those faults (regardless of the use of
set_edt_instances) are classified as DI.EDT faults.

To further reduce the number of AU.EDT faults, you must either use default naming for EDT
blocks, or you must use the add_edt_blocks command to explicitly name EDT blocks. Note that
the tool excludes AU.EDT faults from the relevant fault coverage by default because the chain
and IP test actually checks most of the EDT logic in addition to the scan chains, so other than
properly naming EDT blocks, you need not do anything more. For more information about
relevant coverage, refer to the description of the set_relevant_coverage command in the Tessent
Shell Reference Manual.

AU.PC — PIN_CONSTRAINTS
These are faults that are uncontrollable or that cannot be propagated to an observation point, in
the presence of a constraint value. That is, because the tool can’t toggle the pin, the tool cannot
test the fanout. The only possible solution is to evaluate whether you really need the input
constraint, and if not, to remove the constraint. To help troubleshoot, you can use the
“report_statistics -detailed_analysis” command to report specific pins.

AU.TC — TIED_CELLS
These are faults associated with cells that are always 0 (TIE0) or 1 (TIE1) or X (TIEX) during
capture. One example is test data registers that are loaded during test_setup and then
constrained so as to preserve that value during the rest of scan test. The only possible solution is
to verify that the tied state is really required, and if not, modify the test_setup procedure. To
help troubleshoot, you can use the “report_statistics -detailed_analysis” command to report
specific cells.

AU.CC — CELL_CONSTRAINTS
These are faults that are uncontrollable or that cannot be propagated to an observation point, in
the presence of a constraint value. That is, because the tool can’t toggle a state within the cell,
the tool cannot test the fanout or the faults along the input path that need to propagate through
the cell. The only possible solution is to evaluate whether you really need the cell constraint,
and if not, to remove the constraint. To help troubleshoot, you can use the “report_statistics
-detailed_analysis” command to report specific cells.

For more information about cell constraints and their affect on ATPG, refer to the
add_cell_constraints command description in the Tessent Shell Reference Manual.

AU.FP — FALSE_PATHS
These are faults that can be tested only through a false path. However, if there is any other path
possible that is not false in which the fault exists, then the fault remains in the fault list.

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Understanding Test Types and Fault Models

Therefore, many faults along false paths might not be classified as AU.FP. For example, assume
that the end of a long false path is connected to both a flop A and a flop B. If flop B isn't in a
false path, then the tool can test all the setup faults using flop B and can therefore classify none
as AU.FP.

In the case of hold-time false paths, the tool cannot classify them as AU.FP because a test is
almost always possible that doesn't cause a hold-time violation. For example, consider a circuit
with a hold-time false path that contains a flop A that feeds a buffer connected to flop B. If a
pattern places a 1 on flop A’s D input and a 0 on Q, then ATPG simulation would capture X in
flop B because the tool can’t be certain the capture cycle would appropriately capture 0 in flop
B. That is, the hold-time violation could cause flop A to update and then propagate to flop B
before the clock captures at flop B. However, if flop A’s D input is 0 and Q is 0, then the 0
would be properly captured regardless of the hold-time false path. This is why the tool cannot
remove the faults in this circuit from the fault list and define them as AU.FP.

Note that the tool excludes AU.FP faults from the relevant fault coverage by default. For more
information about relevant coverage, refer to the description of the set_relevant_coverage
command in the Tessent Shell Reference Manual.

AU.MCP — MULTICYCLE_PATHS
These are faults that can be tested only through a multicycle path, which means that the fault
can’t propagate between the launch and capture point within a single clock cycle. For
information about resolving multicycle path issues, refer to “Preventing Pattern Failures Due to
Timing Exception Paths” on page 228.

AU.SEQ — SEQUENTIAL_DEPTH
These are faults associated with non-scan cells that require multiple clock cycles to propagate to
an observe point. One possible solution is to increase the sequential depth using the
“set_pattern_type -sequential” command. Another solution is to ensure that you have defined all
clocks in the circuit using the add_clocks command.

UD.AAB — ATPG_ABORT
These are faults that are undetected because the tool reached its abort limit. You can gain
additional information about UD.AAB faults using the report_aborted_faults command. You
can raise the abort limit to increase coverage using the set_abort_limit command.

UD.UNS — UNSUCCESS
These are faults that are undetected for unknown reasons. There is nothing you can do about
faults in this sub-class.

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Understanding Test Types and Fault Models

UD.EAB — EDT_ABORT
These are faults that are undetected for one of these reasons:

• Your design’s chain:channel ratio is insufficient to detect the fault. One solution is to
run compression analysis on your design to determine the most aggressive chain:channel
ratio using the analyze_compression command, and then re-create the EDT logic in your
design. Another solution is to insert test points to improve the compressibility of faults.
• The tool cannot compress the test due to the scan cells being clustered (which is reported
at the end of ATPG). The solution is to insert test points to improve the compressibility
of faults.

Fault Reporting
When reporting faults, the tool identifies each fault by three ordered fields:
• fault value (0 for stuck-at-0 or “slow-to-rise” transition faults; 1 for stuck-at-1 or “slow-
to-fall” transition faults)
• two-character fault class code
• pin pathname of the fault site
If the tools report uncollapsed faults, they display faults of a collapsed fault group together, with
the representative fault first followed by the other members (with EQ fault codes). Use the
report_faults command to report faults.

Testability Calculations
Given the fault classes explained in the previous sections, the tool makes the following
calculations:
• Test Coverage
Test coverage, which is a measure of test quality, is the percentage of faults detected
from among all testable faults. Typically, this is the number of most concern when you
consider the testability of your design.
The tool calculates test coverage using the formula:
#DT + (#PD * posdet_credit)
——————————————————————————— x 100
#testable

In this formula, posdet_credit is the user-selectable detection credit (the default is 50%)
given to possible detected faults with the set_possible_credit command.
• Fault Coverage
Fault coverage consists of the percentage of faults detected from among all faults that
the test pattern set tests—treating untestable faults the same as undetected faults.

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Understanding Test Types and Fault Models

The tool calculates fault coverage using the formula:

#DT + (#PD * posdet_credit)
——————————————————————————— x 100
#full

• ATPG Effectiveness
ATPG effectiveness measures the ATPG tool’s ability to either create a test for a fault,
or prove that a test cannot be created for the fault under the restrictions placed on the
tool.
The tool calculates ATPG effectiveness using the formula:
#DT + #UT + #AU + #PU +(#PT *posdet_credit)
——————————————————————————————————————————— x 100
#full

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 Chapter 3
Understanding Common Tool Terminology
and Concepts

Now that you understand the basic ideas behind DFT, scan design, and ATPG, you can
concentrate on the Mentor Graphics DFT tools and how they operate. Tessent Scan and the
ATPG tools not only work toward a common goal (to improve test coverage), they also share
common terminology, internal processes, and other tool concepts, such as how to view the
design and the scan circuitry.
Figure 3-1 shows the range of subjects common to these tools.

Figure 3-1. Common Tool Concepts

Understand 1. Scan Terminology

DFT Basics
2. Scan Architectures
3. Test Procedure Files
Understand
Tool Concepts 4. Model Flattening
5. Learning Analysis
Understand 6. ATPG Design Rules Checking
Testability Issues

The following subsections discuss common terminology and concepts associated with scan
insertion and ATPG using Tessent Scan and ATPG tools.

Scan Terminology
This section introduces the scan terminology common to Tessent Scan and the ATPG tools.

Scan Cells
A scan cell is the fundamental, independently-accessible unit of scan circuitry, serving both as a
control and observation point for ATPG and fault simulation. You can think of a scan cell as a

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black box composed of an input, an output, and a procedure specifying how data gets from the
input to the output. The circuitry inside the black box is not important as long as the specified
procedure shifts data from input to output properly.
Because scan cell operation depends on an external procedure, scan cells are tightly linked to
the notion of test procedure files. “Test Procedure Files” on page 74 discusses test procedure
files in detail. Figure 3-2 illustrates the black box concept of a scan cell and its reliance on a test
procedure.

Figure 3-2. Generic Scan Cell

sc_in Scan sc_out

(scan data in) Cell (scan data out)

sc_in -> sc_out

specified by shift procedure
A scan cell contains at least one memory element (flip-flop or latch) that lies in the scan chain
path. The cell can also contain additional memory elements that may or may not be in the scan
chain path, as well as data inversion and gated logic between the memory elements.

Figure 3-3 gives one example of a scan cell implementation (for the mux-DFF scan type).

Figure 3-3. Generic Mux-DFF Scan Cell Implementation

MUX
data

sc_in sc_in D Q sc_out

clk DFF
sc_en sc_en
data Q'
clk

mux-DFF
data D1 Q sc_out
sc_in D2
sc_en EN
clk CK Q'

Each memory element may have a set and/or reset line in addition to clock-data ports. The
ATPG process controls the scan cell by placing either normal or inverted data into its memory
elements. The scan cell observation point is the memory element at the output of the scan cell.
Other memory elements can also be observable, but may require a procedure for propagating

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Scan Terminology

their values to the scan cell’s output. The following subsections describe the different memory
elements a scan cell may contain.

Master Element
The master element, the primary memory element of a scan cell, captures data directly from the
output of the previous scan cell. Each scan cell must contain one and only one master element.
For example, Figure 3-3 shows a mux-DFF scan cell, which contains only a master element.
However, scan cells can contain memory elements in addition to the master. Figures 3-4
through 3-7 illustrate examples of master elements in a variety of other scan cells.

The shift procedure in the test procedure file controls the master element. If the scan cell
contains no additional independently-clocked memory elements in the scan path, this procedure
also observes the master. If the scan cell contains additional memory elements, you may need to
define a separate observation procedure (called master_observe) for propagating the master
element’s value to the output of the scan cell.

Slave Element
The slave element, an independently-clocked scan cell memory element, resides in the scan
chain path. It cannot capture data directly from the previous scan cell. When used, it stores the
output of the scan cell. The shift procedure both controls and observes the slave element. The
value of the slave may be inverted relative to the master element.
Figure 3-4 shows a slave element within a scan cell.

Figure 3-4. LSSD Master/Slave Element Example

Bclk
Aclk Q
sc_in
sys_clk Latch Slave
data Element
Master Latch sc_out
Element

In the example of Figure 3-4, Aclk controls scan data input. Activating Aclk, with sys_clk
(which controls system data) held off, shifts scan data into the scan cell. Activating Bclk
propagates scan data to the output.

Shadow Element
The shadow element, either dependently- or independently-clocked, resides outside the scan
chain path. It can be inside or outside of a scan cell.

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Figure 3-5 gives an example of a scan cell with a dependently-clocked, non-observable shadow
element with a non-inverted value.

Figure 3-5. Dependently-clocked Mux-DFF/Shadow Element Example

Shadow
Master FF Element
Element
clk
data FF sc_out
MUX
sc_in S
sc_en

Figure 3-6 shows a similar example where the shadow element is independently-clocked.

Figure 3-6. Independently-clocked Mux-DFF/Shadow Element Example

sys_clk Shadow
Master FF Element
Element
clk
data FF sc_out
MUX
sc_in S
sc_en

You load a data value into the dependently-clocked shadow element with the shift procedure. If
the shadow element is independently clocked, you use a separate procedure called
shadow_control to load it. You can optionally make a shadow observable using the
shadow_observe procedure. A scan cell may contain multiple shadows but only one may be
observable, because the tools allow only one shadow_observe procedure. A shadow element’s
value may be the inverse of the master’s value.

The definition of a shadow element is based on the shadow having the same (or inverse) value
as the master element it shadows. A variety of interconnections of the master and shadow will
accomplish this. In Figure 3-5, the shadow’s data input is connected to the master’s data input,
and both FFs are triggered by the same clock edge. The definition would also be met if the
shadow’s data input was connected to the master’s output and the shadow was triggered on the
trailing edge, the master on the leading edge, of the same clock.

Copy Element
The copy element is a memory element that lies in the scan chain path and can contain the same
(or inverted) data as the associated master or slave element in the scan cell.

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Figure 3-7 gives an example of a copy element within a scan cell in which a master element
provides data to the copy.

Figure 3-7. Mux-DFF/Copy Element Example

clk
FF sc_out
Master
Element
data FF
MUX Copy
sc_in S Element
sc_en

The clock pulse that captures data into the copy’s associated scan cell element also captures data
into the copy. Data transfers from the associated scan cell element to the copy element in the
second half of the same clock cycle.

During the shift procedure, a copy contains the same data as that in its associated memory
element. However, during system data capture, some types of scan cells allow copy elements to
capture different data. When the copy’s value differs from its associated element, the copy
becomes the observation point of the scan cell. When the copy holds the same data as its
associated element, the associated element becomes the observation point.

Extra Element
The extra element is an additional, independently-clocked memory element of a scan cell. An
extra element is any element that lies in the scan chain path between the master and slave
elements. The shift procedure controls data capture into the extra elements. These elements are
not observable. Scan cells can contain multiple extras. Extras can contain inverted data with
respect to the master element.

Scan Chains
A scan chain is a set of serially linked scan cells. Each scan chain contains an external input pin
and an external output pin that provide access to the scan cells.
Figure 3-8 shows a scan chain, with scan input “sc_in” and scan output “sc_out”.

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Figure 3-8. Generic Scan Chain

sc_in 0
N-1 N-2 N-3
clk sc_out
sc_en
data

The scan chain length (N) is the number of scan cells within the scan chain. By convention, the
scan cell closest to the external output pin is number 0, its predecessor is number 1, and so on.
Because the numbering starts at 0, the number for the scan cell connected to the external input
pin is equal to the scan chain length minus one (N-1).

Scan Groups
A scan chain group is a set of scan chains that operate in parallel and share a common test
procedure file. The test procedure file defines how to access the scan cells in all of the scan
chains of the group. Normally, all of a circuit’s scan chains operate in parallel and are thus in a
single scan chain group.

Figure 3-9. Generic Scan Group

sci1 0
N-1 N-2 N-3
clk sco1
sc_en

sci2 0
N-1 N-2 N-3
sco2

You may have two clocks, A and B, each of which clocks different scan chains. You often can
clock, and therefore operate, the A and B chains concurrently, as shown in Figure 3-9.
However, if two chains share a single scan input pin, these chains cannot be operated in parallel.

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Scan Architectures

Regardless of operation, all defined scan chains in a circuit must be associated with a scan
group. A scan group is a concept used by Mentor Graphics DFT and ATPG tools.

Scan groups are a way to group scan chains based on operation. All scan chains in a group must
be able to operate in parallel, which is normal for scan chains in a circuit. However when scan
chains cannot operate in parallel, such as in the example above (sharing a common scan input
pin), the operation of each must be specified separately. This means the scan chains belong to
different scan groups.

Scan Clocks
Scan clocks are external pins capable of capturing values into scan cell elements. Scan clocks
include set and reset lines, as well as traditional clocks. Any pin defined as a clock can act as a
capture clock during ATPG.
Figure 3-10 shows a scan cell whose scan clock signals are shown in bold.

Figure 3-10. Scan Clocks Example

D1 CLR
D2 Q1
Q2
CK1 Q1'
CK2 Q2'

In addition to capturing data into scan cells, scan clocks, in their off state, ensure that the cells
hold their data. Design rule checks ensure that clocks perform both functions. A clock’s off-
state is the primary input value that results in a scan element’s clock input being at its inactive
state (for latches) or state prior to a capturing transition (for edge-triggered devices). In the case
of Figure 3-10, the off-state for the CLR signal is 1, and the off-states for CK1 and CK2 are
both 0.

Scan Architectures
Tessent Scan supports the insertion of mux-DFF (mux-scan) architecture. Additionally, Tessent
Scan supports all standard scan types, or combinations thereof, in designs containing pre-
existing scan circuitry.
Each scan style provides different benefits. Mux-DFF or clocked-scan are generally the best
choice for designs with edge-triggered flip-flops.

The following subsection details the mux-DFF architecture.

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Test Procedure Files

Mux-DFF
A mux-DFF cell contains a single D flip-flop with a multiplexed input line that allows selection
of either normal system data or scan data.
Figure 3-11 shows the replacement of an original design flip-flop with mux-DFF circuitry.

Figure 3-11. Mux-DFF Replacement

Original Replaced by
Flip-flop mux-DFF Scan Cell

data
D MUX
Q D Q sc_out
sc_in S
(Q)
CLK sc_en DFF
clk CLK

In normal operation (sc_en = 0), system data passes through the multiplexer to the D input of
the flip-flop, and then to the output Q. In scan mode (sc_en = 1), scan input data (sc_in) passes
to the flip-flop, and then to the scan output (sc_out).

Test Procedure Files

Test procedure files describe, for the ATPG tool, the scan circuitry operation within a design.
Test procedure files contain cycle-based procedures and timing definitions that tell the ATPG
tool how to operate the scan structures within a design. In order to use the scan circuitry in your
design, you must do the following:
• Define the scan circuitry for the tool.
• Create a test procedure file to describe the scan circuitry operation. Tessent Scan can
create test procedure files for you.
• Perform DRC process. This occurs when you exit from setup mode.
Once the scan circuitry operation passes DRC, the ATPG tool processes assume the scan
circuitry works properly.

If your design contains scan circuitry, the ATPG tool requires a test procedure file. You must
create one before running ATPG.

For more information about the test procedure file format, see “Test Procedure File” in the
Tessent Shell User’s Manual, which describes the syntax and rules of test procedure files, give

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Model Flattening

examples for the various types of scan architectures, and outline the checking that determines
whether the circuitry is operating correctly.

Model Flattening
To work properly, the ATPG tool and Tessent Scan must use their own internal representations
of the design. The tools create these internal design models by flattening the model and
replacing the design cells in the netlist (described in the library) with their own primitives. The
tools flatten the model when you initially attempt to exit setup mode, just prior to design rules
checking. The ATPG tool also provides the create_flat_model command, which allows
flattening of the design model while still in setup mode.
If a flattened model already exists when you exit setup mode, the tools will only reflatten the
model if you have since issued commands that would affect the internal representation of the
design. For example, adding or deleting primary inputs, tying signals, and changing the internal
faulting strategy are changes that affect the design model. With these types of changes, the tool
must re-create or re-flatten the design model. If the model is undisturbed, the tool keeps the
original flattened model and does not attempt to reflatten.

For a list of the specific Tessent Scan commands that cause flattening, refer to the
set_system_mode description in the Tessent Shell Reference Manual.

• create_flat_model — Creates a primitive gate simulation representation of the design.

• report_flattener_rules — Displays either a summary of all the flattening rule violations
or the data for a specific violation.
• set_flattener_rule_handling — Specifies how the tool handles flattening violations.

Understanding Design Object Naming

Tessent Scan and the ATPG tool use special terminology to describe different objects in the
design hierarchy. The following list describes the most common:
• Instance — A specific occurrence of a library model or functional block in the design.
• Hierarchical instance — An instance that contains additional instances and/or gates
underneath it.
• Module — A Verilog functional block (module) that can be repeated multiple times.
Each occurrence of the module is a hierarchical instance.

The Flattening Process

The flattened model contains only simulation primitives and connectivity, which makes it an
optimal representation for the processes of fault simulation and ATPG.

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Model Flattening

Figure 3-12 shows an example of circuitry containing an AND-OR-Invert cell and an AND
gate, before flattening.

Figure 3-12. Design Before Flattening

/Top
A
AOI1 AND1
B A Z
C Y B
D AOI
E

Figure 3-13 shows this same design once it has been flattened.

Figure 3-13. Design After Flattening

Pin Pathname
/Top/AOI1/B

/Top/AOI1
B
/Top/AND1
C /Top/AOI1 A
Z
Y B
/Top/AOI1
D
E

Unnamed Pin Pathname

Pins /Top/AND1/B

After flattening, only naming preserves the design hierarchy; that is, the flattened netlist
maintains the hierarchy through instance naming. Figures 3-12 and 3-13 show this hierarchy
preservation. /Top is the name of the hierarchy’s top level. The simulation primitives (two AND
gates and a NOR gate) represent the flattened instance AOI1 within /Top. Each of these
flattened gates retains the original design hierarchy in its naming—in this case, /Top/AOI1.

The tools identify pins from the original instances by hierarchical pathnames as well. For
example, /Top/AOI1/B in the flattened design specifies input pin B of instance AOI1. This
naming distinguishes it from input pin B of instance AND1, which has the pathname
/Top/AND1/B. By default, pins introduced by the flattening process remain unnamed and are not
valid fault sites. If you request gate reporting on one of the flattened gates, the NOR gate for

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example, you will see a system-defined pin name shown in quotes. If you want internal faulting
in your library cells, you must specify internal pin names within the library model. The
flattening process then retains these pin names.

You should be aware that in some cases, the design flattening process can appear to introduce
new gates into the design. For example, flattening decomposes a DFF gate into a DFF
simulation primitive, the Q and Q’ outputs require buffer and inverter gates, respectively. If
your design wires together multiple drivers, flattening would add wire gates or bus gates.
Bidirectional pins are another special case that requires additional gates in the flattened
representation.

Simulation Primitives of the Flattened Model

Tessent Scan and the ATPG tool select from a number of simulation primitives when they
create the flattened circuitry. The simulation primitives are multiple-input (zero to four), single-
output gates, except for the RAM, ROM, LA, and DFF primitives. The following list describes
these simulation primitives:
• PI, PO - primary inputs are gates with no inputs and a single output, while primary
outputs are gates with a single input and no fanout.
• BUF - a single-input gate that passes the values 0, 1, or X through to the output.
• FB_BUF - a single-input gate, similar to the BUF gate, that provides a one iteration
delay in the data evaluation phase of a simulation. The tools use the FB_BUF gate to
break some combinational loops and provide more optimistic behavior than when TIEX
gates are used.

Note
There can be one or more loops in a feedback path. In analysis mode, you can display the
loops with the report_loops command. In setup mode, use report_feedback_paths.

The default loop handling is simulation-based, with the tools using the FB_BUF to
break the combinational loops. In setup mode, you can change the default with the
set_loop_handling command. Be aware that changes to loop handling will have an
impact during the flattening process.
• ZVAL - a single-input gate that acts as a buffer unless Z is the input value. When a Z is
the input value, the output is an X. You can modify this behavior with the
set_z_handling command.
• INV - a single-input gate whose output value is the opposite of the input value. The INV
gate cannot accept a Z input value.
• AND, NAND - multiple-input gates (two to four) that act as standard AND and NAND
gates.
• OR, NOR - multiple-input (two to four) gates that act as standard OR and NOR gates.

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• XOR, XNOR - 2-input gates that act as XOR and XNOR gates, except that when either
input is an X, the output is an X.
• MUX - a 2x1 mux gate whose pins are order dependent, as shown in Figure 3-14.

Figure 3-14. 2x1 MUX Example

sel
d1 MUX out
d2

The sel input is the first defined pin, followed by the first data input and then the second
data input. When sel=0, the output is d1. When sel=1, the output is d2.
• LA, DFF - state elements, whose order dependent inputs include set, reset, and
clock/data pairs, as shown in Figure 3-15.

Figure 3-15. LA, DFF Example

set
reset
C1 out
D1
C2
D2
Set and reset lines are always level sensitive, active high signals. DFF clock ports are
edge-triggered while LA clock ports are level sensitive. When set=1, out=1. When
reset=1, out=0. When a clock is active (for example C1=1), the output reflects its
associated data line value (D1). If multiple clocks are active and the data they are trying
to place on the output differs, the output becomes an X.
• TLA, STLA, STFF - special types of learned gates that act as, and pass the design rule
checks for, transparent latch, sequential transparent latch, or sequential transparent flip-
flop. These gates propagate values without holding state.
• TIE0, TIE1, TIEX, TIEZ - zero-input, single-output gates that represent the effect of a
signal tied to ground or power, or a pin or state element constrained to a specific value
(0,1,X, or Z). The rules checker may also determine that state elements exhibit tied
behavior and replace them with the appropriate tie gates.
• TSD, TSH - a 2-input gate that acts as a tri-state™ driver, as shown in Figure 3-16.

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Model Flattening

Figure 3-16. TSD, TSH Example

en
TSD out
d

When en=1, out=d. When en=0, out=Z. The data line, d, cannot be a Z. The ATPG tool
uses the TSD gate for the same purpose.
• SW, NMOS - a 2-input gate that acts like a tri-state driver but can also propagate a Z
from input to output. The ATPG tool uses the SW gate uses the NMOS gate for the same
purpose.
• BUS - a multiple-input (up to four) gate whose drivers must include at least one TSD or
SW gate. If you bus more than four tri-state drivers together, the tool creates cascaded
BUS gates. The last bus gate in the cascade is considered the dominant bus gate.
• WIRE - a multiple-input gate that differs from a bus in that none of its drivers are tri-
statable.
• PBUS, SWBUS - a 2-input pull bus gate, for use when you combine strong bus and
weak bus signals together, as shown in Figure 3-17.

Figure 3-17. PBUS, SWBUS Example

(strong)
BUS
(weak) PBUS ZVAL

TIE0

The strong value always goes to the output, unless the value is a Z, in which case the
weak value propagates to the output. These gates model pull-up and pull-down resistors.
The ATPG tool uses the PBUS gate.
• ZHOLD - a single-input buskeeper gate (see page 87 for more information on
buskeepers) associated with a tri-state network that exhibits sequential behavior. If the
input is a binary value, the gate acts as a buffer. If the input value is a Z, the output
depends on the gate’s hold capability. There are three ZHOLD gate types, each with a
different hold capability:
o ZHOLD0 - When the input is a Z, the output is a 0 if its previous state was 0. If its
previous state was a 1, the output is a Z.
o ZHOLD1 - When the input is a Z, the output is a 1 if its previous state was a 1. If its
previous state was a 0, the output is a Z.

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Understanding Common Tool Terminology and Concepts
Learning Analysis

o ZHOLD0,1 - When the input is a Z, the output is a 0 if its previous state was a 0, or
the output is a 1 if its previous state was a 1.
In all three cases, if the previous value is unknown, the output is X.
• RAM, ROM- multiple-input gates that model the effects of RAM and ROM in the
circuit. RAM and ROM differ from other gates in that they have multiple outputs.
• OUT - gates that convert the outputs of multiple output gates (such as RAM and ROM
simulation gates) to a single output.

Learning Analysis
After design flattening, the ATPG tool performs extensive analysis on the design to learn
behavior that may be useful for intelligent decision making in later processes, such as fault
simulation and ATPG. You have the ability to turn learning analysis off, which may be
desirable if you do not want to perform ATPG during the session.
For more information on turning learning analysis off, refer to the set_static_learning
description in the Tessent Shell Reference Manual.

The ATPG tools perform static learning only once—after flattening. Because pin and ATPG
constraints can change the behavior of the design, static learning does not consider these
constraints. Static learning involves gate-by-gate local simulation to determine information
about the design. The following subsections describe the types of analysis performed during
static learning.

Equivalence Relationships
During this analysis, simulation traces back from the inputs of a multiple-input gate through a
limited number of gates to identify points in the circuit that always have the same values in the
good machine.
Figure 3-18 shows an example of two of these equivalence points within some circuitry.

Figure 3-18. Equivalence Relationship Example

Equivalence
Points

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Learning Analysis

Logic Behavior
During logic behavior analysis, simulation determines a circuit’s functional behavior.
For example, Figure 3-19 shows some circuitry that, according to the analysis, acts as an
inverter.

Figure 3-19. Example of Learned Logic Behavior

1
0 1

1 Has Complement Here

Value Here
During gate function learning, the tool identifies the circuitry that acts as gate types TIE (tied 0,
1, or X values), BUF (buffer), INV (inverter), XOR (2-input exclusive OR), MUX (single select
line, 2-data-line MUX gate), AND (2-input AND), and OR (2-input OR). For AND and OR
function checking, the tool checks for busses acting as 2-input AND or OR gates. The tool then
reports the learned logic gate function information with the messages:

Learned gate functions: #<gatetype>=<number> ...

Learned tied gates: #<gatetype>=<number> ...

If the analysis process yields no information for a particular category, it does not issue the
corresponding message.

Implied Relationships
This type of analysis consists of contrapositive relation learning, or learning implications, to
determine that one value implies another. This learning analysis simulates nearly every gate in
the design, attempting to learn every relationship possible.
Figure 3-20 shows the implied learning the analysis derives from a piece of circuitry.

Figure 3-20. Example of Implied Relationship Learning

A 1
B
1 1
1

“1” here always means a “1” here

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Learning Analysis

The analysis process can derive a very powerful relationship from this circuitry. If the value of
gate A=1 implies that the value of gate B=1, then B=0 implies A=0. This type of learning
establishes circuit dependencies due to reconvergent fanout and buses, which are the main
obstacles for ATPG. Thus, implied relationship learning significantly reduces the number of
bad ATPG decisions.

Forbidden Relationships
During forbidden relationship analysis, which is restricted to bus gates, simulation determines
that one gate cannot be at a certain value if another gate is at a certain value.
Figure 3-21 shows an example of such behavior.

Figure 3-21. Forbidden Relationship Example

0
1 TSD
TSD Tie 1
Tie 1 1 Z
1 0
BUS BUS
0 Z 1 0
TSD TSD
Tie 0 Tie 0
A 1 at each output would be forbidden

Dominance Relationships
During dominance relationship analysis, simulation determines which gates are dominators. If
all the fanouts of a gate go to a second gate, the second gate is the dominator of the first.
Figure 3-22 shows an example of this relationship.

Figure 3-22. Dominance Relationship Example

Gate B is
B Dominator
A of Gate A

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ATPG Design Rules Checking

ATPG Design Rules Checking

Tessent Scan and the ATPG tool perform design rules checking (DRC) after design flattening.
While not all of the tools perform the exact same checks, design rules checking generally
consists of the following processes, done in the order shown:
1. General Rules Checking
2. Procedure Rules Checking
3. Bus Mutual Exclusivity Analysis
4. Scan Chain Tracing
5. Shadow Latch Identification
6. Data Rules Checking
7. Transparent Latch Identification
8. Clock Rules Checking
9. RAM Rules Checking
10. Bus Keeper Analysis
11. Extra Rules Checking
12. Scanability Rules Checking
13. Constrained/Forbidden/Block Value Calculations

General Rules Checking

General rules checking searches for very-high-level problems in the information defined for the
design. For example, it checks to ensure the scan circuitry, clock, and RAM definitions all make
sense. General rules violations are errors and you cannot change their handling.
The “General Rules” section in the Tessent Shell Reference Manual describes the general rules
in detail.

Procedure Rules Checking

Procedure rules checking examines the test procedure file. These checks look for parsing or
syntax errors and ensure adherence to each procedure’s rules. Procedure rules violations are
errors and you cannot change their handling.
The “Procedure Rules” section in the Tessent Shell Reference Manual describes the procedure
rules in detail.

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Bus Mutual Exclusivity Analysis

Buses in circuitry can cause the following problems for ATPG:
• Bus contention during ATPG
• Testing stuck-at faults on tri-state drivers of buses.
This section addresses the first concern, that ATPG must place buses in a non-contending state.
For information on how to handle testing of tri-state devices, see “Tri-State Devices” on
page 103.

Figure 3-23 shows a bus system that can have contention.

Figure 3-23. Bus Contention Example

1
TSD 0
0
1 BUS
1
1 TSD

Many designs contain buses, but good design practices usually prevent bus contention. As a
check, the learning analysis for buses determines if a contention condition can occur within the
given circuitry. Once learning determines that contention cannot occur, none of the later
processes, such as ATPG, ever check for the condition.

Buses in a Z-state network can be classified as dominant or non-dominant and strong or weak.
Weak buses and pull buses are allowed to have contention. Thus the process only analyzes
strong, dominant buses, examining all drivers of these gates and performing full ATPG analysis
of all combinations of two drivers being forced to opposite values. Figure 3-24 demonstrates
this process on a simple bus system.

Figure 3-24. Bus Contention Analysis

E1
D1 TSD
BUS
E2
TSD
D2
Analysis tries:
E1=1, E2=1, D1=0, D2=1
E1=1, E2=1, D1=1, D2=0
If ATPG analysis determines that either of the two conditions shown can be met, the bus fails
bus mutual-exclusivity checking. Likewise, if the analysis proves the condition is never

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ATPG Design Rules Checking

possible, the bus passes these checks. A third possibility is that the analysis aborts before it
completes trying all of the possibilities. In this circuit, there are only two drivers, so ATPG
analysis need try only two combinations. However, as the number of drivers increases, the
ATPG analysis effort grows significantly.

You should resolve bus mutual-exclusivity before ATPG. Extra rules E4, E7, E9, E10, E11,
E12, and E13 perform bus analysis and contention checking. Refer to “Extra Rules” in the
Tessent Shell Reference Manual for more information on these bus checking rules.

Scan Chain Tracing

The purpose of scan chain tracing is for the tool to identify the scan cells in the chain and
determine how to use them for control and observe points. Using the information from the test
procedure file (which has already been checked for general errors during the procedure rules
checks) and the defined scan data, the tool identifies the scan cells in each defined chain and
simulates the operation specified by the load_unload procedure to ensure proper operation.
Scan chain tracing takes place during the trace rules checks, which trace back through the
sensitized path from output to input. Successful scan chain tracing ensures that the tools can use
the cells in the chain as control and observe points during ATPG.
Trace rules violations are either errors or warnings, and for most rules you cannot change the
handling. The “Scan Chain Trace Rules” section in the Tessent Shell Reference Manual
describes the trace rules in detail.

Shadow Latch Identification

Shadows are state elements that contain the same data as an associated scan cell element, but do
not lie in the scan chain path. So while these elements are technically non-scan elements, their
identification facilitates the ATPG process. This is because if a shadow elements’s content is
the same as the associated element’s content, you always know the shadow’s state at that point.
Thus, a shadow can be used as a control point in the circuit.
If the circuitry allows, you can also make a shadow an observation point by writing a
shadow_observe test procedure. The section entitled “Shadow Element” on page 69 discusses
shadows in more detail.

The DRC process identifies shadow latches under the following conditions:

1. The element must not be part of an already identified scan cell.

2. Plus any one of the following:
• At the time the clock to the shadow latch is active, there must be a single sensitized
path from the data input of the shadow latch up to the output of a scan latch.
Additionally the final shift pulse must occur at the scan latch no later than the clock
pulse to the shadow latch (strictly before, if the shadow is edge triggered).

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• The shadow latch is loaded before the final shift pulse to the scan latch is identified
by tracing back the data input of the shadow latch. In this case, the shadow will be a
shadow of the next scan cell closer to scan out than the scan cell identified by
tracing. If there is no scan cell close to scan out, then the sequential element is not a
valid shadow.
• The shadow latch is sensitized to a scan chain input pin during the last shift cycle. In
this case, the shadow latch will be a shadow of the scan cell closest to scan in.

Data Rules Checking

Data rules checking ensures the proper transfer of data within the scan chain. Data rules
violations are either errors or warnings, however, you can change the handling.
The “Scan Cell Data Rules” section in the Tessent Shell Reference Manual describes the data
rules in detail.

Transparent Latch Identification

Transparent latches are latches that can propagate values but do not hold state. A basic scan
pattern contains the following events:
1. Load scan chain
Latch must behave as 2. Force values on primary inputs
transparent between 3. Measure values on primary outputs
events 2 and 3
4. Pulse the capture clock
5. Unload the scan chain
Between the PI force and PO measure, the tool constrains all pins and sets all clocks off. Thus,
for a latch to qualify as transparent, the analysis must determine that it can be turned on when
clocks are off and pins are constrained. TLA simulation gates, which rank as combinational,
represent transparent latches.

Clock Rules Checking

After the scan chain trace, clock rules checking is the next most important analysis. Clock rules
checks ensure data stability and capturability in the chain. Clock rules violations are either
errors or warnings, however, you can change the handling.
The “Clock Rules” section in the Tessent Shell Reference Manual describes the clock rules in
detail.

RAM Rules Checking

RAM rules checking ensures consistency with the defined RAM information and the chosen
testing mode. RAM rules violations are all warnings, however, you can change their handling.

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The “RAM Rules” section in the Tessent Shell Reference Manual describes the RAM rules in
detail.

Bus Keeper Analysis

Bus keepers model the ability of an undriven bus to retain its previous binary state. You specify
bus keeper modeling with a bus_keeper attribute in the model definition. When you use the
bus_keeper attribute, the tool uses a ZHOLD gate to model the bus keeper behavior during
design flattening.
In this situation, the design’s simulation model becomes that shown in Figure 3-25:

Figure 3-25. Simulation Model with Bus Keeper

Tri-State
Device

BUS ZHOLD

Tri-State
Device

Rules checking determines the values of ZHOLD gates when clocks are off, pin constraints are
set, and the gates are connected to clock, write, and read lines. ZHOLD gates connected to
clock, write, and read lines do not retain values unless the clock off-states and constrained pins
result in binary values.

During rules checking, if a design contains ZHOLD gates, messages indicate when ZHOLD
checking begins, the number and type of ZHOLD gates, the number of ZHOLD gates connected
to clock, write, and read lines, and the number of ZHOLD gates set to a binary value during the
clock off-state condition.

Note
Only the ATPG tool requires this type of analysis, because of the way it “flattens” or
simulates a number of events in a single operation.

For information on the bus_keeper model attribute, refer to “Attributes” in the Tessent Cell
Library Manual.

Extra Rules Checking

Excluding rule E10, which performs bus mutual-exclusivity checking, most extra rules checks
do not have an impact on Tessent Scan and the ATPG tool processes. However, they may be

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ATPG Design Rules Checking

useful for enforcing certain design rules. By default, most extra rules violations are set to
ignore, which means they are not even checked during DRC. However, you may change the
handling.
For more information, refer to “Extra Rules” in the Tessent Shell Reference Manual for more
information.

Scanability Rules Checking

Each design contains a certain number of memory elements. Tessent Scan examines all these
elements and performs scannability checking on them, which consists mainly of the audits
performed by rules S1, S2, S3, and S4. Scannability rules are all warnings, and you cannot
change their handling, except for the S3 rule.
For more information, refer to “Scanability Rules (S Rules)” in the Tessent Shell Reference
Manual.

Constrained/Forbidden/Block Value Calculations

This analysis determines constrained, forbidden, and blocked circuitry. The checking process
simulates forward from the point of the constrained, forbidden, or blocked circuitry to
determine its effects on other circuitry. This information facilitates downstream processes, such
as ATPG.
Figure 3-26 gives an example of a tie value gate that constrains some surrounding circuitry.

Figure 3-26. Constrained Values in Circuitry

0
PI 0
(TIE0)
Resulting Constrained
Constrained Value Value
Figure 3-27 gives an example of a tied gate, and the resulting forbidden values of the
surrounding circuitry.

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ATPG Design Rules Checking

Figure 3-27. Forbidden Values in Circuitry

1
0,1
TIEX

Resulting Forbidden
Forbidden Values
Value
Figure 3-28 gives an example of a tied gate that blocks fault effects in the surrounding circuitry.

Figure 3-28. Blocked Values in Circuitry

Fault effect Fault Effect Blocked

from circuitry
X
X
TIEX

Output Always X
Tied Value

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 Chapter 4
Understanding Testability Issues

Testability naturally varies from design to design. Some features and design styles make a
design difficult, if not impossible, to test, while others enhance a design's testability.
Figure 4-1shows the testability issues this section discusses.

Figure 4-1. Testability Issues

Understand
Tool Concepts
1. Synchronous Circuitry
2. Asynchronous Circuitry
Understand
Testability Issues 3. Scannability Checking
4. Support for Special Testability Cases
Insert/Verify
BS Circuitry

The following subsections discuss these design features and describe their effect on the design's
testability.

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Synchronous Circuitry

Synchronous Circuitry
Using synchronous design practices, you can help ensure that your design will be both testable
and manufacturable. In the past, designers used asynchronous design techniques with TTL and
small PAL-based circuits. Today, however, designers can no longer use those techniques
because the organization of most gate arrays and FPGAs necessitates the use of synchronous
logic in their design.
A synchronous circuit operates properly and predictably in all modes of operation, from static
DC up to the maximum clock rate. Inputs to the circuit do not cause the circuit to assume
unknown states. And regardless of the relationship between the clock and input signals, the
circuit avoids improper operation.

Truly synchronous designs are inherently testable designs. You can implement many scan
strategies, and run the ATPG process with greater success, if you use synchronous design
techniques. Moreover, you can create most designs following these practices with no loss of
speed or functionality.

Synchronous Design Techniques

Your design’s level of synchronicity depends on how closely you observe the following
techniques:
• The system has a minimum number of clocks—optimally only one.
• You register all design inputs and account for metastability. That is, you should treat the
metastability time as another delay in the path. If the propagation delay plus the
metastability time is less than the clock period, the system is synchronous. If it is greater
than or equal to the clock period, you need to add an extra flip-flop to ensure the proper
data enters the circuit.
• No combinational logic drives the set, reset, or clock inputs of the flip-flops.
• No asynchronous signals set or reset the flip-flops.
• Buffers or other delay elements do not delay clock signals.
• Do not use logic to delay signals.
• Do not assume logic delays are longer than routing delays.
If you adhere to these design rules, you are much more likely to produce a design that is
manufacturable, testable, and operates properly over a wide range of temperature, voltage, and
other circuit parameters.

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Asynchronous Circuitry

Asynchronous Circuitry
A small percentage of designs need some asynchronous circuitry due to the nature of the
system. Because asynchronous circuitry is often very difficult to test, you should place the
asynchronous portions of your design in one block and isolate it from the rest of the circuitry. In
this way, you can still utilize DFT techniques on the synchronous portions of your design.

Scannability Checking
Tessent Scan performs the scannability checking process on a design’s sequential elements. For
the tool to insert scan circuitry into a design, it must replace existing sequential elements with
their scannable equivalents. Before beginning substitution, the original sequential elements in
the design must pass scannability checks; that is, the tool determines if it can convert sequential
elements to scan elements without additional circuit modifications.
Scannable sequential elements pass the following checks:

1. When all clocks are off, all clock inputs (including set and reset inputs) of the sequential
element must be in their inactive state (initial state of a capturing transition). This
prevents disturbance of the scan chain data before application of the test pattern at the
primary input. If the sequential element does not pass this check, its scan values could
become unstable when the test tool applies primary input values. This checking is a
modification of rule C1. For more information on this rule, refer to “C1” in the Tessent
Shell Reference Manual.
2. Each clock input (not including set and reset inputs) of the sequential element must be
capable of capturing data when a single clock primary input goes active while all other
clocks are inactive. This rule ensures that this particular storage element can capture
system data. If the sequential element does not meet this rule, some loss of test coverage
could result. This checking is a modification of rule C7. For more information on this
rule, refer to “C7” in the Tessent Shell Reference Manual.
When a sequential element passes these checks, it becomes a scan candidate, meaning that
Tessent Scan can insert its scan equivalent into the scan chain. However, even if the element
fails to pass one of these checks, it may still be possible to convert the element to scan. In many
cases, you can add additional logic, called test logic, to the design to remedy the situation. For
more information on test logic, refer to “Enabling Test Logic Insertion” on page 124.

Note
If TIE0 and TIE1 nonscan cells are scannable, they are considered for scan. However, if
these cells are used to hold off sets and resets of other cells so that another cell can be
scannable, you must use the add_nonscan_instances command to make them nonscan.

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Support for Special Testability Cases

Scannability Checking of Latches

By default, Tessent Scan performs scannability checking on all flip-flops and latches.
When latches do not pass scannability checks, Tessent Scan considers them non-scan elements
and then classifies them into one of the categories explained in “Non-Scan Cell Handling” on
page 104. However, if you want Tessent Scan to perform transparency checking on the non-
scan latches, you must turn off checking of rule D6 prior to scannability checking. For more
information on this rule, refer to “D6” in the Tessent Shell Reference Manual.

Support for Special Testability Cases

The following subsections explain certain design features that can pose design testability
problems and describe how Mentor Graphics DFT tools handle these situations.

Feedback Loops
Designs containing loop circuitry have inherent testability problems. A structural loop exists
when a design contains a portion of circuitry whose output, in some manner, feeds back to one
of its inputs. A structural combinational loop occurs when the feedback loop, the path from the
output back to the input, passes through only combinational logic. A structural sequential loop
occurs when the feedback path passes through one or more sequential elements.
The ATPG tool and Tessent Scan all provide some common loop analysis and handling.
However, loop treatment can vary depending on the tool. The following subsections discuss the
treatment of structural combinational and structural sequential loops.

Structural Combinational Loops and Loop-Cutting

Methods
Structural combinational loops are hardwired feedback paths in combinational circuits that
make that circuit difficult to test.
Figure 4-2 shows an example of a structural combinational loop. Notice that the A=1, B=0, C=1
state causes unknown (oscillatory) behavior, which poses a testability problem.

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Figure 4-2. Structural Combinational Loop Example

ABC P
0 0 0 0
0 0 1 1
A 0 1 0 0
0 1 1 0
B 1 0 0 0
C P 1 0 1 X
1 1 0 0
1 1 1 0
The flattening process, which each tool runs as it attempts to exit setup mode, identifies and
cuts, or breaks, all structural combinational loops. The tools classify and cut each loop using the
appropriate methods for each category.

The following list presents the loop classifications, as well as the loop-cutting methods
established for each. The order of the categories presented indicates the least to most pessimistic
loop cutting solutions.

1. Constant value
This loop cutting method involves those loops blocked by tied logic or pin constraints.
After the initial loop identification, the tools simulate TIE0/TIE1 gates and constrained
inputs. Loops containing constant value gates as a result of this simulation, fall into this
category.
Figure 4-3 shows a loop with a constrained primary input value that blocks the loop’s
feedback effects.

Figure 4-3. Loop Naturally-Blocked by Constant Value

Combinational
Logic

C0 PI 0
0

These types of loops lend themselves to the simplest and least pessimistic breaking
procedures. For this class of loops, the tool inserts a TIE-X gate at a non-constrained
input (which lies in the feedback path) of the constant value gate, as Figure 4-4 shows.

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Figure 4-4. Cutting Constant Value Loops

Combinational
Logic
TIEX

C0 PI 0
0

This loop cutting technique yields good circuit simulation that always matches the actual
circuit behavior, and thus, the tools employ this technique whenever possible. The tools
can use this loop cutting method for blocked loops containing AND, OR, NAND, and
NOR gates, as well as MUX gates with constrained select lines and tri-state drivers with
constrained enable lines.
2. Single gate with “multiple fanout”
This loop cutting method involves loops containing only a single gate with multiple
fanout.
Figure 4-2 on page 95 shows the circuitry and truth table for a single multiple-fanout
loop. For this class of loops, the tool cuts the loop by inserting a TIE-X gate at one of the
fanouts of this “multiple fanout gate” that lie in the loop path, as Figure 4-5 shows.

Figure 4-5. Cutting Single Multiple-Fanout Loops

ABC P
TIEX
0 0 0 0
0 0 1 1
0 1 0 0
A 0 1 1 0
1 0 0 0
B 1 0 1 X
C P 1 1 0 0
1 1 1 0

3. Gate duplication for multiple gate with multiple fanout

This method involves duplicating some of the loop logic—when it proves practical to do
so. The tools use this method when it can reduce the simulation pessimism caused by
breaking combinational loops with TIE-X gates. The process analyzes a loop, picks a
connection point, duplicates the logic (inserting a TIE-X gate into the copy), and
connects the original circuitry to the copy at the connection point.
Figure 4-6 shows a simple loop that the tools would target for gate duplication.

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Figure 4-6. Loop Candidate for Duplication

P
Q
A
R AB PQR
B
0 0 0 0 1
0 1 XX X
1 0 0 1 0
1 1 0 1 0

Figure 4-7 shows how TIE-X insertion would add some pessimism to the simulation at
output P.

Figure 4-7. TIE-X Insertion Simulation Pessimism

X
P
1
Q
A 1 1 0
R AB PQR
B 1 X 0 0 0 0 1
X 0 1 XX X
0 1 0 0 1 0
TIEX
X 1 1 X 1 0
Ambiguity added
by TIE-X Insertion

The loop breaking technique proves beneficial in many cases. Figure 4-8 provides a
more accurate simulation model than the direct TIE-X insertion approach.

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Figure 4-8. Cutting Loops by Gate Duplication

A 1 1 0
R AB PQR
B 1 X 0 0 0 0 1
0 1 XX X
X 1 0 0 1 0
1 1 0 1 0
TIEX 1 1
Q
1 Ambiguity
0 0 removed by
P duplication
0
technique

However, it also has some drawbacks. While less pessimistic than the other approaches
(except breaking constant value loops), the gate duplication process can still introduce
some pessimism into the simulation model.
Additionally, this technique can prove costly in terms of gate count as the loop size
increases. Also, the tools cannot use this method on complex or coupled loops—those
loops that connect with other loops (because gate duplication may create loops as well).
4. Coupling loops
The tools use this technique to break loops when two or more loops share a common
gate. This method involves inserting a TIE-X gate at the input of one of the components
within a loop. The process selects the cut point carefully to ensure the TIE-X gate cuts as
many of the coupled loops as possible.
For example, assume the SR latch shown in Figure 4-6 was part of a larger, more
complex, loop coupling network. In this case, loop circuitry duplication would turn into
an iterative process that would never converge. So, the tools would have to cut the loop
as shown in Figure 4-9.

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Figure 4-9. Cutting Coupling Loops

A Modified
P
Truth Table
AB PQ
0 0 1 1
B Q 0 1 1 X
1 0 0 1
1 1 X X
TIEX

The modified truth table shown in Figure 4-9 demonstrates that this method yields the
most pessimistic simulation results of all the loop-cutting methods. Because this is the
most pessimistic solution to the loop cutting problem, the tools only use this technique
when they cannot use any of the previous methods.

ATPG-Specific Combinational Loop Handling Issues

By default, the ATPG tool performs parallel pattern simulation of circuits containing
combinational feedback networks.
This is controlled by using the set_loop_handling command.

A learning process identifies feedback networks after flattening, and an iterative simulation is
used in the feedback network. For an iterative simulation, the ATPG tool inserts FB_BUF gates
to break the combinational loops.

The ATPG tool also has the ability to insert TIE-X gates to break the combinational loops. The
gate duplication option reduces the impact that a TIE-X gate places on the circuit to break
combinational loops. By default, this duplication switch is off.

Note
The set_loop_handling command replaces functionality previously available by the Set
Loop Duplication command.

Tessent Scan-Specific Combinational Loop Handling

Issues
Tessent Scan identifies combinational loops during flattening.
By default, it performs TIE-X insertion using the methods specified in “Structural
Combinational Loops and Loop-Cutting Methods” on page 94 to break all loops detected by the

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initial loop analysis. You can turn loop duplication off using the set_loop_duplication
command.

You can report on loops using the report_loops or the report_feedback_paths commands. While
both involved with loop reporting, these commands behave somewhat differently. Refer to the
Tessent Shell Reference Manual for details. You can write all identified structural
combinational loops to a file using the write_loops command.

You can use the loop information Tessent Scan provides to handle each loop in the most
desirable way. For example, assuming you wanted to improve the test coverage for a coupling
loop, you could use the add_control_points/add_observe_points commands within Tessent Scan
to insert a test point to control or observe values at a certain location within the loop.

Structural Sequential Loops and Handling

Sequential feedback loops occur when the output of a latch or flip-flop feeds back to one of its
inputs, either directly or through some other logic.
Figure 4-10 shows an example of a structural sequential feedback loop.

Figure 4-10. Sequential Feedback Loop

RST
D Q
Flip-flop

Note
The tools model RAM and ROM gates as combinational gates, and thus, they consider
loops involving only combinational gates and RAMs (or ROMs) as combinational loops–
not sequential loops.

The following sections provide tool-specific issues regarding sequential loop handling.

ATPG-Specific Sequential Loop Handling

While the ATPG tool can suffer some loss of test coverage due to sequential loops, these loops
do not cause the tool the extensive problems that combinational loops do.

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By its very nature, the ATPG tool re-models the non-scan sequential elements in the design
using the simulation primitives described in “ATPG Handling of Non-Scan Cells” on page 105.
Each of these primitives, when inserted, automatically breaks the loops in some manner.

Within the ATPG tool, sequential loops typically trigger C3 and C4 design rules violations.
When one sequential element (a source gate) feeds a value to another sequential element (a sink
gate), the tool simulates old data at the sink. You can change this simulation method using the
set_capture_handling command. For more information on the C3 and C4 rules, refer to “Clock
Rules” in the Tessent Shell Reference Manual. For more information about the
set_capture_handling command, refer to its description in the Tessent Shell Reference Manual.

Redundant Logic
In most cases, you should avoid using redundant logic because a circuit with redundant logic
poses testability problems. First, classifying redundant faults takes a great deal of analysis
effort.
Additionally, redundant faults, by their nature, are untestable and therefore lower your fault
coverage. Figure 2-18 on page 58 gives an example of redundant circuitry.

Some circuitry requires redundant logic; for example, circuitry to eliminate race conditions or
circuitry which builds high reliability into the design. In these cases, you should add test points
to remove redundancy during the testing process.

Asynchronous Sets and Resets

Scannability checking treats sequential elements driven by uncontrollable set and reset lines as
unscannable. You can remedy this situation in one of two ways: you can add test logic to make
the signals controllable, or you can use initialization patterns during test to control these
internally-generated signals. Tessent Scan provides capabilities to aid you in both solutions.
Figure 4-11 shows a situation with an asynchronous reset line and the test logic added to control
the asynchronous reset line.

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Figure 4-11. Test Logic Added to Control Asynchronous Reset

B B

D Q D Q

Clk Clk
R R
A A
RST RST Q
D Q D

Clk Clk
test_mode

In this example, Tessent Scan adds an OR gate that uses the test_mode (not scan_enable) signal
to keep the reset of flip-flop B inactive during the testing process. You would then constrain the
test_mode signal to be a 1, so flip-flop B could never be reset during testing. To insert this type
of test logic, you can use the Tessent Scan command set_test_logic (see page 124 for more
information).

Tessent Scan also allows you to specify an initialization sequence in the test procedure file to
avoid the use of this additional test logic. For additional information, refer to the
add_scan_groups description in the Tessent Shell Reference Manual.

Gated Clocks
Primary inputs typically cannot control the gated clock signals of sequential devices. In order to
make some of these sequential elements scannable, you may need to add test logic to modify
their clock circuitry.
For example, Figure 4-12 shows an example of a clock that requires some test logic to control it
during test mode.

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Figure 4-12. Test Logic Added to Control Gated Clock

D Q

D Q Clk

Clk

D Q

D Q Clk

Clk
test_clock
test_mode

In this example, Tessent Scan makes the element scannable by adding a test clock, for both scan
loading/unloading and data capture, and multiplexing it with the original clock signal. It also
adds a signal called test_mode to control the added multiplexer. The test_mode signal differs
from the scan_mode or scan_enable signals in that it is active during the entire duration of the
test—not just during scan chain loading/unloading. To add this type of test logic into your
design, you can use the set_test_logic and set_scan_signals commands. For more information
about these commands, refer to pages 124 and 142, respectively.

Tri-State Devices
Tri-state™ buses are another testability challenge. Faults on tri-state bus enables can cause one
of two problems: bus contention, which means there is more than one active driver, or bus float,
which means there is no active driver. Either of these conditions can cause unpredictable logic
values on the bus, which allows the enable line fault to go undetected.
Figure 4-13 shows a tri-state bus with bus contention caused by a stuck-at-1 fault.

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Figure 4-13. Tri-state Bus Contention

Enable line stuck-at-1

Unpredictable voltage on
bus may cause fault to
0 0 go unnoticed.

Enable line active

1 1

Tessent Scan can add gating logic that turns off the tri-state devices during scan chain shifting.
The tool gates the tri-state device enable lines with the scan_enable signal so they are inactive
and thus prevent bus contention during scan data shifting. To insert this type of gating logic, you
can use the Tessent Scan command set_tristate_gating (see page 124 for more information).

In addition, the ATPG tool lets you specify the fault effect of bus contention on tri-state nets.
This capability increases the testability of the enable line of the tri-state drivers. Refer to the
set_net_dominance description in the Tessent Shell Reference Manual for details.

Non-Scan Cell Handling

During rules checking and learning analysis, the ATPG tool learns the behavior of all state
elements that are not part of the scan circuitry. This learning involves how the non-scan element
behaves after the scan loading operation. As a result of the learning analysis, the ATPG tool
categorizes each of the non-scan cells.

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ATPG Handling of Non-Scan Cells

The ATPG tool places non-scan cells in one of the following categories:
• TIEX — In this category, the ATPG tool considers the output of a flip-flop or latch to
always be an X value during test. This condition may prevent the detection of a number
of faults.
• TIE0 — In this category, the ATPG tool considers the output of a flip-flop or latch to
always be a 0 value during test. This condition may prevent the detection of a number of
faults.
• TIE1 — In this category, the ATPG tool considers the output of a flip-flop or latch to
always be a 1 value during test. This condition may prevent the detection of a number of
faults.
• Transparent (combinational) — In this category, the non-scan cell is a latch, and the
latch behaves transparently. When a latch behaves transparently, it acts, in effect, as a
buffer—passing the data input value to the data output. The TLA simulation gate models
this behavior. Figure 4-14 shows the point at which the latch must exhibit transparent
behavior.

Figure 4-14. Requirement for Combinationally Transparent Latches

Basic Scan Pattern

---------------------------
Transparent Load scan chains
Behavior Force primary inputs
Here Measure primary outputs
Pulse capture clock
Unload scan chains
Transparency occurs if the clock input of the latch is inactive during the time between
the force of the primary inputs and the measure of the primary outputs. If your latch is
set up to behave transparently, you should not experience any significant fault detection
problems (except for faults on the clock, set, and reset lines). However, only in limited
cases do non-scan cells truly behave transparently. For the tool to consider the latch
transparent, it must meet the following conditions:
o The latch must not create a potential feedback path, unless the path is broken by scan
cells or non-scan cells (other than transparent latches).
o The latch must have a path that propagates to an observable point.
o The latch must be able to pass a data value to the output when all clocks are off.
o The latch must have clock, set, and reset signals that can be set to a determined
value.

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For more information on the transparent latch checking procedure, refer to “D6” in the
Tessent Shell Reference Manual.
• Sequential transparent — Sequential transparency extends the notion of transparency
to include non-scan elements that can be forced to behave transparently at the same
point in which natural transparency occurs. In this case, the non-scan element can be
either a flip-flop, a latch, or a RAM read port. A non-scan cell behaves as sequentially
transparent if, given a sequence of events, it can capture a value and pass this value to its
output, without disturbing critical scan cells.
Sequential transparent handling of non-scan cells lets you describe the events that place
the non-scan cell in transparent mode. You do this by specifying a procedure, called
seq_transparent, in your test procedure file. This procedure contains the events
necessary to create transparent behavior of the non-scan cell(s). After the tool loads the
scan chain, forces the primary inputs, and forces all clocks off, the seq_transparent
procedure pulses the clocks of all the non-scan cells or performs other specified events
to pass data through the cell “transparently”.
Figure 4-15 shows an example of a scan design with a non-scan element that is a
candidate for sequential transparency.

Figure 4-15. Example of Sequential Transparency

clock2
SI SO
Seq_trans Procedure
------------------------
scan Region 1 DFF Region 2 scan force clock2 0 0;
cell1 cell2 force clock2 1 1;
force clock2 0 2;
restore_pis;
PIs/scan cells PIs/scan cells

The DFF shown in Figure 4-15 behaves sequentially transparent when the tool pulses its
clock input, clock2. The sequential transparent procedure shows the events that enable
transparent behavior.

Note
To be compatible with combinational ATPG, the value on the data input line of the non-
scan cell must have combinational behavior, as depicted by the combinational Region 1.
Also, the output of the state element, in order to be useful for ATPG, must propagate to
an observable point.

Benefits of sequential transparent handling include more flexibility of use compared to

transparent handling, and the ability to use this technique for creating “structured partial
scan” (to minimize area overhead while still obtaining predictable high test coverage).

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Also, the notion of sequential transparency supports the design practice of using a cell
called a transparent slave. A transparent slave is a non-scan latch that uses the slave
clock to capture its data. Additionally, you can define and use up to 32 different,
uniquely-named seq_transparent procedures in your test procedure file to handle the
various types of non-scan cell circuitry in your design.
Rules checking determines if non-scan cells qualify for sequential transparency via these
procedures. Specifically, the cells must satisfy rules P5, P6, P41, P44, P45, P46, D3, and
D9. For more information on these rules, refer to “Design Rule Checking” in the Tessent
Shell Reference Manual. Clock rules checking treats sequential transparent elements the
same as scan cells.
Limitations of sequential transparent cell handling include the following:
o Impaired ability to detect AC defects (transition fault type causes sequential
transparent elements to appear as tie-X gates).
o Cannot make non-scan cells clocked by scan cells sequentially transparent without
condition statements.
o Limited usability of the sequential transparent procedure if applying it disturbs the
scan cells (contents of scan cells change during the seq_transparent procedure).
o Feedback paths to non-scan cells, unless broken by scan cells, prevent treating the
non-scan cells as sequentially transparent.
• Clock sequential — If a non-scan cell obeys the standard scan clock rules—that is, if
the cell holds its value with all clocks off—the tool treats it as a clock sequential cell. In
this case, after the tool loads the scan chains, it forces the primary inputs and pulses the
clock/write/read lines multiple times (based on the sequential depth of the non-scan
cells) to set up the conditions for a test. A normal observe cycle then follows. Figure 4-
16 shows a clock sequential scan pattern.

Figure 4-16. Clocked Sequential Scan Pattern Events

Clock Sequential Scan Pattern

--------------------------------------------
Repeat “N” Load scan chains
times for
sequential Force primary inputs
depth Pulse clock/read/write signals
Force primary inputs
Measure primary outputs
Pulse capture clock
Unload scan chains
This technique of repeating the primary input force and clock pulse allows the tool to
keep track of new values on scan cells and within feedback paths.

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When DRC performs scan cell checking, it also checks non-scan cells. When the
checking process completes, the rules checker issues a message indicating the number of
non-scan cells that qualify for clock sequential handling.
You instruct the tool to use clock sequential handling by selecting the -Sequential option
to the set_pattern_type command. During test generation, the tool generates test patterns
for target faults by first attempting combinational, and then RAM sequential techniques.
If unsuccessful with these techniques, the tool performs clock sequential test generation
if you specify a non-zero sequential depth.

Note
Setting the -Sequential switch to either 0 (the default) or 1 results in patterns with a
maximum sequential depth of one, but the tool creates clock sequential patterns only if
the setting is 1 or higher.

To report on clock sequential cells, you use the report_nonscan_cells command. For
more information on setting up and reporting on clock sequential test generation, refer to
the set_pattern_type and report_nonscan_cells descriptions in the Tessent Shell
Reference Manual.
Limitations of clock sequential non-scan cell handling include:
o The maximum allowable sequential depth is 255 (a typical depth would range from 2
to 5).
o Copy and shadow cells cannot behave sequentially.
o The tool cannot detect faults on clock/set/reset lines.
o You cannot use the read-only mode of RAM testing with clock sequential pattern
generation.
o The tool simulates cells that capture data on a trailing clock edge (when data changes
on the leading edge) using the original values on the data inputs.
o Non-scan cells that maintain a constant value after load_unload simulation are
treated as tied latches.
o This type of testing has high memory and performance costs.

Clock Dividers
Some designs contain uncontrollable clock circuitry; that is, internally-generated signals that
can clock, set, or reset flip-flops. If these signals remain uncontrollable, Tessent Scan will not
consider the sequential elements controlled by these signals “scannable”. And consequently,
they could disturb sequential elements during scan shifting. Thus, the system cannot convert
these elements to scan.

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Figure 4-17 shows an example of a sequential element (B) driven by a clock divider signal and
with the appropriate circuitry added to control the divided clock signal.

Figure 4-17. Clock Divider

DATA
DATA D Q
D Q
B B
D Q
D Q'
Q Q' A
A CLK Q'
CLK Q' TST_CLK
TST_EN

Tessent Scan can assist you in modifying your circuit for maximum controllability (and thus,
maximum scannability of sequential elements) by inserting special circuitry, called test logic, at
these nodes when necessary. Tessent Scan typically gates the uncontrollable circuitry with chip-
level test pins. In the case of uncontrollable clocks, Tessent Scan adds a MUX controlled by the
test_clk and test_en signals.

For more information on test logic, refer to “Enabling Test Logic Insertion” on page 124.

Pulse Generators
A pulse generator is circuitry that creates a pulse at its output when active.
Figure 4-18 gives an example of pulse generator circuitry.

Figure 4-18. Example Pulse Generator Circuitry

A
A
C B
B C

When designers use this circuitry in clock paths, there is no way to create a stable on state.
Without a stable on state, the fault simulator and test generator have no way to capture data into
the scan cells. Pulse generators also find use in write control circuitry, a use that impedes RAM
testing.

By default, the ATPG tool identifies the reconvergent pulse generator sink (PGS) gates, or
simply “pulse generators”, during the learning process. For the tools to provide support, a
“pulse generator” must satisfy the following requirements:

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• The “pulse generator” gate must have a connection (at C in Figure 4-18) to a clock input
of a memory element or a write line of a RAM.
• The “pulse generator” gate must be an AND, NAND, OR, or NOR gate.
• Two inputs of the “pulse generator” gate must come from one reconvergent source gate.
• The two reconvergent paths may only contain inverters and buffers.
• There must be an inversion difference in the two reconvergent paths.
• The two paths must have different lengths (propagation times).
• In the long path, the inverter or buffer that connects to the “pulse generator” input must
only go to gates of the same gate type as shown in (a) in Figure 4-19. A fanout to gates
of different types as in (b) in the figure is not supported. The tools model this input gate
as tied to the non-controlling value of the “pulse generator” gate (TIE1 for AND and
NAND gates, TIE0 for OR and NOR gates).

Figure 4-19. Long Path Input Gate Must Go to Gates of the Same Type
(a) Supported: (b) Not Supported:
A A
C C
B B

Input gate fans out to

gates of different types
Input gate fans out to
gates of the same type

Rules checking includes some checking for “pulse generator” gates. Specifically, Trace rules
#16 and #17 check to ensure proper usage of “pulse generator” gates. Refer to “T16” and “T17”
in the Tessent Shell Reference Manual for more details about these rules.

The ATPG tool supports pulse generators with multiple timed outputs. For detailed information
about this support, refer to “Pulse Generators with User Defined Timing” in the Tessent Cell
Library User’s Manual.

JTAG-Based Circuits
Boundary scan circuitry, as defined by IEEE standard 1149.1, can result in a complex
environment for the internal scan structure and the ATPG process. The two main issues with
boundary scan circuitry are 1) connecting the boundary scan circuitry with the internal scan
circuitry, and 2) ensuring that the boundary scan circuitry is set up properly during ATPG.

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Testing RAM and ROM

The three basic problems of testing designs that contain RAM and ROM are 1) modeling the
behavior, 2) passing rules checking to allow testing, and 3) detecting faults during ATPG.
The “RAM and ROM” section in the Tessent Cell Library User’s Manual discusses modeling
RAM and ROM behavior. The “RAM Rules” section in the Tessent Shell Reference Manual
discusses RAM rules checking. This section primarily discusses the techniques for detecting
faults in circuits with RAM and ROM during ATPG. The “RAM Summary Results and Test
Capability” section of the Tessent Shell Reference Manual discusses displayed DRC summary
results upon completion of RAM rules checking.

The ATPG tool does not test the internals of the RAM/ROM, although MacroTest (separately
licensed but available in the ATPG tool) lets you create tests for small memories such as register
files by converting a functional test sequence or algorithm into a sequence of scan tests. For
large memories, built-in test structures within the chip itself are the best methods of testing the
internal RAM or ROM.

However, the ATPG tool needs to model the behavior of the RAM/ROM so that tests can be
generated for the logic on either side of the embedded memory. This allows the tool to generate
tests for the circuitry around the RAM/ROM, as well as the read and write controls, data lines,
and address lines of the RAM/ROM unit itself.

Figure 4-20 shows a typical configuration for a circuit containing embedded RAM.

Figure 4-20. Design with Embedded RAM

L L
O CONTROL O
G D G
I E I
C DATA POs
PIs C OUT C
ADDR O RAM
B D B and SLs
and SLs L E L
O R O
DATA IN C
C
K K

A B

ATPG must be able to operate the illustrated RAM to observe faults in logic block A, as well as
to control the values in logic block B to test faults located there. The ATPG tool has unique
strategies for operating the RAMs.

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RAM/ROM Support
The tool treats a ROM as a strictly combinational gate. Once a ROM is initialized, it is a simple
task to generate tests because the contents of the ROM do not change. Testing RAM however, is
more of a challenge, because of the sequential behavior of writing data to and reading data from
the RAM.
The tool supports the following strategies for propagating fault effects through the RAM:

• Read-only mode — The tool assumes the RAM is initialized prior to scan test and this
initialization must not change during scan. This assumption allows the tool to treat a
RAM as a ROM. As such, there is no requirement to write to the RAM prior to reading,
so the test pattern only performs a read operation. Important considerations for read-
only mode test patterns are as follows:
o The read-only testing mode of RAM only tests for faults on data out and read
address lines, just as it would for a ROM. The tool does not test the write port I/O.
o To use read-only mode, the circuit must pass rules A1 and A6.
o Values placed on the RAM are limited to initialized values.
o Random patterns can be useful for all RAM configurations.
o You must define initial values and assume responsibility that those values are
successfully placed on the correct RAM memory cells. The tool does not perform
any audit to verify this is correct, nor will the patterns reflect what needs to be done
for this to occur.
o Because the tester may require excessive time to fully initialize the RAM, it is
allowed to do a partial initialization.
• Pass-through mode — The tool has two separate pass-through testing modes:
o Static pass-through — To detect faults on data input lines, you must write a known
value into some address, read that value from the address, and propagate the effect to
an observation point. In this situation, the tool handles RAM transparently, similar to
the handling of a transparent latch. This requires several simultaneous operations.
The write and read operations are both active and thus writing to and reading from
the same address. While this is a typical RAM operation, it allows testing faults on
the data input and data output lines. It is not adequate for testing faults on read and
write address lines.
o Dynamic pass-through — This testing technique is similar to static pass-through
testing except one pulse of the write clock performs both the write and read
operation (if the write and read control lines are complementary). While static pass-
through testing is comparable to transparent latch handling, dynamic pass-through
testing compares to sequential transparent testing.

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• Sequential RAM test mode — This is the recommended approach to RAM testing.
While the previous testing modes provide techniques for detecting some faults, they
treat the RAM operations as combinational. Thus, they are generally inadequate for
generating tests for circuits with embedded RAM. In contrast, this testing mode tries to
separately model all events necessary to test a RAM, which requires modeling
sequential behavior. This enables testing of faults that require detection of multiple
pulses of the write control lines. These faults include RAM address and write control
lines.
RAM sequential testing requires its own specialized pattern type. RAM sequential
patterns consist of one scan pattern with multiple scan chain loads. A typical RAM
sequential pattern contains the events shown in Figure 4-21.

Note
For RAM sequential testing, the RAM’s read_enable/write_enable control(s) can be
generated internally. However, the RAM’s read/write clock should be generated from a
PI. This ensures RAM sequencing is synchronized with the RAM sequential patterns.

Figure 4-21. RAM Sequential Example

RAM Sequential Pattern

load scan chains

write into
force primary inputs
one address
pulse write control lines

write into load scan chains

second address force primary inputs
pulse write control lines

get data on load scan chains

outputs force primary inputs
pulse read control lines

basic pattern load scan chain

events force primary inputs
measure primary outputs
pulse capture clock
unload scan chains

In this example of an address line test, assume that the MSB address line is stuck at 0.
The first write would write data into an address whose MSB is 0 to match the faulty
value, such as 0000. The second write operation would write different data into a
different address (the one obtained by complementing the faulty bit). For this example, it
would write into 1000. The read operation then reads from the first address, 0000. If the
highest order address bit is stuck-at-0, the 2nd write would have overwritten the original

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data at address 0, and faulty circuitry data would be read from that address in the 3rd
step.
Another technique that may be useful for detecting faults in circuits with embedded RAM is
clock sequential test generation. It is a more flexible technique, which effectively detects faults
associated with RAM. “Clock Sequential Patterns” on page 174 discusses clock sequential test
generation in more detail.

Common Read and Clock Lines

Ram_sequential simulation supports RAMs whose read line is common with a scan clock. The
tool assumes that the read and capture operation can occur at the same time and that the value
captured into the scan cell is a function of the value read out from the RAM.
If the clock that captures the data from the RAM is the same clock which is used for reading, the
tool issues a C6 clock rules violation. This indicates that you must set the clock timing so that
the scan cell can successfully capture the newly read data.

If the clock that captures the data from the RAM is not the same clock that is used for reading,
you will likely need to turn on multiple clocks to detect faults. The default
“set_clock_restriction On” command is conservative, so the tool will not allow these patterns,
resulting in a loss in test coverage. If you issue the “set_clock_restriction Off” command, the
tool allows these patterns, but there is a risk of inaccurate simulation results because the
simulator does not propagate captured data effects.

Common Write and Clock Lines

The tool supports common write and clock lines. The following shows the support for common
write and clock lines:
• You can define a pin as both a write control line and a clock if the off-states are the same
value. the tool then displays a warning message indicating that a common write control
and clock has been defined.
• The rules checker issues a C3 clock rule violation if a clock can propagate to a write line
of a RAM, and the corresponding address or data-in lines are connected to scan latches
which has a connection to the same clock.
• The rules checker issues a C3 clock rule violation if a clock can propagate to a read line
of a RAM, and the corresponding address lines are connected to scan latches which has
a connection to the same clock.
• The rules checker issues a C3 clock rule violation if a clock can capture data into a scan
latch that comes from a RAM read port that has input connectivity to latches which has a
connection to the same clock.
• If you set the simulation mode to Ram_sequential, the rules checker will not issue an A2
RAM rule violation if a clock is connected to a write input of a RAM. Any clock
connection to any other input (including the read lines) will continue to be a violation.

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• If a RAM write line is connected to a clock, you cannot use the dynamic pass through
test mode.
• Patterns which use a common clock and write control for writing into a RAM will be in
the form of ram_sequential patterns. This requires you to set the simulation mode to
Ram_sequential.
• If you change the value of a common write control and clock line during a test
procedure, you must hold all write, set, and reset inputs of a RAM off. The tool
considers failure to satisfy this condition as an A6 RAM rule violation and disqualifies
the RAM from being tested using read_only and ram_sequential patterns.

RAM/ROM Support Commands

The tool requires certain knowledge about the design prior to test generation. For circuits with
RAM, you must define write controls, and if the RAM has data hold capabilities, you must also
define read controls. Just as you must define clocks so the tool can effectively write scan
patterns, you must also define these control lines so it can effectively write patterns for testing
RAM. And similar to clocks, you must define these signals in setup mode, prior to rules
checking.
The commands in Table 4-1 support the testing of designs with RAM and/or ROM.

Table 4-1. RAM/ROM Commands

Command Name Description
add_read_controls Defines a PI as a read control and specifies its off value.
add_write_controls Defines a PI as a write control and specifies its off value.
create_initialization_ Creates RAM initialization patterns and places them in the internal
patterns pattern set.
delete_read_controls Removes the read control line definitions from the specified
primary input pins.
delete_write_controls Removes the write control line definitions from the specified
primary input pins.
read_modelfile Initializes the specified RAM or ROM gate using the memory states
contained in the specified modelfile.
report_read_controls Displays all of the currently defined read control lines.
report_write_controls Displays all of the currently defined write control lines.
set_pattern_type Specifies whether the ATPG simulation run uses combinational or
sequential RAM test patterns.
set_ram_initialization Specifies whether to initialize RAM and ROM gates that do not
have initialization files.

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Table 4-1. RAM/ROM Commands (cont.)

Command Name Description
set_ram_test Sets the RAM testing mode to either read_only, pass_thru, or
static_pass_thru.
write_modelfile Writes all internal states for a RAM or ROM gate into the file that
you specify.

Basic ROM/RAM Rules Checking

The rules checker performs the following audits for RAMs and ROMs:
• The checker reads the RAM/ROM initialization files and checks them for errors. If you
selected random value initialization, the tool gives random values to all RAM and ROM
gates without an initialized file. If there are no initialized RAMs, you cannot use the
read-only test mode. If any ROM is not initialized, an error condition occurs. A ROM
must have an initialization file but it may contain all Xs. Refer to the read_modelfile
description in the Tessent Shell Reference Manual for details on initialization of
RAM/ROM.
• The RAM/ROM instance name given must contain a single RAM or ROM gate. If no
RAM or ROM gate exists in the specified instance, an error condition occurs.
• If you define write control lines and there are no RAM gates in the circuit, an error
condition occurs. To correct this error, delete the write control lines.
• When the write control lines are off, the RAM set and reset inputs must be off and the
write enable inputs of all write ports must be off. You cannot use RAMs that fail this
rule in read-only test mode. If any RAM fails this check, you cannot use dynamic pass-
through. If you defined an initialization file for a RAM that failed this check, an error
condition occurs. To correct this error, properly define all write control lines or use
lineholds (pin constraints).
• A RAM gate must not propagate to another RAM gate. If any RAM fails this check, you
cannot use dynamic pass-through.
• A defined scan clock must not propagate directly (unbroken by scan or non-scan cells)
to a RAM gate. If any RAM fails this check, you cannot use dynamic pass-through.
• The tool checks the write and read control lines for connectivity to the address and data
inputs of all RAM gates. It gives a warning message for all occurrences and if
connectivity fails, there is a risk of race conditions for all pass-through patterns.
• A RAM that uses the edge-triggered attribute must also have the read_off attribute set to
hold. Failure to satisfy this condition results in an error condition when the design
flattening process is complete.
• If the RAM rules checking identifies at least one RAM that the tool can test in read-only
mode, it sets the RAM test mode to read-only. Otherwise, if the RAM rules checking

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passes all checks, it sets the RAM test mode to dynamic pass-through. If it cannot set the
RAM test mode to read-only or dynamic pass-through, it sets the test mode to static
pass-through.
• A RAM with the read_off attribute set to hold must pass Design Rule A7 (when read
control lines are off, place read inputs at 0). The tool treats RAMs that fail this rule as:
o a TIE-X gate, if the read lines are edge-triggered.
o a read_off value of X, if the read lines are not edge-triggered.
• The read inputs of RAMs that have the read_off attribute set to hold must be at 0 during
all times of all test procedures, except the test_setup procedure.
• The read control lines must be off at time 0 of the load_unload procedure.
• A clock cone stops at read ports of RAMs that have the read_off attribute set to hold,
and the effect cone propagates from its outputs.
For more information on the RAM rules checking process, refer to “RAM Rules” in the Tessent
Shell Reference Manual.

Incomplete Designs
The ATPG tool and Tessent Scan can iread incomplete Verilog designs due to their ability to
generate black boxes. The Verilog parser can blackbox any instantiated module or instance that
is not defined in either the ATPG library or the design netlist. The tool issues a warning
message for each blackboxed module similar to the following:
// WARNING: Following modules are undefined:
// ao21
// and02
// Use "add_black_box -auto" to treat these as black boxes.

If the tool instantiates an undefined module, it generates a module declaration based on the
instantiation. If ports are connected by name, the tool uses those port names in the generated
module. If ports are connected by position, the parser generates the port names. Calculating port
directions is problematic and must be done by looking at the other pins on the net connected to
the given instance pin. For each instance pin, if the connected net has a non-Z-producing driver,
the tool considers the generated module port an input, otherwise the port is an output. The tool
never generates inout ports since they cannot be inferred from the other pins on the net.

Modules that are automatically blackboxed default to driving X on their outputs. Faults that
propagate to the black box inputs are classified as ATPG_untestable (AU). To change the
output values driven, refer to the add_black_box description in the Tessent Shell Reference
Manual.

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Inserting Internal Scan
and Test Circuitry

This chapter explains the how to use Tessent Scan to insert scan circuitry in your design.
Figure 5-1 shows the process of inserting scan and other test circuitry with Tessent Scan or
Tessent Shell operating in “dft -scan” context.

Figure 5-1. Internal Scan Insertion Procedure

Insert/Verify 1. The Tessent Scan Process Flow

BScan Circuitry
2. Preparing for Test Structure Insertion

Insert Internal 3. Setting Up a Basic Scan Insertion Run

Scan/Test Circuitry 4. Inserting Test Structures
(Tessent Scan)
5. Saving the New Design and ATPG Setup
Generate/Verify
Test Patterns 6. Inserting Scan Block-by-Block
(ATPG Tool)

The Tessent Scan Process Flow

The following figure shows the basic flow for synthesizing scan circuitry with Tessent Scan.

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The Tessent Scan Process Flow

Figure 5-2. Basic Scan Insertion Flow with Tessent Scan

From
Synthesis
DFT Synthesized
Library Netlist

Setup
Mode Set Up Circuit and
Tool Information

Run Design Rules

and Testability
Analysis

Analysis
Mode N
Pass Troubleshoot
Checks? Problem

Y
Identify
Test Structures

Insert
Test Structures Test
Procedure
File
Netlist with Save Design and
Test ATPG Information
Structures

Dofile
To ATPG

You start with a DFT library and a synthesized design netlist. The library is the same one that
the ATPG tool uses. “Tessent Scan Inputs and Outputs” on page 121 describes the netlist
formats you can use with Tessent Scan. The design netlist you use as input may be an individual
block of the design, or the entire design.

After invoking the tool, your first task is to set up information about the design—this includes
both circuit information and information about the test structures you want to insert. “Preparing
for Test Structure Insertion” on page 123 describes the procedure for this task. The next task
after setup is to run rules checking and testability analysis, and debug any violations that you
encounter. “Changing the System Mode (Running Rules Checking)” on page 131 documents
the procedure for this task.

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The Tessent Scan Process Flow

Note
To catch design violations early in the design process, you should run and debug design
rules on each block as it is synthesized.

After successfully completing rules checking, you will be in the analysis system mode. At this
point, if you have any existing scan you want to remove, you can do so. “Handling Existing
Boundary Scan Circuitry” on page 130 describes the procedure for doing this. You can then set
up specific information about the scan or other testability circuitry you want added and identify
which sequential elements you want converted to scan. “Setting Up a Basic Scan Insertion Run”
on page 131 describes the procedure for accomplishing this. Finally, with these tasks
completed, you can insert the desired test structures into your design. “Inserting Test
Structures” on page 141 describes the procedure for this insertion.

Tessent Scan Inputs and Outputs

The following figure shows the inputs used and the outputs produced by Tessent Scan.

Figure 5-3. The Inputs and Outputs of Tessent Scan

Circuit
Setup
Design (Dofile) Library

Test
Tessent Scan Procedure
File

ATPG
Design Setup
(Dofile)

Tessent Scan uses the following inputs:

• Design (netlist) — The supported design data format is gate-level Verilog.

• Circuit Setup (or Dofile) — This is the set of commands that gives the tool information
about the circuit and how to insert test structures. You can issue these commands
interactively in the tool session or place them in a dofile.
• Library — The design library contains descriptions of all the cells the design uses. The
library also includes information that the tool uses to map non-scan cells to scan cells

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and to select components for added test logic circuitry. The tool uses the library to
translate the design data into a flat, gate-level simulation model on which it runs its
internal processes.
• Test Procedure File — This file defines the stimulus for shifting scan data through the
defined scan chains. This input is only necessary on designs containing preexisting scan
circuitry or requiring test setup patterns.
Tessent Scan produces the following outputs:

• Design (Netlist) — This netlist contains the original design modified with the inserted
test structures. The output netlist format is gate-level Verilog.
• ATPG Setup (Dofile and Test Procedure File) — The tool can automatically create a
dofile and test procedure file that you can supply to the ATPG tool. These files contain
the circuit setup information that you specified to the tool as well as information on the
test structures that the tool inserted into the design. The tool creates these files for you
when you issue the write_atpg_setup command.

Note
The timeplate in the test procedure file defines a generic waveform for all of the clocks in
the design (including set and reset signals that were defined using the add_clocks
command) and not just for those clocks that are required for scan chain shifting.
However, the generated shift procedure only pulses clocks that drive scan cells. Internal
clock nets that can be mapped to a top level pin (by tracing a sensitized path) are not
included in the dofile and test procedure file, but internal clocks that cannot be traced
(such as PLL outputs) are included.

• Test Procedure File — When you issue the write_atpg_setup command, the tool writes
a simple test procedure file for the scan circuitry it inserted into the design. You use this
file with the downstream ATPG tool.

Test Structures Supported by Tessent Scan

Tessent Scan can identify and insert a variety of test structures, including several different scan
architectures and test points.
The following list briefly describes the test structures the tool supports:

• Scan — A flow where the tool converts all sequential elements that pass scannability
checking into scan cells. “Understanding Scan” on page 24 discusses the full scan style.
• Wrapper chains — A flow where the tool identifies sequential elements that interact
with input and output pins. These memory elements are converted into scan chains, and
the remaining sequential elements are not affected. For more information, see
“Understanding Wrapper Chains” on page 25.

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• Scan and/or Wrapper — A flow where the tool converts into scan cells those
sequential elements that interact with primary input and output pins, and then stitches
the scan cells into dedicated wrapper chains. The tool converts the remaining sequential
elements into scan cells and stitches them into separate chains, which are called core
chains.
• Test points — A flow where the tool inserts control and observe points at user specified
locations. “Understanding Test Points” on page 27 discusses the test points method.
Tessent Scan provides the ability to insert test points at user specified locations. If both scan and
test points are enabled during an identification run, the tool performs scan identification
followed by test point identification.

Invoking Tessent Scan

You access Tessent Scan functionality by invoking Tessent Shell and then setting the context to
“dft -scan.”

Procedure
% tessent -shell

SETUP> set_context dft -scan

The tool invokes in setup mode, ready for you to begin loading or working on your design. You
use this setup mode to define the circuit and scan data which is the next step in the process.

Preparing for Test Structure Insertion

The following subsections discuss the steps you would typically take to prepare for the insertion
of test structures into your design. When the tool invokes, you are in setup mode. All of the
setup steps shown in the following subsections occur in setup mode.

Defining Scan Cell and Scan Output Mapping

The tool uses the default mapping defined within the ATPG library. Each scan model in the
library describes how the non-scan models map to scan model in the scan_definition section of
the model.
For more information on the default mapping of the library model, refer to “Defining Cell
Information” in the Tessent Cell Library Manual.

You have the option to customize the scan cell and the cell’s scan output mapping behavior.
You can change the mapping for an individual instance, all instances under a hierarchical
instance, all instances in all occurrences of a module in the design, or all occurrences of the
model in the entire design, using the set_cell_model_mapping command.

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For example, you can map the fd1 nonscan model to the fd1s scan model for all occurrences of
the model in the design by entering:

set_cell_model_mapping -new_model fd1s -model fd1

The following example maps the fd1 nonscan model to the fd1s scan model for all matching
instances in the “counter” module and for all occurrences of that module in the design:

set_cell_model_mapping -new_model fd1s -model fd1 -module counter

Additionally, you can change the scan output pin of the scan model in the same manner as the
scan cell. Within the scan_definition section of the model, the scan_out attribute defines which
pin is used as the scan output pin. During the scan stitching process, the tool selects the output
pin based on the lowest fanout count of each of the possible pins. If you have a preference as to
which pin to use for a particular model or instance, you can also issue the
set_cell_model_mapping command to define that pin.

For example, if you want to use “qn” instead of “q” for all occurrences of the fd1s scan model in
the design, enter:

set_cell_model_mapping -new_model fd1s -output qn

Enabling Test Logic Insertion

Test logic is circuitry that the tool adds to improve the testability of a design. If so enabled, the
tool inserts test logic during scan insertion based on the analysis performed during the design
rules and scannability checking processes.
Test logic provides a useful solution to a variety of common problems. First, some designs
contain uncontrollable clock circuitry; that is, internally-generated signals that can clock, set, or
reset flip-flops. If these signals remain uncontrollable, the tool will not consider the sequential
elements controlled by these signals scannable. Second, you might want to prevent bus
contention caused by tri-state devices during scan shifting.

Tessent Scan can assist you in modifying your circuit for maximum controllability (and thus,
maximum scannability of sequential elements) and bus contention prevention by inserting test
logic circuitry at these nodes when necessary.

Note
Tessent Scan does not attempt to add test logic to user-defined non-scan instances or
models; that is, those specified by the add_nonscan_instances or the
add_nonscan_models commands.

Tessent Scan typically gates the uncontrollable circuitry with a chip-level test pin. Figure 5-4
shows an example of test logic circuitry.

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Figure 5-4. Test Logic Insertion

Before After
Uncontrollable Clock Added Test Logic
Test_en

DRQ DRQ D R Q CL DRQ

CL Sc_in
Sc_in Sc_in Sc_in
Sc_en Sc_en Sc_en Sc_en
CK CK CK CK

You can specify which signals to insert test logic on with the set_test_logic.

You can add test logic to all uncontrollable (set, reset, clock, or RAM write control) signals
during the scan insertion process. By default, Tessent Scan does not add test logic. You must
explicitly enable the use of test logic with this command.

In adding the test logic circuitry, the tool performs some basic optimizations in order to reduce
the overall amount of test logic needed. For example, if the reset line to several flip-flops is a
common internally-generated signal, the tool gates it at its source before it fans out to all the
flip-flops.

Note
You must turn the appropriate test logic on if you want the tool to consider latches as scan
candidates. Refer to “D6” in the Tessent Shell Reference Manual for more information on
scan insertion with latches.

If your design uses bidirectional pins as scan I/Os, the tool controls the scan direction for the
bidirectional pins for correct shift operation.

This can be specified with set_bidi_gating. If the enable signal of the bidirectional pin is
controlled by a primary input pin, then the tool adds a “force” statement for the enable pin in the
new load_unload procedure to enable/disable the correct direction. Otherwise, the tool inserts
gating logic to control the enable line. The gate added to the bidirectional enable line is either a
2-input AND or OR. By default, no bidirectional gating is inserted and you must make sure that
the inserted scan chains function properly by sensitizing the enable lines of any bidirectional
ports in the scan path.

There are four possible cases between the scan direction and the active values of a tri-state
driver, as shown in Table 5-1. The second input of the gate is controlled from the scan_enable

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signal, which might be inverted. You will need to specify AND and OR models through the
cell_type keyword in the ATPG library or use the add_cell_models command.

Table 5-1. Scan Direction and Active Values

Driver Scan Direction Gate Type
active high input AND
active high output OR
active low input OR
active low output AND

If you enable the gating of bidirectional pins, the tool controls all bidirectional pins. The
bidirectional pins not used as scan I/Os are put into input mode (Z state) during scan shifting by
either “force” statements in the new load_unload procedure or by using gating logic.

The tool adds a “force Z” statement in the test procedure file for the output of the bidirectional
pin if it is used as scan output pin. This ensures that the bus is not driven by the tristate drivers
of both bidirectional pin and the tester at the same time.

Specifying the Models to use for Test Logic

When adding test logic circuitry, the tool uses a number of gates from the library. The cell_type
attribute in the library model descriptions tells the tool which components are available for use
as test logic.
If the library does not contain this information, you can instead specify which library models to
use with the add_cell_models command.

Note
Tessent Scan treats any Verilog module enclosed in `celldefine / `endcelldefine
directives as a library cell and prevents any logic changes to these modules.

Issues Concerning Test Logic Insertion and Test Clocks

Because inserting test logic actually adds circuitry to the design, you should first try to increase
circuit controllability using other options. These options might include such things as
performing proper circuit setup or, potentially, adding test points to the circuit prior to scan.
Additionally, you should re-optimize a design to ensure that fanout resulting from test logic is
correctly compensated and passes electrical rules checks.
In some cases, inserting test logic requires the addition of multiple test clocks. Analysis run
during DRC determines how many test clocks the tool needs to insert. The report_scan_chains
command reports the test clock pins used in the scan chains.

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Related Test Logic Commands

In addition to commands mentioned in previous sections, the following commands are useful
when working with test logic:
• delete_cell_models — Deletes the information specified by the add_cell_models
command.
• report_cell_models — Displays a list of library cell models to be used for adding test
logic circuitry.
• report_test_logic — Displays a list of test logic added during scan insertion.

Specifying Clock Signals

Tessent Scan must be aware of the circuit clocks to determine which sequential elements are
eligible for scan. The tool considers clocks to be any signals that have the ability to alter the
state of a sequential device (such as system clocks, sets, and resets).
Therefore, you need to tell the tool about these “clock signals” by adding them to the clock list
with the add_clocks command.

With this command, you must specify the off-state for pins you add to the clock list. The off-
state is the state in which clock inputs of latches are inactive. For edge-triggered devices, the off
state is the clock value prior to the clock’s capturing transition.

For example, you might have two system clocks, called “clk1” and “clk2”, whose off-states are
0 and a global reset line called “rst_l” whose off-state is 1 in your circuit. You can specify these
as clock lines as follows:

SETUP> add_clocks 0 clk1 clk2

SETUP> add_clocks 1 rst_1

You can specify multiple clock pins with the same command if they have the same off-state.
You must define clock pins prior to entering analysis mode. Otherwise, none of the non-scan
sequential elements will successfully pass through scannability checks. Although you can still
enter analysis mode without specifying the clocks, the tool will not be able to convert elements
that the unspecified clocks control.

Note
If you are unsure of the clocks within a design, you can use the analyze_control_signals
command to identify and then define all the clocks. It also defines the other control
signals in the design.

• delete_clocks — Deletes primary input pins from the clock list.

• report_clocks — Displays a list of all clocks.

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• report_primary_inputs — Displays a list of primary inputs.

• write_primary_inputs — Writes a list of primary inputs to a file.

Specifying Existing Scan Information

You may have a design that already contains some existing internal scan circuitry. For example,
one block of your design may be reused from another design, and thus, may already contain its
own scan chain. You may also have used a third-party tool to insert scan before invoking the
tool. If either of these is your situation, there are several ways in which you may want to handle
the existing scan data, including, leaving the existing scan alone, deleting the existing scan, and
adding additional scan circuitry.
Note
If you are performing block-by-block scan synthesis, you should refer to “Inserting Scan
Block-by-Block” on page 148.

If your design contains existing scan chains that you want to use, you must specify this
information to the tool while you are in setup mode; that is, before design rules checking. If you
do not specify existing scan circuitry, the tool treats all the scan cells as non-scan cells and
performs non-scan cell checks on them to determine if they are scan candidates.

Common methodologies for handling existing scan circuitry include:

• Remove the existing scan chain(s) from the design and reverse the scan insertion
process. the tool will replace the scan cells with their non-scan equivalent cells. The
design can then be treated as you would any other new design to which you want to add
scan circuitry. This technique is often used when re-stitching scan cells based on
placement and routing results.
• Add additional scan chains based on the non-scan cells while leaving the original scan
chains intact.
• Stitch together existing scan cells that were previously unstitched.
The remainder of this section includes details related to these methodologies.

Specifying Existing Scan Groups

A scan chain group consists of a set of scan chains that are controlled through the same
procedures; that is, the same test procedure file controls the operation of all chains in the group.
If your design contains existing scan chains, you must specify the scan group to which they
belong, as well as the test procedure file that controls the group. To specify an existing scan
group, use the add_scan_groups command.

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For example, you can specify a group name of “group1” controlled by the test procedure file
“group1.test_proc” as follows:

SETUP> add_scan_groups group1 group1.test_proc

For information on creating test procedure files, refer to “Test Procedure Files” on page 74.

Specifying Existing Scan Chains

After specifying the existing scan group, you need to communicate to the tool which scan
chains belong to this group.
To specify existing scan chains, use the add_scan_chains command.

You need to specify the scan chain name, the scan group to which it belongs, and the primary
input and output pins of the scan chain. For example, assume your design has two existing scan
chains, “chain1” and “chain2”, that are part of “group1”. The scan input and output pins of
chain1 are “sc_in1” and “sc_out1”, and the scan input and output pins of chain2 are “sc_in2”
and “sc_out2”, respectively. You can specify this information as follows:

SETUP> add_scan_chains chain1 group1 sc_in1 sc_out1

SETUP> add_scan_chains chain2 group1 sc_in2 sc_out2

Specifying Existing Scan Cells

If the design has existing scan cells that are not stitched together in a scan chain, you need to
identify these cells for Tessent Scan. (You cannot define scan chains if the scan cells are not
stitched together.) This situation can occur if scan cells are used in the functional design to
provide actual timing.
Tessent Scan can insert scan cells without stitching if you use the -Connect {Tied | Loop |
Buffer} arguments to the insert_test_logic command.

Additionally, defining these existing scan cells prevents the tool from performing possibly
undesirable default actions, such as scan cell mapping and generation of unnecessary mux gates.

New Scan Cell Mapping

If you have existing scan cells, you must identify them as such to prevent the tool from
classifying them as replaceable by new scan cells. One or the other of the following criteria is
necessary for the tool to identify existing scan cells and not map them to new scan cells:
1. Declare the “data_in = <port_name>” in the scan_definition section of the scan cell’s
model in the ATPG library.
If you have a hierarchy of scan cell definitions, where one library cell can have another
library cell as its scan version, using the data_in declaration in a model causes the tool to

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consider that model as the end of the scan definition hierarchy, so that no mapping of
instances of that model will occur.

Note
It is not recommended that you create a hierarchy of scan cell model definitions. If, for
instance, your data_in declaration is in the scan_definitions section of the third model in
the definitions hierarchy, but the tool encounters an instance of the first model in the
hierarchy, it will replace the first model with the second model in the hierarchy, not the
desired third model. If you have such a hierarchy, you can use the
set_cell_model_mapping command to point to the desired model.
set_cell_model_mapping overrides the mapping defined in the library model.

2. The scan enable port of the instance of the cell model must be either dangling or tied (0
or 1) or pre-connected to a global scan enable pin(s). In addition, the scan input port
must be dangling or tied or connected to the cell’s scan output port as a self loop or a self
loop with (multiple) buffers or inverters.
Dangling implies that there are no connected fan-ins from other pins except tied pins or
tied nets. To identify an existing (global) scan enable, use the set_scan_signals
command.
Issue the set_scan_signals command before the insert_test_logic command.

Additional Mux Gates

Another consequence of not specifying existing scan cells is the addition of unnecessary
multiplexers, creating an undesirable area and routing overhead.
If you use criteria (a) as the means of preventing scan cell mapping, the tool also checks the
scan enable and scan in ports. If either one is driven by system logic, then the tool inserts a new
mux gate before the data input and uses it as a mux in front of the preexisting scan cell. (This is
only for mux-DFF scan; this mux is not inserted for LSSD or clocked_scan types of scan.)

If you use a combination of criteria (a) and (b), or just criteria (b), as the means of preventing
scan cell mapping, the tool will not insert a mux gate before the data input.

Once the tool can identify existing scan cells, they can be stitched into scan chains in the normal
scan insertion process.

Handling Existing Boundary Scan Circuitry

If your design contains boundary scan circuitry and existing internal scan circuitry, you must
integrate the boundary scan circuitry with the internal test circuitry. You must ensure proper
connection of the scan chains’ scan_in and scan_out ports to the TAP controller.

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Changing the System Mode (Running Rules

Checking)
Tessent Scan performs model flattening, learning analysis, rules checking, and scannability
checking when you try to exit the setup system mode.
“Understanding Common Tool Terminology and Concepts” on page 67 explains these
processes in detail. If you are finished with all the setup you need to perform, you can change
the system mode by entering the set_system_mode command as follows:

SETUP> set_system_mode analysis

If an error occurs during the rules checking process, the application remains in setup mode,
where you must correct the error. You can clearly identify and easily resolve the cause of many
errors. Other errors, such as those associated with proper clock definitions and test procedure
files, can be more complex. “Troubleshooting Rules Violations” in the Tessent Shell Reference
Manual discusses the procedure for debugging rules violations. You can also use
DFTVisualizer to visually investigate the causes of DRC violations. For more information, refer
to “DFTVisualizer” in the Tessent Shell User’s Manual.

Setting Up a Basic Scan Insertion Run

Tessent Scan’s default mode of operation is to replace all candidate non-scan cells with scan
cells and then stitch them into scan chains based on a number of internal criteria including
minimizing the number of clock domain crossings in the scan path, and also minimizing the
routing overhead.
The actual netlist processing is triggered by the insert_test_logic command. The default
operation of the insert_test_logic command is to perform scan substitution and stitching.

Setting Up for Wrapper Chain Identification

Tessent Scan provides the ability to improve performance and test coverage of hierarchical
designs by reducing the visibility of the internal core scan chains of the design submodules. The
only requirements for testing submodules at the top-level is that the logic inputs must be
controllable using scan chains (that is, must be driven by registers) and the logic outputs must
be observable through scan chains (that is, must be registered).
These requirements can be met using wrapper scan chains which are scan chains around the
periphery of the submodules that connect to each input and output (except user-defined clocks,
scan-related I/O pins, and user-excluded pins) of the submodules to be tested. Wrapper chains
comprising the wrapper cells reachable from a submodule’s inputs are treated as the Input type
wrapper chains and wrapper chains comprising the wrapper cells reachable from a submodule’s
outputs are treated as the Output type wrapper chains. The Input and Output wrapper chains are
used to provide proper test coverage of hierarchical designs during the INTEST and EXTEST
modes.

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In INTEST mode, all inputs to submodules are controllable using the Input wrapper scan chains
and all outputs are observable through the Output wrapper scan chains. This provides the ability
to independently generate a complete set of ATPG patterns for submodules.

In EXTEST mode, all outputs from submodules are controllable using the Output wrapper scan
chains and all inputs are observable through the Input wrapper scan chains. This provides the
ability to test/optimize the logic surrounding the submodules at the top-level without requiring
internal visibility into the submodules (that is, core scan chains are not needed).

Generating Wrapper Chains

Tessent Scan generates separate Input and Output wrapper chains based on the following
conditions:
• The scan cells comprising the wrapper scan chains are identified as the Input and Output
wrapper cells using the set_wrapper_chains and analyze_wrapper_cells commands and
are stitched into the separate Input and Output wrapper chains using the insert_test_logic
command.
o When you issue the analyze_wrapper_cells command, Tessent Scan performs the
wrapper cells analysis and identifies the Input and Output wrapper cells, which will
be stitched into the separate Input and Output wrapper chains.
o If the set_wrapper_chains command was not issued, the analyze_wrapper_cells
command will perform the default wrapper cells identification similar to issuing the
set_wrapper_chains command with no arguments.
o If the analyze_wrapper_cells command is issued in conjunction with the
“set_wrapper_chains -io_registration on” command or the set_registered_io
command, the dedicated wrapper cells added to the primary Inputs will be stitched
into the appropriate Input wrapper chains, while the dedicated wrapper cells added
to the primary Outputs will be stitched into the appropriate Output wrapper chains.
• When the set_registered_io command is issued and the analyze_wrapper_cells
command is not issued, the dedicated wrapper cells added to the primary I/Os are treated
as Core cells and are stitched into the appropriate Core scan chains.
o Dedicated wrapper cells are normally analyzed and inserted into scan chains when
you issue the insert_test_logic command.
o For backward compatibility, the report_wrapper_cells command will trigger the
dedicated wrapper cells analysis when the set_registered_io command was issued,
but the analyze_wrapper_cells command was not.
• If you issue the insert_test_logic command after the set_wrapper_chains command, but
have not issued the analyze_wrapper_cells command, you will get an error message
stating that the wrapper chains analysis is required prior to insertion.
Error: Wrapper cells information is out of date:

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use 'analyze_wrapper_cells' to re-analyze wrapper cells prior to

insertion.

• If you use the add_sub_chains command with the “-type input_wrapper” or “-type
output_wrapper” options to explicitly specify sub-chains to be of wrapper type
input_wrapper or output_wrapper, the sub-chains will be stitched into the appropriate
Input or Output wrapper chains only if the analyze_wrapper_chains command was
previously issued; otherwise they will be stitched into the Core chains and will be
controlled by the Core scan enable signals.
• The following commands can affect the I/O registration of dedicated wrapper cells and
the identification of shared wrapper cells:
o add_nonscan_instances, delete_nonscan_instances
o add_nonscan_models, delete_nonscan_models
o add_scan_instances, delete_scan_instances
o add_scan_models, delete_scan_models
If you use any of these commands in conjunction with the set_registered_io and
set_wrapper_chains commands, you will get the following warning message:
Warning: Wrapper cells information is out of date:
use 'analyze_wrapper_cells' to re-analyze wrapper cells prior to
insertion.

Constraining Input Partition Pins

Input partition pins are block input pins that you cannot directly control from chip-level primary
inputs.
Referring to Figure 2-4 and Figure 2-5 on pages 26 and 27, the input partition pins are those
inputs that come into Block A from Block B. Because these are uncontrollable inputs, you must
constrain them to an X value using the add_input_constraints command.

Masking Output Partition Pins

Output partition pins are block output pins that you cannot directly observe from chip-level
primary outputs.
Referring to Figure 2-4 and Figure 2-5 on pages 26 and 27, the output partition pins are those
outputs that go to Block B and Block C. Because these are unobservable outputs, you must
mask them with the add_output_masks command.

To ensure that masked primary outputs drive inactive values during the testing of other
partitions, you can specify that the primary outputs hold a 0 or 1 value during test mode. Special
cells called output hold-0 or output hold-1 wrapper cells serve this purpose. By default, the tool
uses regular output wrapper cells.

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Manually Specifying Control and Observe Points

You can manually specify which control and observe points to add.
You add the control and observe points using the add_control_points and add_observe_points
commands. Tessent Scan considers the test points you manually specify to be user-class test
points.

When using these commands, the tp_pin_pathname argument specifies the pin pathname of the
location where you want to add a control or observe point. If the location is to be a control point,
you specify the Control argument with the name of the model to insert (which you define with
add_cell_models or the cell_type attribute in the library description) and pin(s) to which you
want to connect the added gate.

If the location is to be an observe point, you must specify the primary output in which to
connect the observe point. You can also specify whether to add a scan cell at the control or
observe point.

The following locations in the design are possible candidates for placing control or observe test
points:

• Any site in the fanout cone of a declared clock (defined with the add_clocks command).
• The outputs of scanned latches or flip-flops.
• The internal gates of library cells. Only gates driving the top library boundary can have
test points.
• Notest points which are set using the add_notest_points command.
• The outputs of primitives that can be tri-state.
• The primary inputs for control or observation points.
• The primary outputs for observation points. A primary output driver which also fans out
to internal logic could have a control point added, if needed.
• No control points at unobservable sites.
• No observation points at uncontrollable sites.

Manually Including and Excluding Cells for Scan

Regardless of what type of scan you want to insert, you can manually specify instances or
models to either convert or not convert to scan. Tessent Scan uses lists of scan cell candidates
and non-scan cells when it selects which sequential elements to convert to scan. You can add
specific instances or models to either of these lists. When you manually specify instances or
models to be in these lists, these instances are called user-class instances. System-class

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instances are those Tessent Scan selects. The following subsections describe how you
accomplish this.

Handling Cells Without Scan Replacements

When Tessent Scan switches from setup to analysis mode, it issues warnings when it encounters
sequential elements that have no corresponding scan equivalents. Tessent Scan treats elements
without scan replacements as non-scan models and automatically adds them as system-class
elements to the non-scan model list.
You can display the non-scan model list using the report_nonscan_models or report_dft_check
command.

In many cases, a sequential element may not have a scan equivalent of the currently selected
scan type. For example, a cell may have an equivalent mux-DFF scan cell but not an equivalent
LSSD scan cell. If you set the scan type to LSSD, Tessent Scan places these models in the non-
scan model list. However, if you change the scan type to mux-DFF, Tessent Scan updates the
non-scan model list, in this case removing the models from the non-scan model list.

Specifying Non-Scan Components

Tessent Scan keeps a list of which components it must exclude from scan identification and
replacement.
To exclude particular instances from the scan identification process, you use the
add_nonscan_instances command.

For example, you can specify that I$155/I$117 and /I$155/I$37 are sequential instances you do
not want converted to scan cells by specifying:

SETUP> add_nonscan_instances /I$155/I$117 /I$155/I$37

Another method of eliminating some components from consideration for scan cell conversion is
to specify that certain models should not be converted to scan. To exclude all instances of a
particular model type, you can use the add_nonscan_models command.

For example, the following command would exclude all instances of the dff_3 and dff_4
components from scan cell conversion.

SETUP> add_nonscan_models dff_3 dff_4

Note
Tessent Scan automatically treats sequential models without scan equivalents as non-scan
models, adding them to the nonscan model list.

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Specifying Scan Components

After you decide which specific instances or models you do not want included in the scan
conversion process, you are ready to identify those sequential elements you do want converted
to scan. The instances you add to the scan instance list are called user-class instances.
To include particular instances in the scan identification process, use the add_scan_instances
command. This command lets you specify individual instances, hierarchical instances (for
which all lower-level instances are converted to scan), or control signals (for which all instances
controlled by the signals are converted to scan).

For example, the following command ensures the conversion of instances /I$145/I$116 and
/I$145/I$138 to scan cells when Tessent Scan inserts scan circuitry.

SETUP> add_scan_instances /I$145/I$116 /I$145/I$138

To include all instances of a particular model type for conversion to scan, use the
add_scan_models command. For example, the following command ensures the conversion of
all instances of the component models dff_1 and dff_2 to scan cells when Tessent Scan inserts
scan circuitry.

SETUP> add_scan_models dff_1 dff_2

Related Scan and Nonscan Commands

In addition to commands mentioned in previous sections, the following commands are useful
when working with scan and nonscan instances:
• delete_nonscan_instances — Deletes instances from the non-scan instance list.
• delete_nonscan_models — Deletes models from the non-scan model list.
• delete_scan_instances — Deletes instances from the scan instance list.
• delete_scan_models — Deletes models from the scan model list.
• report_nonscan_models — Displays the models in the non-scan instance list.
• report_sequential_instances — Displays information and testability data for sequential
instances.
• report_scan_models — Displays models in the scan model list.

Reporting Scannability Information

Scannability checking is a modified version of clock rules checking that determines which non-
scan sequential instances to consider for scan. You may want to examine information regarding
the scannability status of all the non-scan sequential instances in your design.

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To display this information, you use the report_dft_check command. This command displays
the results of scannability checking for the specified non-scan instances, for either the entire
design or the specified (potentially hierarchical) instance.

When you perform a report_dft_check command there is typically a large number of nonscan
instances displayed, as shown in the sample report in Figure 5-5.
Figure 5-5. Example Report from report_dft_check Command
SCANNABLE IDENTIFIED CLK0_7 /I_3 dff (156)
SCANNABLE IDENTIFIED CLK0_7 /I_2 dff (157)
SCANNABLE IDENTIFIED CLK0_7 /I_235 dff (158)
SCANNABLE IDENTIFIED CLK0_7 /I_237 dff (159)
SCANNABLE IDENTIFIED CLK0_7 /I_236 dff (160)
SCANNABLE IDENTIFIED Test-logic /I_265 dff (161)
Clock #1: F /I_265/clk
SCANNABLE IDENTIFIED Test-logic /I_295 dff (162)
Clock #1: F /I_295/clk
SCANNABLE IDENTIFIED Test-logic /I_298 dff (163)
Clock #1: F /I_298/clk
SCANNABLE IDENTIFIED Test-logic /I_296 dff (164)
Clock #1: F /I_296/clk
SCANNABLE IDENTIFIED Test-logic /I_268 dff (165)
Clock #1: F /I_268/clk
SCANNABLE IDENTIFIED CLK0_7 /I_4 dff (166)
SCANNABLE IDENTIFIED CLK0_7 /I_1 dff (167)
SCANNABLE DEFINED-NONSCAN Test-logic /I_266 dfscc (168) Stable-high
Clock #1: F /I_266/clk
SCANNABLE DEFINED-NONSCAN CLK0_7 /I_238 dfscc (169)
SCANNABLE DEFINED-NONSCAN Test-logic /I_297 dfscc (170) Stable-high
Clock #1: F /I_297/clk
SCANNABLE DEFINED-NONSCAN Test-logic /I_267 dfscc (171) Stable-high
Clock #1: F /I_267/clk

The fields at the end of each line in the nonscan instance report provide additional information
regarding the classification of a sequential instance. Using the instance /I_266 (highlighted in
maroon), the “Clock” statement indicates a problem with the clock input of the sequential
instance. In this case, when the tool does a trace back of the clock, the signal doesn’t trace back
to a defined clock. The message indicates that the signal traced connects to the clock input of
this non-scan instance, and doesn’t trace back to a primary input defined as a clock. If several
nodes are listed (similarly for “Reset” and “Set), it means that the line is connected to several
endpoints (sequential instances or primary inputs).

This “Clock # 1 F /I_266/clk” issue can be resolved by either defining the specified input as a
clock or allowing Tessent Scan to add a test clock for this instance.

• report_control_signals — Displays control signal information.

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• report_statistics — Displays a statistics report.

• report_sequential_instances — Displays information and testability data for sequential
instances.

Automatic Recognition of Existing Shift Registers

Tessent Scan automatically identifies the shift register structures in the input netlist and tries to
preserve their original connections (shift orders) when they are stitched into scan chains. In this
attempt, Tessent Scan converts only the first flip-flop of a shift register into a scan cell (if
originally a nonscan cell), and preserves the remaining flip-slops in the shift register as nonscan
(or replaces them with nonscan cells if originally scan cells).
This approach has several potential benefits such as the reduction of scan cells in the design,
and therefore fewer scan path muxes, and the better locality of scan path connections due to the
preservation of functional connections.

The following sections describe the process by which Tessent Scan identifies and converts scan
cells.

Identifying the Shift Registers

Tessent Scan identifies shift registers by performing a structural backward tracing in the
flattened model of the design.
The tracing starts from the data pin of the _dff primitive of a sequential cell and ends at the
output pin of the _dff primitive of another sequential cell. The tracing continues only on a single
sensitized path. The sensitization path of a combinational logic gate in the tracing path is
checked based on the state-stability values calculated by the tool during DRC rule checking.
When “-IGNORE_CONSTraint_values ON” is specified with the
set_shift_register_identification command, the state-stability values are ignored during tracing.
As a result, multiple-input gates (AND, OR, MUX, etc.) between sequential cells cannot be
traced during shift register identification. Single-input gates (BUF and INV), on the other hand,
can always be traced. This switch is Off, by default.

The identification occurs when switching to analysis mode, after DRC rule checking is
completed, as indicated in the following transcript.

// ----------------------------------------------------------------------
// Begin shift register identification process for 9971 sequential
// instances.
// ----------------------------------------------------------------------
// Number of shift register flops recorded for scan insertion: 3798
// (38.09%)
// Number of shift registers recorded for scan insertion: 696
// Longest shift register has 15 flops.
// Shortest shift register has 2 flops.
// Potential number of nonscan flops to be converted to scan cells: 696
// Potential number of scan cells to be converted to nonscan flops: 25

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Shift register identification assumes the following:

1. Flip-flops that constitute a shift register reside under the same hierarchical instance.
2. Flip-flops that constitute a shift register use the same clock signal.
3. Multiple clock edges are allowed in the shift register structure as long as no lockup cells
are required (no TE-LE transitions occur). When lockup cells are required (as in LE-TE
transitions), the tool breaks the shift register at this location.
4. Both nonscan and scan flip-flops are considered for identification. However, every
nonscan flip-flop should have a mapping scan flip-flop model in the ATPG library and
every scan flip-flop should have a mapping nonscan flip-flop model in the ATPG
library. In addition, a scan flip-flop should satisfy the following requirements:
o Its scan input pin is not functionally driven (either dangling or tied to a constant, or
looped back from Q/QB output).
o Its scan enable pin is not functionally driven, and is tied to a constant signal to
sensitize the data input pin of the sequential cell such that this input is preserved as
the shift path. The scan enable pin is not considered functionally driven if a global
scan enable pin (defined using set_scan_signals -SEN) is the driver.
5. Shift registers with multiple branches are identified such that each branch is a separate
shift register. The flip-flops on the trunk are included in one of the branches.
6. Shift registers with sequential feedback loops are identified such that the first cell of the
shift register is determined by the tool either randomly or based on the availability of the
scan cell in the loop.

Stitching Shift Register Cells Into Scan Chains

Nonscan flip-flops are replaced with their scan-equivalent flip-flops before stitching them into
scan chain structures. The identified shift register flip-flops are handled as follows.
1. The first flip-flop is replaced with the equivalent scan flip-flop. If the first flip-flop is
originally a scan flip-flop, it is preserved as is.
2. All remaining flip-flops are preserved as nonscan. If they are originally scan flip-flops,
they are converted into nonscan flip-flops.
After performing the scan chain insertion, the report_shift_registers command may list fewer
and shorter shift registers than what were originally identified by switching to DFT mode. The
following lists the main reasons why originally identified shift registers can be modified by the
scan chain stitching.

1. All scan cell candidates are sorted based on clock, edge, and hierarchy before stitching.
The sorting process preserves the cells of the shift registers in exactly the same order as
they are functionally connected. Sorted scan cell candidates are then distributed over

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scan chains. When placing a shift register into a scan chain, it may be cut to a desired
length to satisfy the specified/calculated scan chain length.
2. The first or the last cell in a shift register may be removed from the shift register if it is
positioned to be at the clock transition between the cells in the scan chain, or positioned
to be the first or last cell in the scan chain. These modifications are performed to move
shift register cells from the START and STOP declarations in a scanDEF file that will be
generated with a write_scan_order command. It is unlikely that the first or the last cell in
a shift register will be positioned to be the first or last cell in the scan chain, as the tool
when distributing scan cells to chains tries to avoid placing shift registers at the
beginning or tail of a scan chain. This is likely to happen only when the percentage of
shift registers are very high in the design compared to floating flops, and the tool doesn’t
have a choice to place any other potential scan cell other than a shift register at the
beginning/end of a chain.
The stitching of the flip-flops inside a shift register structure is skipped but the following
connections are performed:

1. The scan input pin of the first flip-flop.

2. The scan enable pin of the first flip-flop.
3. The scan output pin of the last flip-flop. The scan output pin on the last flip-flop will be
determined by checking the load on the Q and QN ports.

Reporting Shift Registers

You may want to report_shift_registers in the design.
To display this information, you use the report_shift_registers command. This command reports
the identified shift registers in the design after switching to analysis mode. The tool tries to
preserve the original connections inside the identified shift. For each identified shift register,
this command reports the following information:

• Length
• Hierarchical path where the shift register flip-flops reside
• First and last flip-flop instance name unless the -verbose switch is specified in which
case all flip-flops in the shift registers are reported
Scan cells can also be reported by the report_scan_cells command after the insert_test_logic
command is executed. If any shift registers are identified in the netlist, a column is added to the
report. The column contains a tool-assigned shift register ID number and a cell number that
indicates the order in which the flip-flops are originally connected in the shift register
structures.

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The Identification Process

Once you complete the proper setup, the identification process for any of the test structures is
done automatically when you switch to dft or analysis mode.
During the identification process, a number of messages may be issued about the identified
structures.

To identify the dedicated and shared wrapper cells, you can use the analyze_wrapper_cells
command.

Note
If you want to start the selection process anew each time, you must use the reset_state
command to clear the existing scan candidate list.

Reporting Identification Information

If you want a statistical report on all aspects of scan cell identification, you can enter the
report_statistics command.
The report_statistics command lists the total number of sequential instances, user-defined non-
scan instances, user-defined scan instances, system-identified scan instances, scannable
instances with test logic, and the scan instances in preexisting chains identified by the rules
checker.

The report_sequential_instances command displays information and testability data for

sequential instances.

Inserting Test Structures

Typically, after identifying the test structures you want, you perform some test synthesis setup
and then insert the structures into the design. The additional setup varies somewhat depending
on the type of test structure you select for insertion. The following logically-ordered subsections
discuss how to perform these tasks.

Setting Up for Internal Scan Insertion

As part of the internal scan insertion setup, you may want to set some scan chain parameters,
such as the scan input and output port names, and the enable and clock ports. If you specify a
port name that matches an existing port of the design, the existing port is used as the scan port.
If the specified port name does not exist, Tessent Scan creates a new port with the specified
name. If you use an existing, connected output port, Tessent Scan also inserts a mux at the
output to select data from either the scan chain or the design, depending on the value of the scan
enable signals.

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Naming Scan Input and Output Ports

Before Tessent Scan stitches the identified scan instances into a scan chain, it needs to know the
names of various pins, such as the scan input and scan output. If the pin names you specify are
existing pins, Tessent Scan will connect the scan circuitry to those pins. If the pin names you
specify do not exist, Tessent Scan adds these pins to the design. By default, Tessent Scan adds
pins for chainX scan ports and names them scan_inX and scan_outX (where X represents the
number of the chain).
To give scan ports specific names (other than those created by default), you can use the
add_scan_pins command.

You must specify the scan chain name, the scan input pin, and the scan output pin. Additionally,
you may specify the name of the scan chain clock. For existing pins, you can specify top module
pins or pins of lower level instances.

After the scan cells are partitioned and grouped into potential scan chains (before scan chain
insertion occurs) Tessent Scan considers some conditions in assigning scan pins to scan chains:

1. Whether the output of the scan candidate is directly connected to a declared output pin.
If yes, then the scan input and output pins are assigned to the scan chain containing the
scan cell candidate.
2. Any scan chains not assigned to scan input/output pins using conditions 1 and 2 are
assigned based on the order in which you declared the scan input/output pins using the
add_scan_pins command.
If a fixed-order file is specified along with the -Fixed option in the insert_test_logic command,
conditions 1 and 2 are ignored and the chain_id in the fixed-order file is then sorted in
increasing order. The chain with the smallest chain_id receives the first specified scan
input/output pins. The chain with the second smallest chain_id receives the second specified
scan input/output pins, and so on. If you did not specify enough scan input/output pins for all
scan chains, then Tessent Scan creates new scan input/output pins for the remaining scan chains.

For information on the format of the fixed-order file, refer to the insert_test_logic description in
the Tessent Shell Reference Manual.

• delete_scan_pins — Deletes scan chain inputs, outputs, and clock names.

• report_scan_pins — Displays scan chain inputs, outputs, and clock names.
• set_scan_pins — Specifies the index or bus naming conventions for scan input and
output pins.

Naming the Enable and Clock Ports

The enable and clock parameters include the pin names of the scan enables, test enable, test
clock, new scan clock, scan master clock, and scan slave clock.

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Additionally, you can specify the names of the set and reset ports and the RAM write and read
ports in which you want to add test logic, along with the type of test logic to use. You do this
using the set_scan_signals command. If you do not specify this command, the default pin names
are scan_en, scan_en_in, scan_en_out, test_en, test_clk, scan_clk, scan_mclk, scan_sclk,
scan_set, scan_reset, write_clk, and read_clk, respectively. If you want to specify the names of
existing pins, you can specify top module pins or dangling pins of lower level modules.

Note
If Tessent Scan adds more than one test clock, it names the first test clock the specified or
default <name> and names subsequent test clocks based on this name plus a unique
number.

The -Muxed and -Disabled switches specify whether Tessent Scan uses an AND gate or MUX
gate when performing the gating. If you specify the -Disabled option, then for gating purposes
Tessent Scan ANDs the test enable signal with the set and reset to disable these inputs of
flip-flops. If you specify the -Muxed option, then for muxing purposes Tessent Scan uses any
set and reset pins defined as clocks to multiplex with the original signal. You can specify the
-Muxed and -Disabled switches for individual pins by successively issuing the set_scan_signals
command.

If Tessent Scan writes out a test procedure file, it places the scan enable at 1 (0) if you specify
-Active High (Low).

Note
If the test enable and scan enable have different active values, you must specify their
active values using different commands. Set the test enable active value using the
set_scan_signals command. Set the scan enable active value using the set_scan_enable
command.

After setting up for internal scan insertion, refer to “Running the Insertion Process” on page 144
to complete insertion of the internal scan circuitry.

Attaching Head and Tail Registers to the Scan Chain

A head register is a non-scan DFF connected at the beginning of a scan chain. A tail register is
a scan DFF connected at the end of the scan chain. You can have Tessent Scan identify certain
cells as the head and tail registers for a scan chain of mux-DFF scan type, and connect them at
the beginning and the end of the chain. The head and tail cells are not inserted by Tessent Scan;
they are only identified in the original design.
During test logic insertion, Tessent Scan attaches the non-scan head register’s output to the
beginning of the scan chain, performs scan replacement on the tail register, and then attaches the
scan tail register’s input to the end of the scan chain. If there is no scan replacement in the
ATPG library for the tail register, a MUX is added to include the tail DFF into the scan chain.

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Note
No design rule checks are performed from the scan_in pin to the output of the head
register and from the output of the tail register to the scan_out pin. You are responsible
for making those paths transparent for scan shifting.

Note
Tessent Scan does not determine the associated top-level pins that are required to be
identified for the add_scan_chains command. You are responsible for adding this
information to the dofile that Tessent Scan creates using the write_atpg_setup command.
You must also provide the pin constraints that cause the correct behavior of the head and
tail registers.

You can specify head and tail registration of a scan chain via the add_scan_pins command. You
need to specify the head register output pin as the scan input pin, and the tail register input pin
as the scan output pin, along with the -Registered switch.

Running the Insertion Process

The insert_test_logic command inserts all of the previously identified test structures into the
design. This includes internal scan (sequential and scan-sequential types), wrapper chain, test
logic, and test points.
When you issue the insert_test_logic command for scan insertion (assuming appropriate prior
setup), the tool converts all identified scannable memory elements to scan elements and then
stitches them into one or more scan chains. If you select wrapper chains for insertion, the tool
converts the non-scan cells identified for wrapper chains to wrapper cells and stitches them into
scan chains separate from internal scan chains.

The scan circuitry insertion process may differ depending on whether you insert scan cells and
connect them up front, or insert and connect them after layout data is available. the tool allows
you to insert scan using both methods.

The insert_test_logic command has a number of different options, most of which apply
primarily to internal scan insertion.

• If you are using specific cell ordering, you can specify a filename of user-identified
instances (in either a fixed or random order) for the stitching order.
• The -Max_length option lets you specify a maximum length to the chains.
• The -NOlimit switch allows an unlimited chain length.
• The -NUmber option lets you specify the number of scan chains for the design.
• The -Clock switch lets you choose whether to merge two or more clocks on a single
chain.

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• The -Edge switch lets you choose whether to merge stable high clocks with stable low
clocks on chains.
The subsection that follows, “Merging Scan Chains with Different Shift Clocks“,
discusses some of the issues surrounding merging chains with different clocks.
• The -COnnect option lets you specify whether to connect the scan cells and scan-
specific pins (scan_in, scan_enable, scan_clock, etc.) to the scan chain (which is the
default mode), or just replace the scan candidates with scan equivalent cells. If you want
to use layout data, you should replace scan cells (using the -connect off switch), perform
layout, obtain a placement order file, and then connect the chain in the appropriate order
(using the -filename filename -fixed options). This option is affected by the settings in
the set_test_logic command. The other options to the -COnnect switch specify how to
handle the input/output scan pins when not stitching the scan cells into a chain.
• The -Scan, -Test_point, and -Ram switches let you turn scan insertion, test point
insertion and RAM gating on or off.
• The -Verilog switch causes the tool to insert buffer instances, rather than use the
“assign” statement, for scan output pins that also fan out as functional outputs.
If you do not specify any options, the tool stitches the identified instances into default scan
chain configurations.

Note
Because the design is significantly changed by the action of this command, the original
flattened, gate-level simulation model created when you entered the analysis system
mode is deleted.

Merging Scan Chains with Different Shift Clocks

You can merge scan cells with different shift clocks into the same scan chain.
You do this using the “-clock merge” option of the insert_test_logic command. To avoid
synchronization problems, the tool places the scan cells using the same shift clock adjacent to
each other in the chain and then inserts lockup cells between the scan cells that use different
shift clocks. Lockup cells are inserted on the scan path to synchronize clock domains in adjacent
scan cells and do not interfere with the functional operation of the circuit.

Use this procedure to merge scan cells with different shift clocks.

Procedure
1. Define latch and inverter library models to use for lockup cells. For example:
add_cell_models dlat1a -type dlat enable data -active high|Low
add_cell_models inv01 -type inv

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The cell type for the latch must be either a DLAT or DFF and the active edge specified
must reflect the overall inversion of the clock pin inside the model.
The inverter is used along with the lockup cell model, when necessary, for correct
lockup cell insertion. The inverter model can also be defined with cell_type attribute in
the model definition.
For more information, see the add_cell_models description in the Tessent Shell
Reference Manual.
2. Set the tool to insert lockup cells. For example:
set_lockup_cell on -type dlat

You can also use a DFF model for a lockup cell, in which case you need to define a DFF
model using the Add Cell Model and use “-type dff” switch for the set_lockup_cell
command.
For more information, see the set_lockup_cell description in the Tessent Shell Reference
Manual.

Note
You can also specify the exact lockup cell locations in scan chains using a cell order file
that specifies the stitching order of scan cells in scan chains. For more information, see
the insert_test_logic description in the Tessent Shell Reference Manual.

Figure 5-6 illustrates lockup cell insertion. The extra inverter on the clock line of the
lockup cell provides a half a cycle delay to ensure synchronization of the clock domains.

Figure 5-6. Lockup Cell Insertion

Before After

d o d o d o
d o d o si si
si si SC LL SC
SC SC clka clk clk clk
clka clk clk

clkb clkb

Examples
The following example defines a group of clocks, identifies a latch model and inverter model to
use for lockup cells, enables lockup cell insertion, and performs the insertions. The -Clock

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Merge option combines the scan cells associated with each of the specified clock groups into a
scan chain when the test logic is inserted.
add_clocks 0 clk1 clk2 clk3
add_cell_models dff04 -type dff clk data
add_cell_models inv -type inv
set_lockup_cell on -type dff
insert_test_logic -scan on -clock merge

The tool creates one scan chain and inserts lockup cells between the clock domains clk1, clk2,
and clk3.
• report_dft_check — Displays and writes the scannability check status for all non-scan
instances.
• report_scan_cells — Displays a list of all scan cells.
• report_scan_chains — Displays scan chain information.
• report_scan_groups — Displays scan chain group information.

Saving the New Design and ATPG Setup

After test structure insertion, the tool releases the current flattened model and has a new
hierarchical netlist in memory. Thus, you should save this new version of your design.
Additionally, you should save any design information that the ATPG process might need.

Writing the Netlist

You can save the netlist for your new design by issuing the write_design command.

Issues with the New Version of the Netlist

The following lists some important issues concerning netlist writing:
• Tessent Scan is not intended for use as a robust netlist translation tool. Thus, you should
always write out the netlist in the same format in which you read the original design.
• If a design contains only one instantiation of a module, and the tool modifies the
instance by adding test structures, the instantiation retains the original module name.
• When the tool identically modifies two or more instances of the same module, all
modified instances retain the original module name. This generally occurs for scan
designs.
• If a design contains multiple instantiations of a module, and the tool modifies them
differently, the tool derives new names for each instance based on the original module
name.

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• Tessent Scan assigns “net” as the prefix for new net names and “uu” as the prefix for
new instance names. It then compares new names with existing names (in a case-
insensitive manner) to check for naming conflicts. If it encounters naming conflicts, it
changes the new name by appending an index number.

Writing the Test Procedure File and Dofile for ATPG

If you plan to use the ATPG tool, you can use Tessent Scan to create a dofile (for setting up the
scan information) and a test procedure file (for operating the inserted scan circuitry).
For information on the new test procedure file format, see “Test Procedure File” in the Tessent
Shell User’s Manual.

To create test procedure files, issue the write_atpg_setup command.

Running Rules Checking on the New Design

You can verify the correctness of the added test circuitry by running the full set of rules checks
on the new design. To do this, return to setup mode after scan insertion, delete the circuit setup,
run the dofile produced for ATPG, and then return to analysis mode. This enables rules
checking on the added scan circuitry to ensure it operates properly before you go to the ATPG
process.
For example, if the tool adds a single scan chain and writes out an ATPG setup file named
scan_design.dofile, enter something like the following:

> set_system_mode setup

> delete_clocks -all
> dofile scan_design.dofile
> set_system_mode analysis

Exiting Tessent Scan

When you finish the Tessent Scan session, enter the exit command.

Inserting Scan Block-by-Block

Scan insertion is “block-by-block” when the tool first inserts scan into lower-level hierarchical
blocks and then connects them together at a higher level of hierarchy.
For example, Figure 5-7 shows a module (Top) with three submodules (A, B, and C).

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Figure 5-7. Hierarchical Design Prior to Scan

top_i Top top_o

a_i b_i c_i

A B C
a_o b_o c_o

Using block-by-block scan insertion, the tool inserts scan (referred to as “sub-chains”) into
blocks A, B, and C, prior to insertion in the Top module. When A, B, and C already contain
scan, inserting scan into the Top module is equivalent to inserting any scan necessary at the top
level, and then connecting the existing scan circuitry in A, B, and C at the top level.

Verilog Flow Example

The following shows the basic procedure for adding scan circuitry block-by-block, as well as
the input and results of each step.
1. Insert scan into block A.
a. Invoke Tessent Scan on a.hdl.
Assume that the module interface is:
A(a_i, a_o)

b. Insert scan.
Set up the circuit, run rules checking, insert the desired scan circuitry.
c. Write out scan-inserted netlist.
Write the scan-inserted netlist to a new filename, such as a_scan.hdl. The new
module interface may differ, for example:
A(a_i, a_o, sc_i, sc_o, sc_en)

d. Write out the subchain dofile.

Use the write_subchain_setup command to write a dofile called a.do for the scan-
inserted version of A. The write_subchain_setup command uses the add_sub_chains
command to specify the scan circuitry in the individual module of the design.
Assuming that you use the mux-DFF scan style and the design block contains 7
sequential elements converted to scan, the subchain setup dofile could appear as
follows:

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DFT> add_sub_chains /user/jdoe/designs/design1/A chain1 sc_i sc_o

7 mux_scan sc_en

e. Exit the tool.

2. Insert scan into block B.
Follow the same procedure as in block A.
3. Insert scan into block C.
Follow the same procedure as in blocks A and B.
4. Concatenate the individual scan-inserted netlists into one file.
$ cat top.hdl a_scan.hdl b_scan.hdl c_scan.hdl > all.hdl

5. Stitch together the chains in blocks A, B, and C.

a. Invoke Tessent Scan on all.hdl.
Assume at this point that the module interface is:
TOP(top_i, top_o)
A(a_i, a_o, sc_i, sc_o, sc_en)
B(b_i, b_o, sc_i, sc_o, sc_en)
C(c_i, c_o, sc_i, sc_o, sc_en)

b. Run each of the scan subchain dofiles (a.do, b.do, c.do).

c. Insert the desired scan circuitry into the all.hdl design.
6. Write out the netlist and exit.
At this point the module interface is:
TOP(top_i, top_o, sc_i, sc_o, sc_en)
A(a_i, a_o, sc_i, sc_o, sc_en)
B(b_i, b_o, sc_i, sc_o, sc_en)
C(c_i, c_o, sc_i, sc_o, sc_en)

Figure 5-8 shows a schematic view of the design with scan connected in the Top
module.

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Figure 5-8. Final Scan-Inserted Design

all.hdl
TOP
top_i Combinational Logic

b_i c_i
a_i
sc_out sc_out sc_out
sc_in A sc_in B sc_in C sc_out
a_o sc_en b_o sc_en c_o
sc_en

Combinational Logic top_o

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 Chapter 6
Core Mapping for Top-Level ATPG

This chapter describes core mapping for ATPG functionality in the Tessent Shell® tool.
Core mapping for ATPG provides an ease-of-use flow by automating the mapping of DFT
information from one level to the next so that you can generate patterns at a higher level.

You can use this functionality in flows in which DFT is implemented bottom-up but pattern
generation is run at a higher level. You can use this functionality for modular EDT flows (in
which EDT and scan is implemented within a core but pattern generation is run at the next
higher level) to map EDT, scan, clock, and procedures from the core level to the level at which
pattern generation is to be run; this can be at a higher level core or at the chip level.

Core mapping for ATPG extends the core description functionality and use model developed for
pattern retargeting to top-level ATPG. Implementing core mapping for ATPG provides you
with the following benefits:

• Clean transfer of core information to the top-level.

• A simple and less error-prone use model by relying on Tessent core description files for
the transfer of information. This is consistent with the pattern retargeting use model.
• Support for all combinations of uncompressed and compressed scan chains.
• Mapping or verification of procedures and clocks.

Tools and Licensing

Core mapping for ATPG requires a Tessent TestKompress or Tessent FastScan license. A
TestKompress license is specifically needed if EDT logic is present at the top level of the design
or in any of the cores.

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Core Mapping for ATPG Process Overview

This section provides an overview of the core mapping for ATPG process. You can use core
mapping to transfer the core-level information to the top level of the design and generate
patterns for the whole chip. The tool uses the Tessent core description (TCD) file to transfer
core-level information to the top level of the design.
An example of the core mapping process is presented in “Core Mapping Examples” on
page 162.

Tessent Core Description File

The Tessent core description (TCD) file contains the information the tool needs to map the core
to the top level, including information such as a description of the EDT hardware and scan
chains.

Test Procedure Retargeting for Core Mapping

The automatic mapping (transfer) of test procedure information from the core level to the top
level of the design occurs when the tool transitions to analysis mode.

load_unload and shift Procedures

When cores are mapped to the chip level, chip-level load_unload and shift test procedures are
also needed. If chip-level load_unload and shift test procedures do not exist, the tool
automatically generates them based on the information in the core-level TCD files. This step
maps the core load_unload and shift procedures from the core level, merges all of the core
load_unload and shift procedures (if there is more than one core), and then merges them with
the top-level test procedures (if there are any and they do not already have the needed events to
drive the core instances).

Any timeplates in the core-level load_unload and shift procedures are updated when they are
mapped to the top level based on the following rules:
• If the timeplate does not have any pulse statements defined, the tool uses an offset and
width of 25% of the period as the pulse template. If the template pulse occurs before
measure_po time, the tool moves the pulse window by another 25%.
• If non-pulse-always clock pulses are defined in the timeplate, the specification of the
clock that is pulsed as the first one is used as a pulse timeplate.
• If only pulse-always clock pulses are defined in the timeplate, the tool uses the first one
as a pulse template.
• If the template clock has multiple pulses, the tool only uses the first pulse specification.

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Clocks and Pin Constraints

By default, the tool maps all clocks and pin constraints that are specified at the core-level to the
top. Clock types are mapped based on the core-level specification. Note, the tool maps any
permanently tied constraints (CT) in those files as constraints that can be overridden (C).
Even though clocks and pin constraints are automatically mapped to the top level by default,
you can still define them at the top, or drive the core-level input constraints through logic such
as a TAP.
Any clocks that are mapped to the top are removed when the tool exits analysis mode.
test_setup and test_end Procedures
You will typically need to provide top-level test_setup procedures and, occasionally, test_end
procedures to configure instruments such as PLLs, OCCs, EDT IPs, channel multiplexer access,
and so on in preparation for scan test. These instruments may be at the top level and/or within
the cores. The test_setup and test_end procedures can consist of a cyclized sequence of events,
and/or IJTAG iCall statements.

If any of the instruments used for scan test are inside a core, a test_setup and/or test_end
procedure at the core level must have been used to initialize them. If IJTAG was used within a
core-level test_setup or test_end procedure, this information is retained in the TCD file. When
an instance of this core is added for mapping, the iCalls that were used at the core level are
automatically mapped to the top level and added to the top-level test_setup or test_end
procedure. In other words, if you use IJTAG to set up the core at the core level, the tool
automatically maps those sequences to the top-level test_setup and test_end procedures without
you doing anything. Of course, the top-level test_setup and test_end procedures may have
additional events to configure top-level instruments.

In order for the tool to perform automatic IJTAG mapping, you must have extracted the
chip-level ICL using the extract_icl command. You must also have sourced the core-level PDL.

Several R DRCs validate the presence of both ICL and PDL. Note that the tool does not store
ICL and PDL in the core-level TCD file; it only stores references to their usage in the test_setup
or test_end procedure.

Note
If you have not previously extracted ICL in a separate step, you can still execute the
normal core mapping flow with one change. After entering the patterns -scan context, you
need to read ICL for each of the cores, extract the ICL using the extract_icl command,
read the PDL file for each module, and then proceed with the rest of the core mapping
flow. For specific instructions on performing ICL extraction, refer to “Step-by-Step ICL
Extraction Procedure” in the Tessent IJTAG User’s Manual.

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Note
You can disable load_unload and shift procedure retargeting by executing the
“set_procedure_retargeting_options -scan off” command. If test procedure retargeting is
disabled and chip-level load_unload and shift test procedures are missing, the tool
generates an error.

You can disable the automated mapping of core-level iCalls stored in the TCD file to the
top level by executing the “set_procedure_retargeting_options -ijtag off” command. (This
will not have any impact on iCalls you explicitly added into the top-level setup
procedures either explicitly or indirectly using the set_test_setup_icall command.) If
IJTAG retargeting is disabled, you must provide the needed test_setup and test_end
procedures.

For more information on using IJTAG to automate test_setup and test_end procedure creation,
see “IJTAG and ATPG in Tessent Shell” in the Tessent IJTAG User’s Manual.

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Core Mapping Process

In addition to scan logic within the cores, you may also have top-level scan logic that will
operate together with the core logic in the current mode when DRCs and ATPG are run. The
standard core mapping process enables you to add the top-level scan logic using standard
commands such as add_scan_chains and add_scan_groups. The tool will merge the core-level
load_unload and shift procedures with the top-level load_unload and shift procedures if you
have read them in.

1. Generate a Tessent core description (TCD) file for each core that you want to map to the
top. This process is illustrated in Figure 6-1.

Note
This is the same process that you use to run DRC and ATPG at the core level, except that
in this step you export the core description to the next higher level using the
write_core_description command.

Figure 6-1. Generation of Core Description Files

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2. Map the core-level TCD files to the top level. This process is illustrated in Figure 6-2.

Figure 6-2. Core Mapping to Top Level

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Alternative Core Mapping Process

If you want to verify your top-level scan logic independently, you can use an alternative
mapping process. The alternative mapping process requires that you create, validate, and then
add the top-level scan logic as a core instance using the “add_core_instances -current_design”
command.

1. Generate a Tessent core description (TCD) file for each core that you want to map to the
top. This process is illustrated in Figure 6-1.
2. Generate a TCD file for the scan logic present at the top level which excludes the scan
logic in the cores that will be mapped and merged later. This process is illustrated in
Figure 6-3. This is identical to what was done at the core level in Figure 6-1 except that
this mode only defines the top-level logic in the design with the exception of the logic in
the core that will be mapped in the next step.
The TCD file you generate in this step only describes the top-level scan logic (it is not
the TCD file that describes everything including the top-level and mapped core
information).

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Figure 6-3. Generation of Top-Level TCD File for Top-Level Logic

3. Map the core-level TCD files to the top level.

This process is illustrated in Figure 6-4. In this alternative flow, if scan logic exists at the
top level, you will have two different views (modes) of the top level: a view that only
includes the top-level scan logic and the procedures that operate that logic (if there is
any), and the merged top-level view created after core mapping which, in addition to the
top-level scan logic and procedures, also includes all of the core-level scan logic
mapped to the top. If there are two views of the top level, you must specify a different

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top-level mode name in either this step or the next step, using the set_current_mode
command.

Figure 6-4. Alternate Process for Core Mapping to Top Level

This flow matches the process shown in Figure 6-2, except that in addition to adding
core instances for the lower-level cores, you use the “add_core_instances
-current_design” command to define the top-level scan logic that was stored in the TCD
in step 2.

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Core Mapping Examples

Core Mapping Examples

The following two examples demonstrate the core mapping use model and the alternative core
mapping use model.
Both of these examples are based on the design shown in Figure 6-5. The design has three
cores: one instance of the piccpu_1 core containing both compressed and uncompressed scan
chains, one instance of the piccpu_2 core containing only compressed chains, and one instance
of the small_core core containing only uncompressed scan chains. TOP is the top design where
both compressed and uncompressed chains exist.

Figure 6-5. Mapping of Cores to Chip Level

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Core Mapping Example

As shown in this example, after you have generated a TCD file for each of the cores in your
design, you map the cores to the chip level using those TCD files, add any additional scan logic,
and then generate patterns for the entire design.

# Set the proper context for core mapping and subsequent ATPG
set_context pattern -scan

# Read cell library (library file)

read_cell_library technology.tcelllib

# Read the top-level netlist and all core-level netlists

read_verilog generated_1_edt_top_gate.vg generated_2_edt_top_gate.vg \
generated_top_edt_top_gate.vg

# Specify the top level of design for all subsequent commands and set mode
set_current_design
set_current_mode top_design_mode

# Read all core description files

read_core_descriptions piccpu_1.tcd
read_core_descriptions piccpu_2.tcd
read_core_descriptions small_core.tcd

# Bind core descriptions to cores

add_core_instances -instance core1_inst -core piccpu_1
add_core_instances -instance core2_inst -core piccpu_2
add_core_instances -instance core3_inst -core small_core

# Specify top-level compressed chains and EDT

dofile generated_top_edt.dofile

# Specify top-level uncompressed chains

add_scan_chains top_chain_1 grp1 top_scan_in_3 top_scan_out_3
add_scan_chains top_chain_2 grp1 top_scan_in_4 top_scan_out_4

# Report instance bindings

report_core_instances

# Change to analysis mode

set_system_mode analysis

# Create patterns
create_patterns

# Write patterns
write_patterns top_patts.stil -stil -replace

# Report procedures used to map the core to the top level (optional)
report_procedures

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Core Mapping Examples

Alternative Core Mapping Example

As shown in this alternative example flow, after you have generated a TCD file for each of the
cores in your design and for the top-level scan logic, you map the cores to the chip level using
those TCD files, and then generate patterns for the entire design.

Notice that in this flow you must add the top-level logic as a core instance using the
“add_core_instances -current_design” command. You must also specify a top-level mode name
using set_current_mode if you did not specify one during TCD file generation; the mode must
have a different name than the mode in the TCD files to avoid overwriting the first one.

The differences between this flow and the standard core mapping flow are shown in red font.

# Set the proper context for core mapping and subsequent ATPG
set_context pattern -scan

# Read cell library (library file)

read_cell_library technology.tcelllib

# Read the top-level netlist and all core-level netlists

read_verilog generated_1_edt_top_gate.vg generated_2_edt_top_gate.vg \
generated_top_edt_top_gate.vg

# Specify the top level of design for all subsequent commands and set mode
set_current_design
set_current_mode top_design_mode

# Read all core description files

read_core_descriptions core1.tcd
read_core_descriptions core2.tcd
read_core_descriptions core3.tcd
read_core_descriptions top_only.tcd

# Bind core descriptions to cores

add_core_instances -instance core1_inst -core piccpu_1
add_core_instances -instance core2_inst -core piccpu_2
add_core_instances -instance core3_inst -core small_core
add_core_instances -current_design -core my_design

# Report instance bindings

report_core_instances

# Change to analysis mode

set_system_mode analysis

# Create patterns
create_patterns

# Write patterns
write_patterns top_patts.stil -stil -replace

# Report procedures used to map the core to the top level (optional)
report_procedures

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Core Mapping Examples

Core Mapping Example With IJTAG

If you used IJTAG at the core level, by default, the tool will try to map any core-level iCalls to
the chip level as described in section “test_setup and test_end Procedures.” As shown in this
example, after you have generated a TCD file for each of the cores in your design, you need to
extract the top-level ICL and source the necessary PDL. When you have done this, you can map
the cores to the chip level using the previously generated TCD files, add any additional scan
logic, and then generate patterns for the entire design.

The differences between this flow and the standard core mapping flow are shown in red font.

# Set the proper context for core mapping and subsequent ATPG
set_context pattern -scan

# Read cell library (library file)

read_cell_library technology.tcelllib

# Read the top-level netlist and all core-level netlists

read_verilog generated_1_edt_top_gate.vg generated_2_edt_top_gate.vg \
generated_top_edt_top_gate.vg

# Read core-level ICL. ICL may be automatically loaded when the

# set_current_design command is issued as described in section
# “Top-Down and Bottom-Up ICL Extraction Flows” in the Tessent IJTAG
# User’s Manual
read_icl piccpu_1.icl
read_icl piccpu_2.icl
read_icl small_core.icl

# Specify the top level of design for all subsequent commands and set mode
set_current_design
set_current_mode top_design_mode

# Extract top-level ICL and write the extracted ICL for later usage
extract_icl
write_icl -output_file top_design.icl

# Source core-level PDL

source generated_1_edt.pdl
source generated_2_edt.pdl
source generated_3_edt.pdl

# Read all core description files

read_core_descriptions piccpu_1.tcd
read_core_descriptions piccpu_2.tcd
read_core_descriptions small_core.tcd

# Bind core descriptions to cores

add_core_instances -instance core1_inst -core piccpu_1
add_core_instances -instance core2_inst -core piccpu_2
add_core_instances -instance core3_inst -core small_core

# Specify top-level compressed chains and EDT

dofile generated_top_edt.dofile

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Limitations

# Specify top-level uncompressed chains

add_scan_chains top_chain_1 grp1 top_scan_in_3 top_scan_out_3
add_scan_chains top_chain_2 grp1 top_scan_in_4 top_scan_out_4

# Report instance bindings

report_core_instances

# Change to analysis mode

set_system_mode analysis

# Create patterns
create_patterns

# Write patterns
write_patterns top_patts.stil -stil -replace

# Report procedures used to map the core to the top level (optional)
report_procedures

Limitations
The limitations of the Core Mapping for ATPG functionality are listed here.
• Core-level scan pins, whether compressed or uncompressed, must connect to the top
level through optional pipeline stages. Uncompressed chains from different cores cannot
be concatenated together before being connected to the top level.

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 Chapter 7
Generating Test Patterns

The following figure shows the process for generating test patterns for your design.

Figure 7-1. Test Generation Procedure

ATPG Basic Tool Flow

Performing Basic Operations
Setting Up Design and Tool Behavior
Checking Rules and Debugging Rules Violations
Running Good/Fault Simulation on Existing Patterns
Insert Internal Running Random Pattern Simulation
Scan Circuitry
(Tessent Scan) Setting Up the Fault Information for ATPG
Performing ATPG
Generate/Verify
Test Patterns Creating an IDDQ Test Set
(ATPG tool)
Net Pair Identification with Calibre for Bridge
Fault Test Patterns
Hand Off Creating a Delay Test Set
to Vendor
Creating a Transition Delay Test Set
Creating a Path Delay Test Set
At-Speed Test Using Named Capture Procedures
Generating Patterns for a Boundary Scan Circuit
Using the MacroTest Capability
Verifying Test Patterns

This section discusses each of the tasks outlined in Figure 7-1. You use the ATPG tool (and
possibly ModelSim, depending on your test strategy) to perform these tasks.

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ATPG Basic Tool Flow

The following figure shows the basic process flow for the ATPG tool.

Figure 7-2. Overview of ATPG Tool Usage

From Test Invoke Tessent Shell

Synthesis set_context patterns -scan

Load Design and Library Synthesized

Library
Netlist

Setup Mode Logfile

Dofile

Design
Flattened? Y

N
create_flat_model

Test Learn Circuitry

Procedure Perform DRC
File

Pass
Checks? N

Analysis Mode

Fault
Add/Read Fault
Fault List File
File

Read/Simulate Create
Patterns Patterns
Patterns

Save
Patterns Patterns

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ATPG Basic Tool Flow

The following list describes the basic process for using the ATPG tool:

1. Invoke Tessent Shell using the “tessent -shell” command. Set the context to
“patterns -scan” using the set_context command, which allows you to access ATPG
functionality.
2. The ATPG tool requires a structural (gate-level) design netlist and a DFT library, which
you accomplish with the read_cell_library and read_verilog commands, respectively.
“ATPG Tool Inputs and Outputs” on page 170 describes which netlist formats you can
use with the ATPG tool. Every element in the netlist must have an equivalent
description in the specified DFT library. The “Design Library” section in the Tessent
Cell Library Manual gives information on the DFT library. The tool reads in the library
and the netlist, parsing and checking each.
3. After reading the library and netlist, the tool goes into setup mode. Within setup mode,
you perform several tasks, using commands either interactively or through the use of a
dofile. You can set up information about the design and the design’s scan circuitry.
“Setting Up Design and Tool Behavior” on page 177 documents this setup procedure.
Within setup mode, you can also specify information that influences simulation model
creation during the design flattening phase.
4. After performing all the desired setup, you can exit setup mode, which triggers a number
of operations. If this is the first attempt to exit setup mode, the tool creates a flattened
design model. This model may already exist if a previous attempt to exit setup mode
failed or you used the create_flat_model command. “Model Flattening” on page 75
provides more details about design flattening.
5. Next, the tool performs extensive learning analysis on this model. “Learning Analysis”
on page 80 explains learning analysis in more detail.
6. Once the tool creates a flattened model and learns its behavior, it begins design rules
checking. The “Design Rule Checking” section in the Tessent Shell Reference Manual
gives a full discussion of the design rules.
7. Once the design passes rules checking, the tool enters analysis mode, where you can
perform simulation on a pattern set for the design. For more information, refer to
“Good-Machine Simulation” on page 192 and “Fault Simulation” on page 190.
8. At this point, you may want to create patterns. You can also perform some additional
setup steps, such as adding the fault list. “Setting Up the Fault Information for ATPG”
on page 193 details this procedure. You can then run ATPG on the fault list. During the
ATPG run, the tool also performs fault simulation to verify that the generated patterns
detect the targeted faults.
In either case (full or partial scan), you can run ATPG under different constraints, or
augment the test vector set with additional test patterns, to achieve higher test coverage.
“Performing ATPG” on page 197 covers this subject.

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ATPG Basic Tool Flow

After generating a test set with the ATPG tool, you should apply timing information to
the patterns and verify the design and patterns before handing them off to the vendor.
“Verifying Test Patterns” on page 302 documents this operation.

ATPG Tool Inputs and Outputs

The following figure shows the inputs and outputs of the ATPG tool.

Figure 7-3. ATPG Tool Inputs and Outputs

Test
Design Procedure ATPG
Netlist File Library

ATPG Tool Fault

Test List
Patterns

ATPG
Info.
Files

The ATPG tool uses the following inputs:

• Design — The supported design data format is gate-level Verilog. Other inputs also
include 1) a cell model from the design library and 2) a previously-saved, flattened
model.
• Test Procedure File — This file defines the operation of the scan circuitry in your
design. You can generate this file by hand, or Tessent Scan can create this file
automatically when you issue the command write_atpg_setup.
• Library — The design library contains descriptions of all the cells used in the design.
The tool uses the library to translate the design data into a flat, gate-level simulation
model for use by the fault simulator and test generator.
• Fault List — The tool can read in an external fault list. The tool uses this list of faults
and their current status as a starting point for test generation.
• Test Patterns — The tool can read in externally generated test patterns and use those
patterns as the source of patterns to be simulated.

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The tool produces the following outputs:

• Test Patterns — The tool generates files containing test patterns. They can generate
these patterns in a number of different simulator and ASIC vendor formats. “Test
Pattern Formatting and Timing” on page 365 discusses the test pattern formats in more
detail.
• ATPG Information Files — These consist of a set of files containing information from
the ATPG session. For example, you can specify creation of a log file for the session.
• Fault List — This is an ASCII-readable file that contains internal fault information in
the standard Mentor Graphics fault format.

Understanding the ATPG Method

To understand how the ATPG tool operates, you should understand the basic ATPG process,
timing model, and basic pattern types that the tool produces. The following subsections discuss
these topics.

Basic ATPG Process

The ATPG tool has default values set, so when you start ATPG for the first time (by issuing the
create_patterns command), the tool performs an efficient combination of random pattern fault
simulation and deterministic test generation on the target fault list.
“The ATPG Process” on page 30 discusses the basics of random and deterministic pattern
generation.

Random Pattern Generation Using the ATPG Tool

The tool first performs random pattern fault simulation for each capture clock, stopping when a
simulation pattern fails to detect at least 0.5% of the remaining faults. The tool then performs
random pattern fault simulation for patterns without a capture clock, as well as those that
measure the primary outputs connected to clock lines.
Note
ATPG constraints and circuitry that can have bus contention are not optimal conditions
for random pattern generation. If you specify ATPG constraints, The tool does not
perform random pattern generation.

Deterministic Test Generation Using the ATPG Tool

Some faults have a very low chance of detection using a random pattern approach. Thus, after it
completes the random pattern simulation, the tool performs deterministic test generation on
selected faults from the current fault list. This process consists of creating test patterns for a set
of (somewhat) randomly chosen faults from the fault list.

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During this process, the tool identifies and removes redundant faults from the fault list. After it
creates enough patterns for a fault simulation pass, it displays a message that indicates the
number of redundant faults, the number of ATPG untestable faults, and the number of aborted
faults that the test generator identifies. The tool then once again invokes the fault simulator,
removing all detected faults from the fault list and placing the effective patterns in the test set.
The tool then selects another set of patterns and iterates through this process until no faults
remain in the current fault list, except those aborted during test generation (that is, those in the
UC or UO categories).

ATPG Tool Timing Model

The tool uses a cycle-based timing model, grouping the test pattern events into test cycles. The
ATPG tool simulator uses the non-scan events: force_pi, measure_po, capture_clock_on,
capture_clock_off, ram_clock_on, and ram_clock_off. The tool uses a fixed test cycle type
for ATPG; that is, you cannot modify it.
The most commonly used test cycle contains the events: force_pi, measure_po,
capture_clock_on, and capture_clock_off. The test vectors used to read or write into RAMs
contain the events force_pi, ram_clock_on, and ram_clock_off. You can associate real times
with each event via the timing file.

ATPG Tool Pattern Types

The ATPG tool has several different types of testing modes. That is, it can generate several
different types of patterns depending on the style and circuitry of the design and the information
you specify. By default, the tool generates basic scan patterns, which assume a full-scan design
methodology. The following subsections describe basic scan patterns, as well as the other types
of patterns that the tool can generate.

Basic Scan Patterns

As mentioned, the tool generates basic scan patterns by default. A scan pattern contains the
events that force a single set of values to all scan cells and primary inputs (force_pi), followed
by observation of the resulting responses at all primary outputs and scan cells (measure_po).
The tool uses any defined scan clock to capture the data into the observable scan cells
(capture_clock_on, capture_clock_off). Scan patterns reference the appropriate test procedures
to define how to control and observe the scan cells. The tool requires that each scan pattern be
independent of all other scan patterns. The basic scan pattern contains the following events:
1. Load values into scan chains.
2. Force values on all non-clock primary inputs (with clocks off and constrained pins at
their constrained values).
3. Measure all primary outputs (except those connected to scan clocks).
4. Pulse a capture clock or apply selected clock procedure.

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5. Unload values from scan chains.

While the list shows the loading and unloading of the scan chain as separate events, more
typically the loading of a pattern occurs simultaneously with the unloading of the preceding
pattern. Thus, when applying the patterns at the tester, you have a single operation that loads in
scan values for a new pattern while unloading the values captured into the scan chains for the
previous pattern.

Because the ATPG tool is optimized for use with scan designs, the basic scan pattern contains
the events from which the tool derives all other pattern types.

Clock PO Patterns
The following figure shows that in some designs, a clock signal may go to a primary output
through some combinational logic.

Figure 7-4. Clock-PO Circuitry

Comb.
Logic
Clock Primary
Outputs
...
LA LA

The tool considers any pattern that measures a PO with connectivity to a clock, regardless of
whether or not the clock is active, a clock PO pattern. A normal scan pattern has all clocks off
during the force of the primary inputs and the measure of the primary outputs. However, in the
clocked primary output situation, if the clock is off, a condition necessary to test a fault within
this circuitry might not be met and the fault may go undetected. In this case, in order to detect
the fault, the pattern must turn the clock on during the force and measure. This does not happen
in the basic scan pattern. The tool allows this within a clock PO pattern, to observe primary
outputs connected to clocks.

Clock PO patterns contain the following events:

1. Load values into the scan chains.

2. Force values on all primary inputs, (potentially) including clocks (with constrained pins
at their constrained values).
3. Measure all primary outputs that are connected to scan clocks.
The tool generates clock PO patterns whenever it learns that a clock connects to a primary
output and if it determines that it can only detect faults associated with the circuitry by using a

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clock PO pattern. If you do not want the tool to generate clock PO patterns, you can turn off the
capability as follows:

SETUP> set_pattern_type off

Clock Sequential Patterns

The ATPG tool’s clock sequential pattern type handles limited sequential circuitry, and can also
help in testing designs with RAM. This kind of pattern contains the following events:
1. Load the scan chains.
2. Apply the clock sequential cycle.
a. Force values on all primary inputs, except clocks (with constrained pins at their
constrained values).
b. Pulse the write lines, read lines, capture clock, and/or apply selected clock
procedure.
c. Repeat steps a and b for a total of “N” times, where N is the
clock sequential depth - 1.
3. Apply the capture cycle.
a. Force pi.
b. Measure po.
c. Pulse capture clock.
4. Unload the scan chains as you load the next pattern.
To instruct the tool to generate clock sequential patterns, you must set the sequential depth to
some number greater than one, using the set_pattern_type command as follows:

SETUP> set_pattern_type -sequential 2

A depth of zero indicates combinational circuitry. A depth greater than one indicates limited
sequential circuitry. You should, however, be careful of the depth you specify. You should start
off using the lowest sequential depth and analyzing the run results. You can perform several
runs, if necessary, increasing the sequential depth each time. Although the maximum allowable
depth limit is 255, you should typically limit the value you specify to five or less, for
performance reasons.

Multiple Load Patterns

The tool can optionally include multiple scan chain loads in a clock sequential pattern. By
creating patterns that use multiple loads, the tool can:
• take advantage of a design’s non-scan sequential cells that are capable of retaining their
state through a scan load operation

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• test through a RAM/ROM

You enable the multiple load capability by using “-multiple_load on” with the set_pattern_type
command and setting the sequential depth to some number greater than one. When you activate
this capability, you allow the tool to include a scan load before any pattern cycle.

Note
An exception is at-speed sequences in named capture procedures. A load may not occur
between the at-speed launch and capture cycles. For more information, see the
description of the “load” cycle type in “Defining Internal and External Modes” on
page 247.

Generally, multiple load patterns require a sequential depth for every functional mode clock
pulsed. A minimum sequential depth of 4 is required to enable the tool to create the multiple
cycle patterns necessary for RAM testing. The patterns are very similar to RAM sequential
patterns, but for many designs will give better coverage than RAM sequential patterns. This
method also supports certain tool features (MacroTest, dynamic compression, split-capture
cycle, clock-off simulation) not supported by RAM sequential patterns.

RAM Sequential Patterns

To propagate fault effects through RAM, and to thoroughly test the circuitry associated with a
RAM, the tool generates a special type of pattern called RAM sequential. RAM sequential
patterns are single patterns with multiple loads, which model some sequential events necessary
to test RAM operations. The multiple load events include two address writes and possibly a read
(if the RAM has data hold). This type of pattern contains the following events:
1. Load scan cells.
2. Force primary inputs.
3. Pulse write line(s).
4. Repeat steps 1 through 3 for a different address.
5. Load scan cells.
6. Force primary inputs.
7. Pulse read lines (optional, depending on the RAM’s data hold attribute).
8. Load scan cells.
9. Force primary inputs
10. Measure primary outputs.
11. Pulse capture clock.
12. Unload values from scan cells.

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Performing Basic Operations

The following example explains the operations depicted in this type of pattern. Assume you
want to test a stuck-at-1 fault on the highest order bit of the address lines. You could do this by
writing some data, D, to location 1000. You could then write different data, D’, to location
0000. If a stuck-at-1 fault were present on the highest address bit, the faulty machine would
overwrite location 1000 with the value D’. Next, you would attempt to read from address
location 1000. With the stuck-at-1 fault on the address line, you would read D’.

Conversely, if the fault on the highest order bit of the address line is a stuck-at-0 fault, you
would want to write the initial data, D, to location 0000. You would then write different data,
D’, to location 1000. If a stuck-at-0 fault were present on the highest address bit, the faulty
machine would overwrite location 0000 with the value D’. Next, you would attempt to read
from address location 0000. With the stuck-at-0 fault on the address line, you would read D’.

You can instruct the tool to generate RAM sequential patterns by issuing the set_pattern_type
command as follows:

SETUP> set_pattern_type -ram_sequential on

Sequential Transparent Patterns

Designs containing some non-scan latches can use basic scan patterns if the latches behave
transparently between the time of the primary input force and the primary output measure. A
latch behaves transparently if it passes rule D6.
For latches that do not behave transparently, a user-defined procedure can force some of them to
behave transparently between the primary input force and primary output measure. A test
procedure, which is called seq_transparent, defines the appropriate conditions necessary to
force transparent behavior of some latches. The events in sequential transparent patterns
include:

1. Load scan chains.

2. Force primary inputs.
3. Apply seq_transparent procedure(s).
4. Measure primary outputs.
5. Unload scan chains.
For more information on sequential transparent procedures, refer to “Scan and Clock
Procedures” in the Tessent Shell User’s Manual.

Performing Basic Operations

This section describes the most basic operations you may need to perform with the ATPG tool.

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Invoking the Application

You access ATPG functionality by invoking Tessent Shell and then setting the context to
“patterns -scan”:
% tessent -shell

SETUP> set_context patterns -scan

Setting the System Mode

When Tessent Shell invokes, the tool assumes the first thing you want to do is set up circuit
behavior, so it automatically puts you in setup mode.
To change the system mode to analysis, use the set_system_mode command.

Setting Up Design and Tool Behavior

The first real task you must perform in the basic ATPG flow is to set up information about
design behavior and existing scan circuitry. The following subsections describe how to
accomplish this setup.

Setting Up the Circuit Behavior

The ATPG tool provides a number of commands that let you set up circuit behavior. You must
execute these commands while in setup mode.
A convenient way to execute the circuit setup commands is to place these commands in a dofile,
as explained previously in “Running Tessent Shell as a Batch Job”. The following subsections
describe typical circuit behavior set up tasks.

Defining Equivalent or Inverted Primary Inputs

Within the circuit application environment, often multiple primary inputs of the circuit being
tested must always have the same (equivalent) or opposite values. Specifying pin equivalences
constrains selected primary input pins to equivalent or inverted values relative to the last entered
primary input pin.
The following commands are useful when working with pin equivalences:

• add_input_constraints — Adds pin equivalences.

• delete_input_constraints — Deletes the specified pin equivalences.
• report_input_constraints — Displays the specified pin equivalences.

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Adding Primary Inputs and Outputs

In some cases, you may need to change the test pattern application points (primary inputs) or the
output value measurement points (primary outputs). When you add previously undefined
primary inputs, they are called user class primary inputs, while the original primary inputs are
called system class primary inputs.
To add_primary_inputs to a circuit, at setup mode prompt, use the add_primary_inputs
command. When you add previously undefined primary outputs, they are called user class
primary outputs, while the original primary outputs are called system class primary outputs.

To add_primary_outputs to a circuit, at the setup mode prompt, use the add_primary_outputs

command.

• delete_primary_inputs — Deletes the specified types of primary inputs.

• report_primary_inputs — Reports the specified types of primary inputs.
• delete_primary_outputs — Deletes the specified types of primary outputs.
• report_primary_outputs — Reports the specified types of primary outputs.

Using Bidirectional Pins as Primary Inputs or Outputs

During pattern generation, the ATPG tool automatically determines the mode of bidirectional
pins (bidis) and avoids creating patterns that drive values on these pins when they are not in
input mode. In some situations, however, you might prefer to have the tool treat a bidirectional
pin as a PI or PO. For example, some testers require more memory to store bidirectional pin data
than PI or PO data. Treating each bidi as a PI or PO when generating and saving patterns will
reduce the amount of memory required to store the pin data on these testers.
From the tool’s perspective, a bidi consists of several gates and includes an input port and an
output port. You can use the commands, report_primary_inputs and report_primary_outputs, to
view PIs and POs. Pins that are listed by both commands are bidirectional pins.

Certain other PI-specific and PO-specific commands accept a bidi pinname argument, and
enable you to act on just the applicable port functionality (input or output) of the bidi. For
example, you can use the delete_primary_inputs command with a bidirectional pin argument to
remove the input port of the bidi from the design interface. From then on, the tool will treat that
pin as a PO. You can use the delete_primary_outputs command similarly to delete the output
port of a bidi from the design interface, so the tool treats that bidi as a PI.

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Note
Altering the design’s interface will result in generated patterns that are different than
those the tool would generate for the original interface. It also prevents verification of the
saved patterns using the original netlist interface. If you want to be able to verify saved
patterns by performing simulation using the original netlist interface, you must use the
commands described in the following subsections instead of the
delete_primary_inputs/outputs commands.

Setting Up a Bidirectional Pin as a Primary Output for ATPG Only

With the add_input_constraints command you can get the tool to treat a bidi as a PO during
ATPG only, without altering the design interface within the tool. You do this by constraining
the input part of the bidi to a constant high impedance (CZ) state. The generated patterns will
then contain PO data for the bidi, and you will be able to verify saved patterns by performing
simulation using the original design netlist.

Setting Up a Bidirectional Pin as a Primary Input for ATPG Only

With the add_output_masks command, you can get the tool to treat a bidi as a PI during ATPG
only, without altering the design interface. This command blocks observability of the output
part of the bidi. The generated patterns will then contain PI data for the bidi, and you will be
able to verify saved patterns by performing simulation using the original design netlist.

If the Bidirectional Pin Control Logic is Unknown

Sometimes the control logic for a bidi is unknown. In this situation, you can model the control
logic as a black box. If you want the tool to treat the bidi as a PI, model the output of the black
box to be 0. If you want the bidi treated as a PO, model the output of the black box to be 1.

If the Bidirectional Pin has a Pull-up or Pull-down Resistor

Using default settings, The ATPG tool generates a known value for a bidirectional pad having
pull-up or pull-down resistors. In reality, however, the pull-up or pull-down time is typically
very slow and will result in simulation mismatches when a test is carried out at high speed.
To prevent such mismatches, you should use the add_input_constraints command. This
command changes the tool’s simulation of the I/O pad so that instead of a known value, an X is
captured for all observation points that depend on the pad. The X masks the observation point,
preventing simulation mismatches.

Examples of Setting Up Bidirectional Pins as PIs or POs

The following examples demonstrate the use of the commands described in the preceding
sections about bidirectional pins (bidis). Assume the following pins exist in an example design:
• Bidirectional pins: /my_inout[0]…/my_inout[2]

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• Primary inputs (PIs): /clk, /rst, /scan_in, /scan_en, /my_en

• Primary outputs (POs): /my_out[0]…/my_out[4]
You can view the bidis by issuing the following two commands:

SETUP> report_primary_inputs
SYSTEM: /clk
SYSTEM: /rst
SYSTEM: /scan_in
SYSTEM: /scan_en
SYSTEM: /my_en
SYSTEM: /my_inout[2]
SYSTEM: /my_inout[1]
SYSTEM: /my_inout[0]
SETUP> report_primary_outputs
SYSTEM: /x_out[4]
SYSTEM: /x_out[3]
SYSTEM: /x_out[2]
SYSTEM: /x_out[1]
SYSTEM: /x_out[0]
SYSTEM: /my_inout[2]
SYSTEM: /my_inout[1]
SYSTEM: /my_inout[0]

Pins listed in the output of both commands (shown in bold font) are pins the tool will treat as
bidis during test generation. To force the tool to treat a bidi as a PI or PO, you can remove the
definition of the unwanted input or output port. The following example removes the input port
definition, then reports the PIs and POs. You can see the tool now only reports the bidis as POs,
which reflects how those pins will be treated during ATPG:

SETUP> delete_primary_inputs /my_inout[0] /my_inout[1] /my_inout[2]

SETUP> report_primary_inputs
SYSTEM: /clk
SYSTEM: /rst
SYSTEM: /scan_in
SYSTEM: /scan_en
SYSTEM: /my_en

SETUP> report_primary_outputs
SYSTEM: /x_out[4]
SYSTEM: /x_out[3]
SYSTEM: /x_out[2]
SYSTEM: /x_out[1]
SYSTEM: /x_out[0]
SYSTEM: /my_inout[2]
SYSTEM: /my_inout[1]
SYSTEM: /my_inout[0]

Because the preceding approach alters the design’s interface within the tool, it may not be
acceptable in all cases. Another approach, explained earlier, is to have the tool treat a bidi as a
PI or PO during ATPG only, without altering the design interface. To obtain PO treatment for a

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bidi, constrain the input part of the bidi to the high impedance state. The following command
does this for the /my_inout[0] bidi:

SETUP> add_input_constraints /my_inout[0] -cz

To have the tool treat a bidi as a PI during ATPG only, direct the tool to mask (ignore) the
output part of the bidi. The following example does this for the /my_inout[0] and /my_inout[1]
pins:

SETUP> add_output_masks /my_inout[0] /my_inout[2]

SETUP> report_output_masks
TIEX /my_inout[0]
TIEX /my_inout[2]

The “TIEX” in the output of “report_output_masks” indicates the two pins are now tied to X,
which blocks their observability and prevents the tool from using them during ATPG.

Tying Undriven Signals

Within your design, there could be several undriven nets, which are input signals not tied to
fixed values. When you read a netlist, the application issues a warning message for each
undriven net or floating pin in the module. The ATPG tool must “virtually” tie these pins to a
fixed logic value during ATPG.
If you do not specify a value, the application uses the default value X, which you can change
with the set_tied_signals command.

To add_tied_signals, at the setup mode prompt, use the add_tied_signals command.

This command assigns a fixed value to every named floating net or pin in every module of the
circuit under test.

• set_tied_signals — Sets default for tying unspecified undriven signals.

• delete_tied_signals — Deletes the current list of specified tied signals.
• report_tied_signals — Displays current list of specified tied nets and pins.

Constraining Primary Inputs

The tool can constrain primary inputs during the ATPG process.
To add a pin constraint to a specific pin, use the add_input_constraints command. You can
specify one or more primary input pin pathnames to be constrained to one of the following
formats: constant 0 (C0), constant 1 (C1), high impedance (CZ), or unknown (CX).

For detailed information on the tool-specific usages of this command, refer to

add_input_constraints in the Tessent Shell Reference Manual.

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Masking Primary Outputs

Your design may contain certain primary output pins that have no strobe capability. Or in a
similar situation, you may want to mask certain outputs from observation for design trade-off
experimentation.
In these cases, you could mask these primary outputs using the add_output_masks command.

Note
The tool places faults they can only detect through masked outputs in the AU category—
not the UO category.

Adding Slow Pads

While running tests at high speed, as might be used for path delay test patterns, it is not always
safe to assume that the loopback path from internal registers, via the I/O pad back to internal
registers, can stabilize within a single clock cycle. Assuming that the loopback path stabilizes
within a single clock cycle may cause problems verifying ATPG patterns or may lead to yield
loss during testing.
To prevent a problem caused by this loopback, use the add_input_constraints command to
modify the simulated behavior of the bidirectional I/O pin, on a pin by pin basis.

For a slow pad, the simulation of the I/O pad changes so that the value propagated into the
internal logic is X whenever the primary input is not driven. This causes an X to be captured for
all observation points dependent on the loopback value.

• delete_input_constraints — Resets the specified I/O pin back to the default simulation
mode.
• report_input_constraints — Displays all I/O pins marked as slow.

Setting Up Tool Behavior

In addition to specifying information about the design to the ATPG tool, you can also set up
how you want the ATPG tool to handle certain situations and how much effort to put into
various processes. The following commands are useful for setting up the ATPG tool:
• set_learn_report — Enables access to certain data learned during analysis.
• set_loop_handling — Specifies the method in which to break loops.
• set_pattern_buffer — Enables the use of temporary buffer files for pattern data.
• set_possible_credit — Sets credit for possibly-detected faults.
The following subsections discuss the typical tool setup.

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Checking Bus Contention

If you use contention checking on tri-state driver busses and multiple-port flip-flops and latches,
the tool rejects (from the internal test pattern set) patterns generated by the ATPG process that
can cause bus contention.
To set contention checking, you use the set_contention_check command.

By default, contention checking is on, as are the switches -Warning and -Bus, causing the tool
to check tri-state driver buses and issue a warning if bus contention occurs during simulation.
For more information on the different contention checking options, refer to the
set_contention_check description in the Tessent Shell Reference Manual.

To display the current status of contention checking, use the report_environment command.

• analyze_bus — Analyzes the selected buses for mutual exclusion.

• set_bus_handling — Specifies how to handle contention on buses.
• set_driver_restriction — Specifies whether only a single driver or multiple drivers can
be on for buses or ports.
• report_bus_data — Reports data for either a single bus or a category of buses.
• report_gates — Reports netlist information for the specified gates.

Setting Multi-Driven Net Behavior

When you specify the fault effect of bus contention on tri-state nets with the set_net_dominance
command, you are giving the tool the ability to detect some faults on the enable lines of tri-state
drivers that connect to a tri-state bus.
At the setup mode prompt, you use the set_net_dominance command.

The three choices for bus contention fault effect are And, Or, and Wire (unknown behavior),
Wire being the default. The Wire option means that any different binary value results in an X
state. The truth tables for each type of bus contention fault effect are shown on the references
pages for the set_net_dominance description in the Tessent Shell Reference Manual.

On the other hand, if you have a net with multiple non-tri-state drivers, you may want to specify
this type of net’s output value when its drivers have different values. Using the
set_net_resolution command, you can set the net’s behavior to And, Or, or Wire (unknown
behavior). The default Wire option requires all inputs to be at the same state to create a known
output value. Some loss of test coverage can result unless the behavior is set to And (wired-and)
or Or (wired-or). To set the multi-driver net behavior, at the setup mode prompt, you use the
set_net_resolution command.

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Setting Z-State Handling

If your tester has the ability to distinguish the high impedance (Z) state, you should use the Z
state for fault detection to improve your test coverage. If the tester can distinguish a high
impedance value from a binary value, certain faults may become detectable which otherwise
would at best be possibly detected (pos_det). This capability is particularly important for fault
detection in the enable line circuitry of tri-state drivers.
The default for the ATPG tool is to treat a Z state as an X state. If you want to account for Z
state values during simulation, you can issue the set_z_handling command.

Internal Z handling specifies how to treat the high impedance state when the tri-state network
feeds internal logic gates. External handling specifies how to treat the high impedance state at
the circuit primary outputs. The ability of the tester normally determines this behavior.

To set the internal or external Z handling, use the set_z_handling command at the setup mode
prompt.

For internal tri-state driver nets, you can specify the treatment of high impedance as a 0 state, a
1 state, and an unknown state.

Note
This command is not necessary if the circuit model already reflects the existence of a pull
gate on the tri-state net.

For example, to specify that the tester does not measure high impedance, enter the following:

SETUP> set_z_handling external X

For external tri-state nets, you can also specify that the tool measures high impedance as a 0
state and distinguished from a 1 state (0), measures high impedance as a 1 state and
distinguished from a 0 state (1), measures high impedance as unique and distinguishable from
both a 1 and 0 state (Z).

Controlling the Learning Process

The ATPG tool performs extensive learning on the circuit during the transition from setup to
some other system mode. This learning reduces the amount of effort necessary during ATPG.
The tool allows you to control this learning process.
For example, the tool lets you turn the learning process off or change the amount of effort put
into the analysis. You can accomplish this for combinational logic using the set_static_learning
command.

By default, static learning is on and the simulation activity limit is 1000. This number ensures a
good trade-off between analysis effort and process time. If you want the ATPG tool to perform
maximum circuit learning, you should set the activity limit to the number of gates in the design.

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By default, state transition graph extraction is on. For more information on the learning process,
refer to “Learning Analysis” on page 80.

Setting the Capture Handling

The ATPG tool evaluates gates only once during simulation, simulating all combinational gates
before sequential gates. This default simulation behavior correlates well with the normal
behavior of a synchronous design, if the design model passes design rules checks—particularly
rules C3 and C4. However, if your design fails these checks, you should examine the situation
to see if your design would benefit from a different type of data capture simulation.
For example, examine the design of Figure 7-5. It shows a design fragment which fails the C3
rules check.

Figure 7-5. Data Capture Handling Example

1
d d
0
Q1 Q2
(source) (sink)

C3 violation flagged here

The rules checker flags the C3 rule because Q2 captures data on the trailing edge of the same
clock that Q1 uses. The ATPG tool considers sequential gate Q1 as the data source and Q2 as
the data sink. By default, the tool simulates Q2 capturing old data from Q1. However, this
behavior most likely does not correspond to the way the circuit really operates. In this case, the
C3 violation should alert you that simulation could differ from real circuit operation.

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To allow greater flexibility of capture handling for these types of situations, the tool provides
some commands that alter the default simulation behavior. The set_split_capture_cycle
command, for example, effects whether or not the tool updates simulation data between clock
edges. When set to “on”, the tool is able to determine correct capture values for trailing edge
and level-sensitive state elements despite C3 and C4 violations. If you get these violations, issue
the set_split_capture_cycle ON command.

You can select modified capture handling for level sensitive or trailing edge gates. For these
types of gates, you select whether you want simulation to use old data, new data, or X values.

The set_capture_handling command changes the data capture handling globally for all the
specified types of gates that fail C3 and C4. If you want to selectively change capture handling,
you can use the add_capture_handling command.

You can specify the type of data to capture, whether the specified gate(s) is a source or sink
point, and the gates or objects (identified by ID number, pin names, instance names, or cell
model names) for which to apply the special capture handling.

Note
When you change capture handling to simulate new data, the tool only performs new data
simulation for one additional level of circuitry. That is, sink gates capture new values
from their sources. However, if the sources are also sinks that are set to capture new data,
the tool does not simulate this effect.

For more information about set_capture_handling or add_capture_handling, refer to the Tessent

Shell Reference Manual. For more information about C3 and C4 rules violations, refer to
“Clock Rules” in the Tessent Shell Reference Manual.

• delete_capture_handling — Removes special data capture handling for the specified

objects.
• set_drc_handling — Specifies violation handling for a design rules check.

Setting Transient Detection

You can set how the tool handles zero-width events on the clock lines of state elements.
The tool lets you turn transient detection on or off with the set_transient_detection command.

With transient detection off, DRC simulation treats all events on state elements as valid.
Because the simulator is a zero delay simulator, it is possible for DRC to simulate zero-width
monostable circuits with ideal behavior, which is rarely matched in silicon. The tool treats the
resulting zero-width output pulse from the monostable circuit as a valid clocking event for other
state elements. Thus, state elements change state although their clock lines show no clocking
event.

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With transient detection on, the tool sets state elements to a value of X if the zero-width event
causes a change of state in the state elements. This is the default behavior upon invocation of the
tool.

Defining the Scan Data

You must define the scan clocks and scan chains before the application performs rules checking
(which occurs upon exiting setup mode). The following subsections describe how to define the
various types of scan data.

Defining Scan Clocks

The tool considers any signals that capture data into sequential elements (such as system clocks,
sets, and resets) to be scan clocks. Therefore, to take advantage of the scan circuitry, you need to
define these “clock signals” by adding them to the clock list.
You must specify the off-state for pins you add to the clock list. The off-state is the state in
which clock inputs of latches are inactive. For edge-triggered devices, the off-state is the clock
value prior to the clock’s capturing transition. You add clock pins to the list by using the
add_clocks command.

You can constrain a clock pin to its off-state to suppress its usage as a capture clock during the
ATPG process. The constrained value must be the same as the clock off-state, otherwise an
error occurs. If you add an equivalence pin to the clock list, all of its defined equivalent pins are
also automatically added to the clock list.

• delete_clocks — Deletes the specified pins from the clock list.

• report_clocks — Reports all defined clock pins.

Defining Scan Groups

A scan group contains a set of scan chains controlled by a single test procedure file. You must
create this test procedure file prior to defining the scan chain group that references it.
To define scan groups, you use the add_scan_groups command. These commands are also
useful:

• delete_scan_groups — Deletes specified scan groups and associated chains.

• report_scan_groups — Displays current list of scan chain groups.

Defining Scan Chains

After defining scan groups, you can define the scan chains associated with the groups. For each
scan chain, you must specify the name assigned to the chain, the name of the chain’s group, the
scan chain input pin, and the scan chain output pin.

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Note
Scan chains of a scan group can share a common scan input pin, but this condition
requires that both scan chains contain the same data after loading.

To define scan chains and their associated scan groups, you use the add_scan_chains command.
These commands are also useful:

• delete_scan_chains — Deletes the specified scan chains.

• report_scan_chains — Displays current list of scan chains.

Setting the Clock Restriction

You can specify whether or not to allow the test generator to create patterns that have more than
one non-equivalent capture clock active at the same time.
To set the clock restriction, you use the set_clock_restriction command.

Note
If you choose to turn off the clock restriction, you should verify the generated pattern set
using a timing simulator—to ensure there are no timing errors.

Adding Constraints to Scan Cells

The tool can constrain scan cells to a constant value (C0 or C1) during the ATPG process to
enhance controllability or observability. Additionally, the tools can constrain scan cells to be
either uncontrollable (CX), unobservable (OX), or both (XX).
You identify a scan cell by either a pin pathname or a scan chain name plus the cell’s position in
the scan chain.

To add constraints to scan cells, you use the add_cell_constraints command.

If you specify the pin pathname, it must be the name of an output pin directly connected
(through only buffers and inverters) to a scan memory element. In this case, the tool sets the
scan memory element to a value such that the pin is at the constrained value. An error condition
occurs if the pin pathname does not resolve to a scan memory element.

If you identify the scan cell by chain and position, the scan chain must be a currently-defined
scan chain and the position is a valid scan cell position number. The scan cell closest to the
scan-out pin is in position 0. The tool constrains the scan cell’s MASTER memory element to
the selected value. If there are inverters between the MASTER element and the scan cell output,
they may invert the output’s value.

• delete_cell_constraints — Deletes the constraints from the specified scan cells.

• report_cell_constraints — Reports all defined scan cell constraints.

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Adding Nofault Settings

Within your design, you may have instances that should not have internal faults included in the
fault list. You can label these parts with a nofault setting.
To add a nofault setting, you use the add_nofaults command.

You can specify that the listed pin pathnames, or all the pins on the boundary and inside the
named instances, are not allowed to have faults included in the fault list.

• delete_nofaults — Deletes the specified nofault settings.

• report_nofaults — Displays all specified nofault settings.

Checking Rules and Debugging Rules

Violations
If an error occurs during the rules checking process, the application remains in setup mode so
you can correct the error. You can easily resolve the cause of many such errors; for instance,
those that occur during parsing of the test procedure file. Other errors may be more complex and
difficult to resolve, such as those associated with proper clock definitions or with shifting data
through the scan chain.
The ATPG tool performs model flattening, learning analysis, and rules checking when you try
to exit setup mode. Each of these processes is explained in detail in “Understanding Common
Tool Terminology and Concepts” on page 67. To change from setup to one of the other system
modes, you enter the set_system_mode command.

Note
The ATPG tool does not require the Drc mode because it uses the same internal design
model for all of its processes.

The “Troubleshooting Rules Violations” section in the Tessent Shell Reference Manual
discusses some procedures for debugging rules violations. The Debug window of
DFTVisualizer is especially useful for analyzing and debugging certain rules violations. The
“Attributes and DFTVisualizer” section in the Tessent User’s Manual discusses DFTVisualizer
in detail.

Running Good/Fault Simulation on Existing

Patterns
The purpose of fault simulation is to determine the fault coverage of the current pattern source
for the faults in the active fault list. The purpose of “good” simulation is to verify the simulation

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model. Typically, you use the good and fault simulation capabilities of the ATPG tool to grade
existing hand- or ATPG-generated pattern sets.

Fault Simulation
The following subsections discuss the procedures for setting up and running fault simulation
using the ATPG tool.

Preparing for Fault Simulation

Fault simulation runs in analysis mode without additional setup. You enter analysis mode using
the following command:
SETUP> set_system_mode analysis

Setting the Fault Type

By default, the fault type is stuck-at. If you want to simulate patterns to detect stuck-at faults,
you do not need to issue this command.
If you wish to change the fault type to toggle, pseudo stuck-at (IDDQ), transition, path delay, or
bridge, you can issue the set_fault_type command.

Whenever you change the fault type, the application deletes the current fault list and current
internal pattern set.

Creating the Faults List

Before you can run fault simulation, you need an active fault list from which to run.
You create the faults list using the add_faults command. Typically, you would create this list
using all faults as follows:

> add_faults -all

“Setting Up the Fault Information for ATPG” on page 193 provides more information on
creating the fault list and specifying other fault information.

Setting the Pattern Source

You can have the tools perform simulation and test generation on a selected pattern source,
which you can change at any time.
To use an external pattern source, you use the read_patterns command.

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Note
You may notice a slight drop in test coverage when using an external pattern set as
compared to using generated patterns. This is an artificial drop.

The ATPG tool can perform simulation with a select number of random patterns. Refer to the
Tessent Shell Reference Manual for additional information about these application-specific
simulate_patterns command options.

• set_capture_clock — Specifies the capture clock for random pattern simulation.

• set_random_clocks — Specifies the selection of clock_sequential patterns for random
pattern simulation.
• set_random_patterns — Specifies the number of random patterns to be simulated.

Executing Fault Simulation

You execute the fault simulation process by using the simulate_patterns command. You can
repeat this command as many times as you want for different pattern sources.
• simulate_patterns — Executes the fault simulation process.
• report_faults — Displays faults for selected fault classes.
• report_statistics — Displays a statistics report.

Writing the Undetected Faults List

Typically, after performing fault simulation on an external pattern set, you will want to save the
faults list. You can then use this list as a starting point for ATPG.
To save the faults, you use the write_faults command. For more information, refer to “Writing
Faults to an External File” on page 195.

To read the faults back in for ATPG, go to analysis mode (using set_system_mode) and enter
the read_faults command.

Debugging the Fault Simulation

To debug your fault simulation, you can write a list of pin values that differ between the faulty
and good machine.
Do this using the add_lists and set_list_file commands.

The add_lists command specifies which pins you want reported. The set_list_file command
specifies the name of the file in which to place simulation values for the selected pins. The
default behavior is to write pin values to standard output.

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Resetting Circuit and Fault Status

You can reset the circuit status and status of all testable faults in the fault list to undetected.
Doing so lets you re-run the fault simulation using the current fault list, which does not cause
deletion of the current internal pattern set.
To reset the testable faults in the current fault list, enter the reset_state command.

Good-Machine Simulation
Given a test vector, you use good machine simulation to predict the logic values in the good
(fault-free) circuit at all the circuit outputs. The following subsections discuss the procedures
for running good simulation on existing hand- or ATPG-generated pattern sets using the ATPG
tool.

Preparing for Good-Machine Simulation

Good-machine simulation runs in analysis mode without additional setup. You enter analysis
mode using the following command:
SETUP> set_system_mode analysis

Specifying an External Pattern Source

By default, simulation runs using an internal ATPG-generated pattern source. To run simulation
using an external set of patterns, enter the following command:
ANALYSIS> read_patterns filename

Debugging the Good Machine Simulation

You can debug your good machine simulation in several ways.
If you want to run the simulation and save the values of certain pins in batch mode, you can use
the add_lists and set_list_file commands. The add_lists command specifies which pins to report.
The set_list_file command specifies the name of the file in which you want to place simulation
values for the selected pins.

If you prefer to perform interactive debugging, you can use the simulate_patterns and
report_gates commands to examine internal pin values.

Resetting Circuit Status

In analysis mode, you can reset the circuit status by using the reset_state command.

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Running Random Pattern Simulation

The following subsections show the typical procedure for running random pattern simulation.

Changing to the Fault System Mode

You run random pattern simulation in the analysis system mode.
If you are not already in the analysis system mode, use the set_system_mode command as in the
following example:

SETUP> set_system_mode analysis

Adding the Faults List

You can generate the faults list and eliminate all untestable faults.
To generate the faults list and eliminate all untestable faults, use the add_faults and
delete_faults commands together as in the following example:

> add_faults -all

> delete_faults -untestable

In this example, the delete_faults command with the -untestable switch removes faults from the
fault list that are untestable using random patterns.

Running the Simulation

You can run random pattern simulation.
To run the random pattern simulation, enter the “simulate_patterns -source random” command.
After the simulation run, you can display the undetected faults with the report_faults command.
Some of the undetected faults may be redundant. You can run ATPG on the undetected faults to
identify those that are redundant.

Setting Up the Fault Information for ATPG

Prior to performing test generation, you must set up a list of all faults the application has to
evaluate. The tool can either read the list in from an external source, or generate the list itself.
The type of faults in the fault list vary depending on the fault model and your targeted test type.
For more information on fault modeling and the supported models, refer to “Fault Modeling” on
page 35.

After the application identifies all the faults, it implements a process of structural equivalence
fault collapsing from the original uncollapsed fault list. From this point on, the application

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works on the collapsed fault list. The results, however, are reported for both the uncollapsed and
collapsed fault lists. Executing any command that changes the fault list causes the tool to
discard all patterns in the current internal test pattern set due to the probable introduction of
inconsistencies. Also, whenever you re-enter setup mode, it deletes all faults from the current
fault list. The following subsections describe how to create a fault list and define fault related
information.

Changing to the Analysis System Mode

You can enter the fault list commands from analysis system mode.
You switch from setup to the analysis mode using the set_system_mode command.

For example, assuming your circuit passes rules checking with no violations, you can exit the
setup system mode and enter the analysis system mode as follows:

SETUP> set_system_mode analysis

Setting the Fault Type

By default, the fault type is stuck-at. If you want to generate patterns to detect stuck-at faults,
you do not need to issue the set_fault_type command.
If you wish to change the fault type to toggle, pseudo stuck-at (IDDQ), transition, or path delay,
you can issue the set_fault_type command.

Whenever you change the fault type, the application deletes the current fault list and current
internal pattern set.

Creating the Faults List

The application creates the internal fault list the first time you add_faults or load in external
faults. Typically, you would create a fault list with all possible faults of the selected type,
although you can place some restrictions on the types of faults in the list.
To create a list with all faults of the given type, enter the add_faults command using the -All
switch as in the following example:

ANALYSIS> add_faults -all

If you do not want all possible faults in the list, you can use other options of the add_faults
command to restrict the added faults. You can also specify no-faulted instances to limit placing
faults in the list. You flag instances as “Nofault” while in setup mode. For more information,
refer to “Adding Nofault Settings” on page 189.

When the tool first generates the fault list, it classifies all faults as uncontrolled (UC).

• delete_faults — Deletes the specified faults from the current fault list.

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• report_faults — Displays the specified types of faults.

Adding Faults to an Existing List

You can add new faults to the current fault list.
To add new faults to the current fault list, enter the add_faults command. You must enter either
a list of object names (pin pathnames or instance names) or use the -All switch to indicate the
pins whose faults you want added to the fault list. You can use the -Stuck-at switch to indicate
which stuck faults on the selected pins you want added to the list. If you do not use the Stuck-at
switch, the tool adds both stuck-at-0 and stuck-at-1 faults. The tool initially places faults added
to a fault list in the undetected-uncontrolled (UC) fault class.

Loading Faults from an External List

You can place faults from a previous run (from an external file) into the internal fault list.
To read faults from an external file into the current fault list, enter the read_faults command.

The applications support external fault files in the 3, 4, or 6 column formats. The only data they
use from the external file is the first column (stuck-at value) and the last column (pin
pathname), unless you use the -Restore option.

The -Retain option causes the application to retain the fault class (second column of
information) from the external fault list. The -Delete option deletes all faults in the specified file
from the internal faults list. The -DELETE_Equivalent option, in the ATPG tool, deletes from
the internal fault list all faults listed in the file, as well as all their equivalent faults.

Note
In the ATPG tool, the filename specified cannot have fault information lines with
comments appended to the end of the lines or fault information lines greater than five
columns. The tool will not recognize the line properly and will not add the fault on that
line to the fault list.

Writing Faults to an External File

You can write all or only selected faults from a current fault list into an external file. You can
then edit or load this file to create a new fault list.
To write_faults to a file, enter the write_faults command. You must specify the name of the file
you want to write.

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Setting the Fault Sampling Percentage

By reducing the fault sampling percentage (which by default is 100%), you can decrease the
process time to evaluate a large circuit by telling the application to process only a fraction of the
total collapsed faults.
To set the fault sampling percentage, use the set_fault_sampling command. You must specify a
percentage (between 1 and 100) of the total faults you want processed.

Setting the Fault Mode

You can specify use of either the collapsed or uncollapsed fault list for fault counts, test
coverages, and fault reports. The default is to use uncollapsed faults.
To set the fault mode, you use the set_fault_mode command.

Setting the Possible-Detect Credit

Before reporting test coverage, fault coverage, and ATPG effectiveness, you should specify the
credit you want given to possible-detected faults.
To set the credit to be given to possible-detected faults, use the set_possible_credit command.
The selected credit may be any positive integer less than or equal to 100, the default being 50%.

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Performing ATPG
Obtaining the optimal test set in the least amount of time is a desirable goal.
Figure 7-6 outlines how to most effectively meet this goal.

Figure 7-6. Efficient ATPG Flow

Set Up for
ATPG

Create Patterns

N
Coverage Adjust
Good? ATPG Approach

Write Patterns

The first step in the process is to perform any special setup you may want for ATPG. This
includes such things as setting limits on the pattern creation process itself. The second step is to
create patterns with default settings (see page 205). This is a very fast way to determine how
close you are to your testability goals. You may even obtain the test coverage you desire from
your very first run. However, if your test coverage is not at the required level, you may have to
troubleshoot the reasons for the inadequate coverage and create additional patterns using other
approaches (see page 205).

Setting Up for ATPG

Prior to ATPG, you may need to set certain criteria that aid the test generators in the test
generation process.
If you just want to generate patterns quickly in the ATPG tool using default settings, the
recommended method for pattern creation is using the create_patterns command.

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If the initial patterns are unsatisfactory, then run the create_patterns command a second time. If,
however, you are still unable to create a satisfactory pattern set, then use the set_pattern_type
command in conjunction with the create_patterns command using the following sequence:

ANALYSIS> set_pattern_type -sequential 2

ANALYSIS> create_patterns

A reasonable practice is creating patterns using these two commands with the sequential depth
set to 2. This is described in more detail in “Creating Patterns with Default Settings” on
page 205.

Defining ATPG Constraints

ATPG constraints are similar to pin constraints and scan cell constraints. Pin constraints and
scan cell constraints restrict the values of pins and scan cells, respectively. ATPG constraints
place restrictions on the acceptable kinds of values at any location in the circuit. For example,
you can use ATPG constraints to prevent bus contention or other undesirable events within a
design. Additionally, your design may have certain conditions that can never occur under
normal system operation. If you want to place these same constraints on the circuit during
ATPG, use ATPG constraints.
During deterministic pattern generation, only the restricted values on the constrained circuitry
are allowed. Unlike pin and scan cell constraints, which are only available in setup mode, you
can define ATPG constraints in any system mode after design flattening. If you want to set
ATPG constraints prior to performing design rules checking, you must first create a flattened
model of the design using the create_flat_model command.

ATPG constraints are useful when you know something about the way the circuit behaves that
you want the ATPG process to examine. For example, the design may have a portion of
circuitry that behaves like a bus system; that is, only one of various inputs may be on, or
selected, at a time. Using ATPG constraints, combined with a defined ATPG function, you can
specify this information to the ATPG tool. ATPG functions place artificial Boolean
relationships on circuitry within your design. After defining the functionality of a portion of
circuitry with an ATPG function, you can then constrain the value of the function as desired
with an ATPG constraint. This is more useful than just constraining a point in a design to a
specific value.

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To define ATPG functions, use the add_atpg_functions command. When using this command,
you specify a name, a function type, and an object to which the function applies.

You can specify ATPG constraints with the add_atpg_constraints command. When using this
command, you specify a value, an object, a location, and a type.

Test generation considers all current constraints. However, design rules checking considers only
static constraints. You can only add or delete static constraints in setup mode. Design rules
checking generally does not consider dynamic constraints, but there are some exceptions
detailed in the set_drc_handling command reference description (see the Atpg_analysis and
ATPGC options). You can add or delete dynamic constraints at any time during the session. By
default, ATPG constraints are dynamic.

Figure 7-7 and the following commands give an example of how you use ATPG constraints and
functions together.

Figure 7-7. Circuitry with Natural “Select” Functionality

0
/u1

/u2 0 contention-
/u5 free
1
/u3

0
/u4

The circuitry of Figure 7-7 includes four gates whose outputs are the inputs of a fifth gate.
Assume you know that only one of the four inputs to gate /u5 can be on at a time, such as would
be true of four tri-state enables to a bus gate whose output must be contention-free. You can
specify this using the following commands:

ANALYSIS> add_atpg_functions sel_func1 select1 /u1/o /u2/o /u3/o /u4/o

ANALYSIS> add_atpg_constraints 1 sel_func1

These commands specify that the “select1” function applies to gates /u1, /u2, /u3, and /u4 and
the output of the select1 function should always be a 1. Deterministic pattern generation must

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ensure these conditions are met. The conditions causing this constraint to be true are shown in
Table 7-1. When this constraint is true, gate /u5 will be contention-free.

Table 7-1. ATPG Constraint Conditions

/u1 /u2 /u3 /u4 sel_func1 /u5
0 0 0 1 1 contention-free
0 0 1 0 1 contention-free
0 1 0 0 1 contention-free
1 0 0 0 1 contention-free

Given the defined function and ATPG constraint you placed on the circuitry, the ATPG tool
only generates patterns using the values shown in Table 7-1.

Typically, if you have defined ATPG constraints, the tools do not perform random pattern
generation during ATPG. However, using the ATPG tool you can perform random pattern
simulation using simulate_patterns command. In this situation, the tool rejects patterns during
fault simulation that do not meet the currently-defined ATPG constraints.

• analyze_atpg_constraints — Analyzes a given constraint for either its ability to be

satisfied or for mutual exclusivity.
• analyze_restrictions — Performs an analysis to automatically determine the source of
the problems from a failed ATPG run.
• delete_atpg_constraints — Removes the specified constraint from the list.
• delete_atpg_functions — Removes the specified function definition from the list.
• report_atpg_constraints — Reports all ATPG constraints in the list.
• report_atpg_functions — Reports all defined ATPG functions.

Excluding Power and Ground Ports from the Pattern Set

You can constrain power and ground ports during circuit setup and DRC, and also exclude those
ports from the pattern set. The first step is to specify the power and ground ports by setting the
value of the “function” attribute on a top-level input or inout port. For example, the following
command designates the “vcc” port as a power input:
SETUP> set_attribute_value vcc -name function -value power

Note that the port can only be an input or inout port on a top-level module, and the only
allowable values are “power” and “ground.” Also, the port cannot be an IJTAG port in the ICL
file, if present. And you cannot change the “function” attribute after reading in the flat model.

Setting the value to “power” has the effect of adding an input constraint of CT1 on that port; and
setting the value to “ground” has the effect of adding an input constraint of CT0 on that port.

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The only way to remove these inferred constraints is by using the reset_attribute_value
command. That is, the delete_input_constraints and delete_pin_constraints commands do not
work in this situation.

After specifying the power and ground ports, you can write patterns that exclude those ports
using the “write_patterns -parameter_list” command. Also, the parameter file has a keyword
ALL_EXCLUDE_POWER_GROUND that allows you to control whether power and ground
ports are excluded from tester pattern file formats, such as STIL and WGL. For more
information, refer to the ALL_EXCLUDE_POWER_GROUND keyword description in the
“Parameter File Format and Keywords” section of the Tessent Shell Reference Manual.

Setting ATPG Limits

Normally, there is no need to limit the ATPG process when creating patterns. There may be an
occasional special case, however, when you want the tool to terminate the ATPG process if
CPU time, test coverage, or pattern (cycle) count limits are met.
To set these limits, use the set_atpg_limits command.

Understanding Event Simulation for DFFs and Latches

The following explains how the tool’s simulation kernel models DFFs and latches. The kernel
simulates clock pulses as “010” or “101.” The three-digit notation refers to the state of the three
frames that comprise a clock cycle: clock_off frame, post-LE frame, and post-TE frame. There
are always at least three events per cycle.
The kernel distinguishes only between “edge-triggered” and “level-sensitive” sequential
elements:

• Edge-triggered elements update on 0➜1 transition.

• Level-sensitive elements update on clock=1.
The kernel updates all elements immediately in the same frame. For more information, refer to
Figure 7-8.

Figure 7-8. Simulation Frames

Post-LE Frame Clock_off Frame
Forces PI value event
Post-LE Frame
Forces clock ON, updates LE
DFF/latches
CLK
Post-TE Frame
Forces clock OFF, updates TE
DFF/latches and propagates
Clock_off Frame Clock_off Frame
their values
Post-TE Frame

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Figure 7-9 explains how the simulation kernel models waveforms with DFFs and latches.

Figure 7-9. Waveform Modeling for DFFs and Latches

flop_a (LE) 1. The first frame of the CLK input forces the DFF to output its init
value of 1 on the Q pin.
000 100 2. The second frame of the CLK input forces the CLK ON due to the
D Q
010 011 0➜1 edge, the DFF evaluates its D input and outputs Q=0 for the
CLK QB
second frame.
3. The third frame of the CLK input forces the CLK OFF, the DFF
dff
performs no evaluation and holds state Q=0 for the third frame.

latch_a 1. The first frame of the CLK input forces the latch to output its init
value of 1 on the Q pin.
100 2. The second frame of the CLK input forces the level-sensitive CLK
D 100
010 Q ON, the latch evaluates its D input and outputs Q=0 for the second
CLK
frame.
3. The third frame of the CLK input forces the CLK OFF, the latch
latch
performs no evaluation and holds state Q=0 for the third frame.

flop_b (TE) 1. The first frame of the CLK input forces the DFF to output its init
value of 1 on the Q pin.
100 110 2. The second frame of the CLK input forces the CLK OFF, the DFF
D Q
101 001 holds state and outputs Q=1 for the second frame.
CLK QB
3. The third frame of the CLK input forces the CLK ON due to the
0➜1 edge, the DFF evaluates the third frame of its D input and
dff
outputs Q=0.

Displaying Event Simulation Data for a Gate

You can display event simulation data for a gate.
You do this using the set_gate_report and report_gates commands. For examples of how to do
this refer to “Example 5” in the set_gate_report command description in the Tessent Shell
Reference Manual.

Using a Flattened Model to Save Time and Memory

To save time and memory, you can flatten your design and subsequently use the flattened model
instead of the design netlist with the tool.

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Note
Before you save a flattened version of your design, ensure you have specified all
necessary settings accurately. Some design information, such as that related to hierarchy,
is lost when you flatten the design. Consequently, commands that require this information
do not operate with the flattened netlist. Some settings, once incorporated in the flattened
netlist, cannot be changed (for example, a tied constraint you apply to a primary input
pin).

When you reinvoke the tool, you use this flattened netlist by specifying the “-flat” switch. The
tool reinvokes in the same mode (setup or analysis) the tool was in when you saved the flattened
model.

You flatten your design with the tool using one of the following methods depending on the
tool’s current mode:

• analysis mode — The tool automatically creates the flat model when you change from
setup to analysis mode using the set_system_mode command. After model flattening,
you save the flattened design using the write_flat_model command.
• setup mode — You can manually create a flattened model in setup mode using the
create_flat_model command. After model flattening, you save the flattened design using
the write_flat_model command.
You can read a flat model into the tool in setup mode using the read_flat_model command.

An advantage of using a flattened netlist rather a regular netlist, is that you save memory and
have room for more patterns.

Creating a Pattern Buffer Area

To reduce demands on virtual memory when you are running the tool with large designs, use the
set_pattern_buffer command with the ATPG tool. The tool will then store runtime pattern data
in temporary files rather than in virtual memory.

Using Fault Sampling to Save Processing Time

Another command, set_fault_sampling, enables you to perform quick evaluation runs of large
designs prior to final pattern generation. Intended for trial runs only, you can use this command
to reduce the processing time when you want a quick estimate of the coverage to expect with
your design.

Setting Up for Checkpointing

The term “checkpointing” refers to when the tool automatically saves test patterns at regular
periods, called checkpoints, throughout the pattern creation process. This is useful when ATPG
takes a long time and there is a possibility it could be interrupted accidentally. For example, if a

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system failure occurs during ATPG, checkpointing enables you to recover and continue the run
from close to the interruption point. You do not have to redo the entire pattern creation process
from the beginning. The continuation run uses the data saved at the checkpoint, just prior to the
interruption, saving you the time required to recreate the patterns that would otherwise have
been lost.
The set_checkpointing_options command turns the checkpoint functionality on or off, specifies
the time period between each write of the test patterns, as well as the name of the pattern file to
which the tool writes the patterns.

Example Checkpointing
Suppose a large design takes several days for the ATPG tool to process. You do not want to
restart pattern creation from the beginning if a system failure ends ATPG one day after it
begins. The following dofile segment defines a checkpoint interval of 90 minutes and enables
checkpointing.
set_checkpointing_options on -pattern_file my_checkpoint_file -period 90 \
-replace -pattern_format ascii -faultlist_file my_checkpoint_fault_file

If you need to perform a continuation run, invoking on a flattened model can be much faster
than reflattening the netlist (see “Using a Flattened Model to Save Time and Memory” on
page 202 for more information). After the tool loads the design, but before you continue the
interrupted run, be sure to set all the same constraints you used in the interrupted run. The next
dofile segment uses checkpoint data to resume the interrupted run:

# Load the fault population stored by the checkpoint.

#
# The ATPG process can spend a great deal of time proving
# faults to be redundant (RE) or ATPG untestable (AU). By
# loading the fault population using the -restore option, the
# status of these fault sites will be restored. This will
# save the time required to reevaluate these fault sites.
read_faults my_checkpoint_fault_file -restore
#
# The report_statistics command shows if the fault coverage
# is at the same level as at the last checkpoint the tool
# encountered.
report_statistics
#
# Set the pattern source to the pattern set that was stored
# by the checkpoint. Then fault simulate these patterns.
# During the fault simulation, the external patterns will be
# copied into the tool’s internal pattern set. Then, by
# setting the pattern source back to the internal pattern
# set, additional patterns can be added during a subsequent
# ATPG run. This sequence is accomplished with the following
# segment of the dofile.
#
# Fault grade the checkpoint pattern set.
read_patterns my_checkpoint_file
#

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# Reset the fault status to assure that the patterns

# simulated do detect faults. When the pattern set is fault
# simulated, if no faults are detected, the tool will not
# retain the patterns in the internal pattern set.
reset_state
simulate_patterns
report_statistics
#
# Create additional ATPG patterns
create_patterns

After it executes the above commands, the tool should be at the same fault grade and number of
patterns as when it last saved checkpoint data during the interrupted run. To complete the
pattern creation process, you can now use the create_patterns command as described in the next
section.

Creating Patterns with Default Settings

By default, the create_patterns command initiates an optimal ATPG process that includes
highly efficient pattern compression:
ANALYSIS> create_patterns

Review the transcript for any command or setting changes the tool suggests and implement
those that will help you achieve your test goals. Refer to the create_patterns command
description in the Tessent Shell Reference Manual for more information.

If the first pattern creation run gives inadequate coverage, refer to “Approaches for Improving
ATPG Efficiency” on page 205. To analyze the results if pattern creation fails, use the
analyze_atpg_constraints command and the analyze_restrictions command.

Approaches for Improving ATPG Efficiency

If you are not satisfied with the test coverage after initially creating patterns, or if the resulting
pattern set is unacceptably large, you can make adjustments to several system defaults to
improve results in another ATPG run. The following subsections provide helpful information
and strategies for obtaining better results during pattern creation.

Understanding the Reasons for Low Test Coverage

There are two basic reasons for low test coverage:
• Constraints on the tool
• Abort conditions
A high number of faults in the ATPG_untestable (AU) or PU fault categories indicates the
problem lies with tool constraints. PU faults are a type of possible-detected, or Posdet (PD),
fault. A high number of UC and UO faults, which are both AU faults are categorized into
several predefined fault sub-classes, as listed in Table 2-3 on page 62. faults, indicates the

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problem lies with abort conditions. If you are unfamiliar with these fault categories, refer to
“Fault Classes” on page 55.

When trying to establish the cause of low test coverage, you should examine the messages the
tool prints during the deterministic test generation phase. These messages can alert you to what
might be wrong with respect to Redundant (RE) faults, ATPG_untestable (AU) faults, and
aborts. If you do not like the progress of the run, you can terminate the process with CTRL-C.

If a high number of aborted faults (UC or UO) appears to cause the problem, you can set the
abort limit to a higher number, or modify some command defaults to change the way the
application makes decisions. The number of aborted faults is high if reclassifying them as
Detected (DT) or Posdet (PD) would result in a meaningful improvement in test coverage. In
the tool’s coverage calculation (see “Testability Calculations” on page 65), these reclassified
faults would increase the numerator of the formula. You can quickly estimate how much
improvement would be possible using the formula and the fault statistics from your ATPG run.
The following subsections discuss several ways to handle aborted faults.

Note
Changing the abort limit is not always a viable solution for a low coverage problem. The
tool cannot detect ATPG_untestable (AU) faults, the most common cause of low test
coverage, even with an increased abort limit. Sometimes you may need to analyze why a
fault, or set of faults, remain undetected to understand what you can do.

Also, if you have defined several ATPG constraints or have specified set_contention_check On
-Atpg, the tool may not abort because of the fault, but because it cannot satisfy the required
conditions. In either of these cases, you should analyze the buses or ATPG constraints to ensure
the tool can satisfy the specified requirements.

Analyzing a Specific Fault

You can report on all faults in a specific fault category with the report_faults command.
You can analyze each fault individually, using the pin pathnames and types listed by
report_faults, with the analyze_fault command.

The analyze_fault command runs ATPG on the specified fault, displaying information about the
processing and the end results. The application displays different data depending on the
circumstances. You can optionally display relevant circuitry in DFTVisualizer using the
-Display option. See the analyze_fault description in the Tessent Shell Reference Manual for
more information.

You can also report data from the ATPG run using the report_testability_data command within
the ATPG tool for a specific category of faults. This command displays information about
connectivity surrounding the problem areas. This information can give you some ideas as to
where the problem might lie, such as with RAM or clock PO circuitry. Refer to the
report_testability_data description in the Tessent Shell Reference Manual for more information.

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Reporting on Aborted Faults

During the ATPG process, the tool may terminate attempts to detect certain faults given the
ATPG effort required. The tools place these types of faults, called aborted faults, in the AU
fault class, which includes the UC and UO sub-classes.
You can determine why these faults are undetected by using the report_aborted_faults
command.

Setting the Abort Limit

If the fault list contains a number of aborted faults, the tools may be able to detect these faults if
you change the abort limit. You can increase the abort limit for the number of backtracks, test
cycles, or CPU time and recreate patterns.
To set the abort limit using the ATPG tool, use the set_abort_limit command.

The default for combinational ATPG is 30. The clock sequential abort limit defaults to the limit
set for combinational. Both the report_environment command and a message at the start of
deterministic test generation indicate the combinational and sequential abort limits. If they
differ, the sequential limit follows the combinational abort limit.

The application classifies any faults that remain undetected after reaching the limits as aborted
faults—which it considers undetected faults.

Related Command:

• report_aborted_faults — Displays and identifies the cause of aborted faults.

Setting Random Pattern Usage

The ATPG tool lets you specify whether to use random test generation processes when creating
uncompressed patterns. In general, if you use random patterns, the test generation process runs
faster and the number of test patterns in the set is larger. If not specified, the default is to use
random patterns in addition to deterministic patterns. If you use random patterns exclusively,
test coverage is typically very low.
To set random pattern usage for ATPG, use the set_random_atpg command.

Note
The create_patterns command does not use random patterns when generating compressed
patterns.

Changing the Decision Order

Prior to ATPG, the tool learns which inputs of multiple input gates it can most easily control. It
then orders these inputs from easiest to most difficult to control. Likewise, the tool learns which

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outputs can most easily observe a fault and orders these in a similar manner. Then during
ATPG, the tool uses this information to generate patterns in the simplest way possible.
This facilitates the ATPG process, however, it minimizes random pattern detection. This is not
always desirable, as you typically want generated patterns to randomly detect as many faults as
possible. To maximize random pattern detection, the tool provides the set_decision_order
command to allow flexible selection of control inputs and observe outputs during pattern
generation.

Saving the Test Patterns

To save generated test patterns, enter the write_patterns command.
For more information about the test data formats, refer to “Saving Timing Patterns” on
page 369.

Low-Power ATPG
Low-power ATPG allows you to create test patterns that minimize the amount of switching
activity during test to reduce power consumption. Excessive power consumption can
overwhelm the circuit under test, causing it to malfunction.
Low-power ATPG controls switching activity during scan load/unload and capture as described
in the following topics:

• Low-Power Capture
• Low-Power Shift

Note
Low-power constraints are directly related to the number of test patterns generated in an
application. For example, using stricter low-power constraints results in more test
patterns.

Low-Power Capture
Low-power capture employs the clock gaters in a design to achieve the power target. Clock
gaters controlling untargeted portions of the design are turned off, while clock gaters controlling
targeted portions are turned on.
Power is controlled most effectively in designs that employ clock gaters, and especially
multiple levels of clock gaters (hierarchy), to control a majority of the state elements.

This low-power feature is available using the Capture option of the set_power_control
command.

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Low-Power Shift
Low-power shift minimizes the switching activity during shift with a constant fill algorithm
where random values in scan chains are replaced with constant values as they are shifted
through the core; A repeat fill heuristic is used to generate the constant values.
This low-power feature is available in the ATPG tool using the set_power_control command
during test pattern generation.

Setting up Low-Power ATPG

Use this procedure to enable low-power for capture and shift before test patterns are generated.
Setting up low-power ATPG is an iterative process that includes the setup, generation, and
analysis of test patterns.

Prerequisites
• Gate-level netlist with scan chains inserted.
• DFT strategy for your design. A test strategy helps define the most effective testing
process for your design.

Procedure
1. Invoke Tessent Shell:
% tessent -shell my_gate_scan.v -library my_lib.atpg \
-logfile log/atpg_cg.log -replace

2. Set the context to “patterns -scan”:

SETUP> set_context patterns -scan

3. Set up test patterns.

4. Exit setup mode and run DRC:
SETUP> set_system_mode analysis

5. Correct any DRC violations.

6. Turn on low-power capture. For example:
ANALYSIS> set_power_control capture -switching_threshold_percentage 30 \
-rejection_threshold_percentage 35

Switching during the capture cycle is minimized to 30% and any test patterns that
exceed a 35% rejection threshold are discarded.
7. Turn on low-power shift. For example
ANALYSIS> set_power_control shift on -switching_threshold_percentage 20 \
-rejection_threshold_percentage 25

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Low-Power ATPG

Switching during scan chain loading is minimized to 20% and any test patterns that
exceed a 25% rejection threshold are discarded.
8. Create test patterns:
ANALYSIS> create_patterns

Test patterns are generated and the test pattern statistics and power metrics display.
9. Analyze reports, adjust power and test pattern settings until power and test coverage
goals are met. You can use the report_power_metrics command to report the capture
power usage associated with a specific instance or set of modules.
10. Save test patterns. For example:
ANALYSIS> write_patterns ../generated/patterns_edt_p.stil -stil -replace

Related Topics
Low-Power Capture set_power_control
Low-Power Shift

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Creating an IDDQ Test Set

Creating an IDDQ Test Set

The ATPG tool supports the pseudo stuck-at fault model for IDDQ testing.
This fault model allows detection of most of the common defects in CMOS circuits (such as
resistive shorts) without costly transistor level modeling. “IDDQ Test” on page 33 introduces
IDDQ testing.

Additionally, the tool supports supplemental IDDQ test generation. The tool creates a
supplemental IDDQ test set when it generates an original set of IDDQ patterns based on the
pseudo stuck-at fault model. Before running the supplemental IDDQ process, you must first set
the fault type to IDDQ with the set_fault_type command.

During selective or supplemental IDDQ test generation, the tool classifies faults at the inputs of
sequential devices such as scan cells or non-scan cells as Blocked (BL) faults. This is because
the diversity of flip-flop and latch implementations means the pseudo stuck-at fault model
cannot reliably guide ATPG to create a good IDDQ test. In contrast, a simple combinational
logic gate has one common, fully complementary implementation (a NAND gate, for example,
has two parallel pFETs between its output and Vdd and two series nFETs between its output and
Vss), so the tool can more reliably declare pseudo stuck-at faults as detected. The switch level
implementation of a flip-flop varies so greatly that assuming a particular implementation is
highly suspect. The tool therefore takes a pessimistic view and reports coverage lower than it
actually is, because it is unlikely such defects will go undetected for all IDDQ patterns.

Using the ATPG tool, you can generate IDDQ patterns using several user-specified checks.
These checks can help ensure that the IDDQ test vectors do not increase IDDQ in the good
circuit. The following subsections describe IDDQ test generation, and user-specified checks in
more detail.

Determining When to Perform the Measures

The pre-existing external test set may or may not target IDDQ faults. For example, you can run
ATPG using the stuck-at fault type and then select patterns from this set for IDDQ testing. If the
pattern set does not target IDDQ faults, it will not contain statements that specify IDDQ
measurements. IDDQ test patterns must contain statements that tell the tester to make an IDDQ
measure. In the Text format, this IDDQ measure statement, or label, appears as follows:
measure IDDQ ALL <time>;

By default, the ATPG tool places these statements at the end of patterns (cycles) that can
contain IDDQ measurements. You can manually add these statements to patterns (cycles)
within the external pattern set.

When you want to select patterns from an external set, you must specify which patterns can
contain an IDDQ measurement. If the pattern set contains no IDDQ measure statements, you
can specify that the tools assume the tester can make a measurement at the end of each pattern
or cycle. If the pattern set already contains IDDQ measure statements (if you manually added

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these statements), you can specify that simulation should only occur for those patterns that
already contain an IDDQ measure statement, or label. To set this measurement information, use
the set_iddq_strobe command.

Generating an IDDQ Test Set

The following subsections discuss the basic IDDQ pattern generation process and provide an
example of a typical IDDQ pattern generation run.

Generating the Patterns

Prior to pattern generation, you may want to set up restrictions that the selection process must
abide by when choosing the best IDDQ patterns.
“Specifying Leakage Current Checks” on page 213 discusses these IDDQ restrictions. As with
any other fault type, you issue the create_patterns command within analysis mode. This
generates an internal pattern set targeting the IDDQ faults in the current list. You can turn
dynamic pattern compression on with the “set_atpg_compression On” command, targeting
multiple faults with a single pattern and resulting in a more compact test set.

Running ATPG for IDDQ

The following is an example of running ATPG for the IDDQ fault type.

Procedure
1. Invoke Tessent Shell, set the context to “patterns -scan,” read in the netlist, set up the
appropriate parameters for ATPG run, pass rules checking, and then enter analysis
mode.
...
SETUP> set_system_mode analysis

2. Set the fault type to IDDQ.

ANALYSIS> set_fault_type iddq

3. Specify the number of desirable IDDQ patterns.

ANALYSIS> set_atpg_limits -pattern_count 128

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4. Run ATPG, generating patterns that target the IDDQ faults in the current fault list.
Note that you could use the set_iddq_checks command prior to the ATPG run to place
restrictions on the generated patterns. You can also optionally issue the read_faults
command to add the target fault list at this point.
ANALYSIS> create_patterns -patterns_per_pass 1

The tool then adds all faults automatically.

Note that you did not need to specify which patterns could contain IDDQ measures with
set_iddq_strobe, as the generated internal pattern source already contains the appropriate
measure statements.
5. Save these IDDQ patterns into a file.
ANALYSIS> write_patterns iddq.pats

Specifying Leakage Current Checks

For CMOS circuits with pull-up or pull-down resistors or tri-state buffers, the good circuit
should have a nearly zero IDDQ current. The ATPG tool allows you to specify various IDDQ
measurement checks to ensure that the good circuit does not raise IDDQ current during the
measurement.
Use the set_iddq_checks command to specify these options.

By default, the tool does not perform IDDQ checks. Both ATPG and fault simulation processes
consider the checks you specify.

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Net Pair Identification with Calibre for Bridge Fault Test Patterns

Net Pair Identification with Calibre for Bridge

Fault Test Patterns
You can use the ATPG tool to identify a list of net pairs that should be targeted for the bridging
fault model and generated scan patterns. For complete information, refer to the “Bridge Fault
Test Pattern Generation Flow” section of the Calibre Solutions for Physical Verification manual
in your Calibre software tree or SupportNet.
The remainder of this section covers the specifics of the bridge fault model.

Four-Way Dominant Fault Model

The ATPG tool uses the 4-Way Dominant fault model to target net pairs for bridging. The
4-Way Dominant fault model works by driving a net to a dominant value (0 or 1) and ensuring
that the follower can be driven to the opposite value.
A simplified bridging fault model is adopted by defining the target net pairs as a set of stuck-at
faults. Each of the net pairs targeted by the bridge fault model is classified as dominant or
follower. A dominant net will force the follower net to take the same value as the dominant net
in the faulty circuit when the follower has an opposite logical value than the dominant net.

Let A and B be two gates with output signals sig_A and sig_B. When sig_A and sig_B are
bridged together, four faulty relationships can be defined:

• sig_A has dominant value of 0 (sig_A=0; sig_B s@0)

• sig_A has dominant value of 1 (sig_A=1; sig_B s@1)
• sig_B has dominant value of 0 (sig_B=0; sig_A s@0)
• sig_B has dominant value of 1 (sig_B=1; sig_A s@1)
By default, the ATPG tool targets each net pair using these four faulty relationships by
generating patterns which drive one net to the dominant value and observing the behavior of the
other (follower) net.

Top-level Bridging ATPG

This flow generates patterns for a list of net pairs generated by another tool such as Calibre. The
following input files are used in this flow:
• Gate-level Netlist
• ATPG Library
• Bridge Definition File or Calibre Server output file, here: from_calibre.sites

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Net Pair Identification with Calibre for Bridge Fault Test Patterns

The following commands are specific to bridging fault ATPG and should be issued once in
analysis mode:

set_fault_type bridge
// FAULT TYPE SET TO 4WAY_DOM
read_fault_sites from_calibre.sites
// ALL BRIDGE NET PAIRS ARE LOADED
add_faults all
// GENERATE A LIST OF FAULTS BASED ON THE LOADED BRIDGE NET PAIRS
create_patterns
// GENERATES PATTERNS
write_patterns bridge_patterns.ascii
exit

The set_fault_type command with the bridge argument is the only specification needed for
generation of patterns that target the 4-Way Dominant fault model. The read_fault_sites
command is used to load the list of net pairs that should be targeted for the bridging fault model.

Incremental Multi-Fault ATPG

The following flow generates patterns for a list of net pairs generated by another tool such as
Calibre. After the initial ATPG for bridging, you can generate patterns for other fault models
such as stuck-at. The flow fault simulates each pattern set for other fault models and generate
additional patterns to target the remaining faults. The following input files are used in this flow:
• Gate-level Netlist
• ATPG Library
• Bridge Definition File or Calibre Server output file, here: from_calibre.sites

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Net Pair Identification with Calibre for Bridge Fault Test Patterns

The following example shows generation of patterns for bridging faults followed by stuck-at
faults. The following commands should be issued once in analysis mode:

set_fault_type bridge
// fault type set to 4way_dom
read_fault_sites from_calibre.sites
// all bridge net pairs are loaded
create_patterns
// generate a list of faults based on the loaded bridge net pairs,
// then generate patterns
write_patterns bridge_patterns.ascii
set_fault_type stuck
add_faults all
// adds all stuck-at faults
read_patterns bridge_patterns.ascii
// load external patterns and add to internal patterns
simulate_patterns
// simulate bridge patterns for stuck-at faults
report_statistics
set_fault_protection on
// protect stuck-at faults that were detected by bridge patterns
reset_state
// remove bridge patterns that were effective in detecting stuck-at faults
create_patterns
// generate new patterns for remaining faults
write_patterns stuck_patterns.ascii
exit

The next example shows generation of patterns for stuck-at faults followed by bridging faults.
The following commands should be issued once in analysis mode:

set_fault_type stuck
create_patterns
// adds all stuck-at faults and generates patterns
write_patterns stuck_patterns.ascii
set_fault_type bridge
// fault type set to 4way_dom
read_fault_sites from_calibre.sites
// all bridge net pairs are loaded
read_patterns stuck_patterns.ascii
simulate_patterns
// simulate stuck-at patterns for bridge faults
write_faults bridge_faults.detected -class DT
// save detected bridge faults to file
create_patterns
reset_state
// remove stuck-at patterns that were effective in detecting bridge faults
read_faults bridge_faults.detected -retain
// load detected bridge faults and retain detection status
create_patterns
// generate new patterns for remaining undetected bridge faults
write_patterns patterns_bridge.ascii
exit

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The Bridge Parameters File

The Bridge Parameters File

The bridge definition is a text file which you automatically generate using the Extraction
Package.
This section provides information for interpreting the entries in the bridge parameters file.
Formatting Rules
Normally, you do not modify the generated bridge parameters file. If you do modify this file,
then you must adhere to the following syntax:

• Precede each line of comment text with a pair of slashes (//).

• Do not modify keywords. They can be in upper-or lowercase.
• Use an equal sign to define a value for a keyword.
• Enclose all string values in double quotation marks (" ").
• Bridge declarations must be enclosed in braces ({}).
• A semicolon (;) must separate each entry within the bridge declaration.
Parameters
Use the keywords described in the following table to create a bridge definition file. Keywords
cannot be modified and can be in upper- or lowercase.

Table 7-2. Bridge Definition File Keywords

Keyword (s) Usage Rules
VERSION Required. Used to specify the version of the bridge definition file and
must be declared before any bridge entries. Must be a real number or
integer written in non-scientific format starting with 1.1.
FAULT_TYPE Optional. Used to indicate what type of fault is declared by the FAULTS
keyword. Must be a string value enclosed in quotation marks (" ").
BRIDGE Required. Used to start a bridge entry.
NET1 Required. Identifies the net/pin pair for the bridge entry. Specifies the
NET2 regular or hierarchical net/pin pathname to gate(s). Use the following
guidelines when using these keywords:
• Declare either net or pin value pairs.
• Net/Pin pairs are the first two items declared in the bridge entry.
• Net/Pin pathnames are string values and should be enclosed in
quotation marks (" ").
• A net can be used in multiple bridge entries.
• Nets can be defined in any order.

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The Bridge Parameters File

Table 7-2. Bridge Definition File Keywords

Keyword (s) Usage Rules
FAULTS Optional. Provides a fault classification for each of the four components
of the 4-way bridge fault model. Four classifications must be specified in
a comma-separated list enclosed in braces ({}). You can use any of the
following 2-digit codes for each of the four components:
• UC (uncontrolled)
• UO (unobserved)
• DS (det_simulation)
• PU (posdet_untestable)
• PT (posdet_testable)
• TI (tied)
• BL (blocked)
• AU (ATPG_untestable)
• NF (directs the tool to nofault this component of the bridge)
Using fault classifications, you can filter and display the desired fault
types. For more information on fault classes and codes, see “Fault
Classes” on page 55.
NAME Optional. A unique string, enclosed in quotation marks (" "), that specifies
a name for the bridge.
DISTANCE1 Optional. Real number that specifies the distance attribute in microns
(um) for the bridge entry.
PARALLEL_RUN1 Optional. Real number that specifies the parallel_run attribute in microns
(um) for the bridge entry.
LAYER1,2 Optional. String that specifies the layer attribute for the bridge entry. Must
be enclosed in quotation marks (" ").
WEIGHT1,2 Optional. Real number that specifies the weight attribute for the bridge
entry.
TYPE1,2 Optional. 3-, 4-, or 5-character code that specifies a type identification for
the bridge entry. Must be enclosed in quotation marks (" "). Use one of the
following codes:
• S2S — Side-to-side
• SW2S — Side-Wide-to-Side: same as S2S but at least one of the two
signal lines is a wide metal line
• S2SOW — Side-to-Side-Over-Wide: same as S2S, but the bridge is
located over a wide piece of metal in a layer below
• C2C — Corner-to-Corner
• V2V — Via-to-Via: an S2S for vias
• VC2VC — Via-Corner-to-Via-Corner
• EOL — End-of-Line: the end head of a line faces another metal line
1. This keyword is not used by the software to define any specific value. However, you can use it to specify a
value which can be filtered on by the read_fault_sites command.
2. You can display a histogram of the number of equivalent classes of bridges based on either their layer, weight,
or type attribute using the report_fault_sites or report_faults command.

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Example
The following example shows the format of the bridge definition file. The keywords used in this
file are case insensitive.

// Any string after // is a comment and will be ignored by the tool.

VERSION 1.1// An optional item that declares the version of
// the bridge definition file. VERSION must be
// defined before any bridge entry. The version
// number starts from 1.0.
FAULT_TYPE = BRIDGE_STATIC_4WAY_DOM
// An optional header that indicates the fault type
// to be declared by keywords FAULTS in
// bridge body. Currently, the only valid fault type
// is BRIDGE_STATIC_4WAY_DOM
BRIDGE {// Define the bridge entry body and can be repeated
// as many times as necessary.
NET1 = NET_NAME; // Defines the first net name.
NET2 = NET_NAME; // Defines the second net name.
// NET1 and NET2 must be defined in each bridge
// entry and they must be declared before any other
// items defined in the bridge entry body.
FAULTS = {FAULT_CATEGORY_1, FAULT_CATEGORY_2, FAULT_CATEGORY_3,
FAULT_CATEGORY_4};
// An optional item that defines the fault
// classes for each of the four faulty
// relationships. All four fault categories must be
// declared and must be in the order shown in the
// previous section. Examples of fault classes are
// UC, UO, DS, DI, PU, // PT, TI, BL and AU. A new
// fault class NF (No Fault) can be used to exclude
// any specific faulty relationship for ATPG.
NAME = STRING; // An optional string specifying the name
// of the bridge.
PARALLEL_RUN = floating number; // An optional item specifying the
// parallel run length of the nets of the bridge.
DISTANCE = floating number; // An optional item specifying the distance
// of the nets of the bridge.
WEIGTH = floating number; // An optional item specifying a weight assigned
// to the bridge.
LAYER = LAYER_NAME; // An optional item specifying the name of the
// layer the bridge is in.
} // End of bridge body

The items DISTANCE, PARALLEL_RUN, WEIGTH, and LAYER are collectively referred to
as attributes of the bridge. For ATPG purposes, these attributes are ignored, and by default not
reported. You have to specify the -attribute option to according commands to see them. The unit
of length used for DISTANCE and PARALLEL_RUN is um (10^-6) meters.

Creating a Delay Test Set

Delay, or “at-speed” tests in the ATPG tool are of two types: transition delay and path delay.
Figure 7-10 shows a general flow for creating a delay pattern set.

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Figure 7-10. Flow for Creating a Delay Test Set

Choose Fault Type

(Transition or
Path Delay)

Define Capture
Procedures
(Optional)

Analyze Coverage

Create Patterns

Your process may be different and it may involve multiple iterations through some of the steps,
based on your design and coverage goals. This section describes these two test types in more
detail and how you create them using the ATPG tool. The following topics are covered:

Creating a Transition Delay Test Set . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 220

Creating a Path Delay Test Set . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 234
At-Speed Test Using Named Capture Procedures . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 245
Support for On-Chip Clocks (PLLs). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 246
Mux-DFF Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 253
Generating Test Patterns for Different Fault Models and Fault Grading . . . . . . . . . . . . . . 263

Creating a Transition Delay Test Set

The tool can generate patterns to detect transition faults.
“At-Speed Testing and the Transition Fault Model” on page 40 introduced the transition fault
model. Transition faults model gross delays on gate terminals (or nodes), allowing each
terminal to be tested for slow-to-rise or slow-to-fall behavior. The defects these represent may
include things like partially conducting transistors or interconnections.

Figure 7-11 illustrates the six potential transition faults for a simple AND gate. These are
comprised of slow-to-rise and slow-to-fall transitions for each of the three terminals. Because a
transition delay test checks the speed at which a device can operate, it requires a two cycle test.
First, all the conditions for the test are set. In the figure, A and B are 0 and 1 respectively. Then

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a change is launched on A, which should cause a change on Y within a pre-determined time. At

the end of the test time, a circuit response is captured and the value on Y is measured. Y might
not be stuck at 0, but if the value of Y is still 0 when the measurement is taken at the capture
point, the device is considered faulty. The ATPG tool automatically chooses the launch and
capture scan cells.

Figure 7-11. Transition Delay

A
Y
B AND

Predetermined
test time

Y Measure/Capture

Launch
Y
Fail

Transition Fault Detection

To detect transition faults, the following conditions must be met:
• The corresponding stuck-at fault must be detected.
• Within a single previous cycle, the node value must be at the opposite value than the
value detected in the current cycle.
Figure 7-12 depicts the launch and capture events of a small circuit during transition testing.
Transition faults can be detected on any pin.

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Figure 7-12. Transition Launch and Capture Events

Capture
Launch 0-1 Event
Event PI or 0-1 (measure PO)
(force PI) scan cell AND
X-1
X-0
X-0 NOR
X-0
PO or
X-1 AND scan cell

To detect a transition fault, a typical pattern includes the events in Figure 7-13.

Figure 7-13. Events in a Broadside Pattern

1. Load scan chains
2. Force primary inputs
3. Pulse clock
4. Force primary inputs
5. Measure primary outputs
6. Pulse clock
7. Unload scan chains

This is a clock sequential pattern, commonly referred to as a broadside pattern. It has basic
timing similar to that shown in Figure 7-14 and is the kind of pattern the ATPG tool attempts to
create by default when the clock-sequential depth (the depth of non-scan sequential elements in
the design) is two or larger. You specify this depth with the set_pattern_type command’s
-Sequential switch. The default setting of this switch upon invocation is 0, so you would need to
change it to at least 2 to enable the tool to create broadside patterns.

Typically, this type of pattern eases restrictions on scan enable timing because of the relatively
large amount of time between the last shift and the launch. After the last shift, the clock is
pulsed at speed for the launch and capture cycles.

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Figure 7-14. Basic Broadside Timing

Cycles: Shift Shift Dead Clock sequential Shift

Cycle
(optional) Launch Capture

clk

scan_en

If it fails to create a broadside pattern, the tool next attempts to generate a pattern that includes
the events shown in Figure 7-15.

Figure 7-15. Events in a Launch Off Shift Pattern

1. Init_force primary inputs
2. Load scan chains
3. Force primary inputs
4. Measure primary outputs
5. Pulse clock
6. Unload scan chains

In this type of pattern, commonly referred to as a launch off last shift or just launch off shift
pattern, the transition occurs because of the last shift in the load scan chains procedure (event
#2) or the forcing of the primary inputs (event #3). Figure 7-16 shows the basic timing for a
launch that is triggered by the last shift.

Figure 7-16. Basic Launch Off Shift Timing

Cycles: Shift Shift Last Shift Capture Shift

Launch Capture

clk

scan_en

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If your design cannot support this requirement, you can direct the tool not to create launch off
shift patterns by including the -No_shift_launch switch when specifying transition faults with
the set_fault_type command.

Note
Launch off shift patterns require the scan enable signal for mux-scan designs to transition
from shift to capture mode at speed. Therefore, the scan enable must be globally routed
and timed similar to a clock. Also, because launch off shift patterns detect a certain
number of faults in non-functional paths as well as in the scan path, the test coverage
reported is usually higher than for broadside patterns which do not capture faults in non-
functional paths.

The following are example commands you could use at the command line or in a dofile to
generate broadside transition patterns:

SETUP> add_input_constraintsscan_en -c0 #force for launch & capture.

SETUP> set_output_masks on #do not observe primary outputs.
SETUP> set_transition_holdpi on #freeze primary input values.
ANALYSIS> set_fault_type transition -no_shift_launch #prohibit launch off last shift.
ANALYSIS> set_pattern_type -sequential 2 #sequential depth depends on design.
ANALYSIS> create_patterns

To create transition patterns that launch off the last shift, use a sequence of commands similar to
this:

SETUP> set_output_masks on #don’t observe primary outputs.

...
ANALYSIS> set_fault_type transition
ANALYSIS> set_pattern_type -sequential 0 #prevent broadside patterns.
ANALYSIS> create_patterns

The following is a list of related commands:

• set_abort_limit — Specifies the abort limit for the test pattern generator.
• set_fault_type — Specifies the fault model for which the tool develops or selects ATPG
patterns.
• set_pattern_type — Specifies the type of test patterns the ATPG simulation run uses.

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Basic Procedure for Generating a Transition Test

Set
Use the following basic procedure for generating a transition test set:
1. Perform circuit setup tasks.
2. Constrain the scan enable pin to its inactive state. For example:
SETUP> add_input_constraints scan_en -c0

3. Set the sequential depth to two or greater (optional):

SETUP> set_pattern_type -sequential 2

4. Enter analysis system mode. This triggers the tool’s automatic design flattening and
rules checking processes.
SETUP> set_system_mode analysis

5. Set the fault type to transition:

ANALYSIS> set_fault_type transition

6. Add faults to the fault list:

ANALYSIS> add_faults -all

7. Run test generation:

ANALYSIS> create_patterns

Timing for Transition Delay Tests

This section describes how the timing works for transition delay tests. Basically, the tool obtains
the timing information from the test procedure file. This file describes the scan circuitry
operation to the tool. You can create scan circuitry manually, or use Tessent Scan to create the
scan circuitry for you after it inserts scan circuitry into the design.
The test procedure file contains cycle-based procedures and timing definitions that tell the
ATPG tool how to operate the scan structures within a design. For more information, refer to
“Test Procedure File” in the Tessent Shell User’s Manual.

Within the test procedure file, timeplates are the mechanism used to define tester cycles and
specify where all event edges are placed in each cycle. As shown conceptually in Figure 7-14
for broadside testing, slow cycles are used for shifting (load and unload cycles) and fast cycles
for the launch and capture. Figure 7-17 shows the same diagram with example timing added.

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Figure 7-17. Broadside Timing Example

Cycles: Shift Shift Dead Clock sequential Shift

Cycle
Launch Capture

clk

tp_slow tp_slow tp_fast tp_fast tp_fast tp_slow

scan_en

400 ns 400 ns 40 ns 40 ns 40 ns 400 ns

This diagram now shows 400 nanosecond periods for the slow shift cycles defined in a
timeplate called tp_slow and 40 nanosecond periods for the fast launch and capture cycles
defined in a timeplate called tp_fast.

The following are example timeplates and procedures that would provide the timing shown in
Figure 7-17. For brevity, these excerpts do not comprise a complete test procedure. Normally,
there would be other procedures as well, like setup procedures.

timeplate tp_slow = timeplate tp_fast =

force_pi 0; force_pi 0;
measure_po 100; measure_po 10;
pulse clk 200 100; pulse clk 20 10;
period 400; period 40;
end; end;

procedure load_unload = procedure capture =

scan_group grp1; timeplate tp_fast;
timeplate tp_slow; cycle =
cycle = force_pi;
force clk 0; measure_po;
force scan_en 1; pulse_capture_clock;
end; end;
apply shift 127; end;
end;

procedure shift = procedure clock_sequential =

timeplate tp_slow; timeplate tp_fast;
cycle = cycle =
force_sci; force_pi;
measure_sco; pulse_capture_clock;
pulse clk; pulse_read_clock;
end; pulse_write_clock;
end; end;
end;

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In this example, there are 40 nanoseconds between the launch and capture clocks. If you want to
create this same timing between launch and capture events, but all your clock cycles have the
same period, you can skew the clock pulses within their cycle periods—if your tester can
provide this capability. Figure 7-18 shows how this skewed timing might look.

Figure 7-18. Launch Off Shift (Skewed) Timing Example

Cycles: Shift Shift Last Shift Capture Shift

Launch Capture

clk

tp_late tp_late tp_late tp_early tp_late

scan_en

100 ns 100 ns 100 ns 100 ns 100 ns

The following timeplate and procedure excerpts show how skewed launch off shift pattern
events might be managed by timeplate definitions called tp_late and tp_early, in a test
procedure file:

Note
For brevity, these excerpts do not comprise a complete test procedure. The shift
procedure is not shown and normally there would be other procedures as well, like setup
procedures.

timeplate tp_late = timeplate tp_early =

force_pi 0; force_pi 0;
measure_po 10; measure_po 10;
pulse clk 80 10; pulse clk 20 10;
period 100; period 100;
end; end;

procedure load_unload = procedure capture =

scan_group grp1; timeplate tp_early;
timeplate tp_late; cycle =
cycle = force_pi;
force clock 0; measure_po;
force scan_en 1; pulse_capture_clock;
end; end;
apply shift 7; end;
end;

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By moving the clock pulse later in the period for the load_unload and shift cycles and earlier in
the period for the capture cycle, the 40 nanosecond time period between the launch and capture
clocks is achieved.

Preventing Pattern Failures Due to Timing

Exception Paths
Prior to ATPG, you can perform a timing optimization on a design using a static timing analysis
(STA) tool. This process also defines timing exception paths consisting of one of the following:
• False Path — A path that cannot be sensitized in the functional mode of operation (the
STA tool ignores these paths when determining the timing performance of a circuit).
• Multicycle Path — A path with a signal propagation delay of more than one clock
cycle. Figure 7-19 shows path P1 beginning at flip-flop U1, going through gates G1, G3,
G5, and G6, and ending at flip-flop U5. This path has a total propagation delay longer
than the clock period.

Figure 7-19. Multicycle Path Example

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You should evaluate the effect of timing exception paths for any sequential pattern containing
multiple at-speed capture clock pulses, either from the same clock or from different clocks. This
includes the following pattern types:

• Clock sequential (broadside transition patterns and stuck-at patterns)

• RAM sequential
• Path delay
The ATPG tool automatically evaluates false and multicycle paths before creating the test
patterns. When simulating patterns during ATPG, the tool identifies incorrect capture responses
due to false paths and masks them (modifies them to X) in the resultant patterns. For multicycle
paths, expected values are accurately simulated and stored in the pattern set.

Types of Timing Exception Path Violations

Most designs hold data and control inputs constant for specified time periods before and after
any clock events. In this context, the time period before the clock event is the setup time, and
the time period after the clock event is hold time.
Figure 7-20 illustrates how setup and hold time exceptions can produce the following timing
exception path violations: Setup Time Violations and Hold Time Violations.

Figure 7-20. Setup Time and Hold Time Violations

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Setup Time Violations

A timing exception path with setup time violation does not meet the setup time requirements.
This type of violation can affect test response for at-speed test patterns.

Hold Time Violations

A timing exception path with hold time violation does not meet the hold time requirements.
This type of violation can affect test response of any test pattern and usually occurs across
different clock domains. The ATPG tool simulates hold time false paths for the following
timing exception paths:
• False paths you manually specify with add_false_paths command using the -hold
switch—see “Manually Specifying False Paths for Hold Time Violation Checks”.
• False paths between two clock domains.
• Multicycle paths with a path multiplier of 0 (zero).
Figure 7-21 illustrates a hold time violation occurring across different clock domains.

Figure 7-21. Across Clock Domain Hold Time Violation

Figure 7-21 shows the false paths from the clock domain Clk1 to the clock domain Clk2. In this
figure, when a test pattern has a clock sequence of simultaneously pulsing both Clk1 and Clk2,
there can be a hold time violation from the flip-flop U1 or U2 to the flip-flop U3.

In Figure 7-21, pulsing clocks Clk1 and Clk2 simultaneously places the new value 1 at the D
input of flip-flop U1, creating a rising transition on the flip-flop U1’s Q output. This transition
sensitizes the path from the flip-flop U1 to the flip-flop U3. If the clock Clk2 arrives late at the
flip-flop U3 the new value 0 is captured at the flip-flop U3 instead of the old value 1.

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Reading Timing Exception Paths from an SDC File

STA tools typically provide a command to write out the false and multicycle path information
identified during the STA process into a file or script in Synopsys Design Constraint (SDC)
format. For example, the Synopsys PrimeTime tool has the write_sdc command. You can also
read in both the -setup and -hold SDC information.

Tip: Before using the write_sdc command, use the PrimeTime transform_exceptions
command to eliminate redundant, overridden or invalid exceptions.

If you can get the information into an SDC file, you can use the read_sdc command to read in
the false path definitions from the file. Note that the tool does not infer false and multicycle
paths from the delays in an SDF file.

The following is an example of the use of this command in a typical command sequence for
creating broadside transition patterns:

<Define clocks, scan chains, constraints, and so on>

ANALYSIS> set_fault_type transition -no_shift_launch
ANALYSIS> set_pattern_type -sequential 2
ANALYSIS> read_sdc my_sdc_file
...
ANALYSIS> create_patterns

If you already have a pattern set for your design and want to see the effect of adding the false
and multicycle path information, the command sequence is slightly different:

<Define clocks, scan chains, constraints, and so on>

ANALYSIS> read_sdc my_sdc_file
ANALYSIS> add_faults -all
ANALYSIS> read_patterns my_patterns.ascii
ANALYSIS> simulate_patterns
ANALYSIS> report_statistics

As a result of simulating the patterns using the false and multicycle path information, the
patterns read in from the external pattern file will now be stored in the tool’s internal pattern set,
with some capture values in the internal patterns changed to “X”. These changed values
represent masking the tool applied to adjust for false and multicycle path effects. The Xs will
increase the number of undetected faults slightly and lower the test coverage; however, the
patterns will be more correct and will eliminate mismatches related to those capture values.

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Note
You can save the patterns that include the false and/or multicycle path information as
usual. For example:
ANALYSIS> write_patterns my_patterns_falsepaths.v -verilog
ANALYSIS> write_patterns my_patterns_falsepaths.ascii -ascii

Ensuring the SDC File Contains Valid SDC Information

The read_sdc command reads an SDC file and parses it, looking for supported commands that
are relevant for ATPG. ATPG tools support the following SDC commands and arguments:
• all_clocks
• create_clock [-name clock_name]
• create_generated_clock [-name clock name]
• get_clocks
• get_generated_clocks
• get_pins
• get_ports
• set_case_analysis value port_or_pin_list
• set_clock_groups
• set_disable_timing [-from from_pin_name] [-to to_pin_name] cell_pin_list
• set_false_path [-setup] [-hold] [-from from_list] [-to to_list] [-through through_list]
• set_hierarchy_separator
• set_multicycle_path [-setup] [-hold] [-from from_list] [-to to_list]
[-through through_list]

Note
For complete information on these commands and arguments, refer to your SDC
documentation.

To avoid problems extracting the timing exception paths from the SDC specifications, the best
results are obtained when the SDC file is written out by the PrimeTime static timing analysis
(STA) tool. Mentor Graphics highly recommends that within PrimeTime you use the
transform_exceptions command before saving the SDC file. This command removes any
redundant, invalid and overridden path exceptions. Then use the write_sdc command to save the
updated information to a file you can use with read_sdc.

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The following summarizes the recommended steps:

1. Read the original SDC file(s) in PrimeTime.

2. Execute the transform_exceptions command in PrimeTime.
3. Execute the write_sdc command in PrimeTime, to write out a valid SDC file.
4. In the ATPG tool, use the read_sdc command to read the SDC file written out in step 3.
5. Generate at-speed patterns.

Defining False Paths Manually

Alternatively and using the add_false_paths command, you can manually specify false path
definitions for both specifying false paths for setup time violation checks and specifying false
paths for hold time violation checks.
When performing this operation, you must be specific and accurate when specifying false (and
multicycle) paths during the STA process, and to maintain the same accuracy when using the
add_false_paths command.

For example, defining non-specify false path definition with the following command:

add_false_paths -from U3

would result in the propagation of the false path out through the design in an effect cone
encompassing all possible paths from that node. Figure 7-22 shows an illustration of this.

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Figure 7-22. Effect Cone of a Non-specific False Path Definition

Manually Specifying False Paths for Setup Time Violation Checks

By default, the tool evaluates setup time violations for the false paths you manually specify with
the add_false_paths command.

Manually Specifying False Paths for Hold Time Violation Checks

For hold time violations, you identify the path with the add_false_paths command and also
specify the -hold switch. The following are useful commands for working with false paths:
• add_false_paths — Specifies one or more false paths.
• delete_false_paths — Deletes the specified false path definitions.
• report_false_paths — Displays the specified false path definitions.
• delete_multicycle_paths — Deletes the specified multicycle path definitions.
• report_multicycle_paths — Displays the specified multicycle path definitions.

Creating a Path Delay Test Set

The ATPG tool can generate patterns to detect path delay faults. These patterns determine if
specific user-defined paths operate correctly at-speed.

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“At-Speed Testing and the Path Delay Fault Model” on page 41 introduced the path delay fault
model. You determine the paths you want tested (most people use a static timing analysis tool to
determine these paths), and list them in an ASCII path definition file you create. You then load
the list of paths into the tool. “The Path Definition File” on page 240 describes how to create
and use this file.

Path Delay Fault Detection

Path delay testing requires a logic value transition, which implies two events need to occur to
detect a fault. These events include a launch event and a capture event. Typically, both the
launch and capture occur at scan cells, but they can occur at RAMs or, depending on the timing
and precision of the ATE to test around a chip’s I/O, at PIs and POs.
Figure 7-23 depicts the launch and capture events of a small circuit during a path delay test.

Figure 7-23. Path Delay Launch and Capture Events

Capture
Launch 0-1 Event
Event PI or 0-1 (measure PO)
(force PI) scan cell AND
1-1 1-0
0-0 NOR
1-0
AND PO or
1-1 scan cell

Path delay patterns are a variant of clock-sequential patterns. A typical pattern to detect a path
delay fault includes the following events:

1. Load scan chains

2. Force primary inputs
3. Pulse clock (to create a launch event for a launch point that is a state element)
4. Force primary inputs (to create a launch event for a launch point that is a primary input)
5. Measure primary outputs (to create a capture event for a capture point that is a primary
output)
6. Pulse clock (to create a capture event for a capture point that is a state element)
7. Unload scan chains
The additional force_pi/pulse_clock cycles may occur before or after the launch or capture
events. The cycles depend on the sequential depth required to set the launch conditions or
sensitize the captured value to an observe point.

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Note
Path delay testing often requires greater depth than for stuck-at fault testing. The
sequential depths that the tool calculates and reports are the minimums for stuck-at
testing.

To get maximum benefit from path delay testing, the launch and capture events must have
accurate timing. The timing for all other events is not critical.

The ATPG tool detects a path delay fault with either a robust test, a non-robust test, or a
functional test. If you save a path delay pattern in ASCII format, the tool includes comments in
the file that indicate which of these three types of detection the pattern uses. Robust detection
occurs when the gating inputs used to sensitize the path are stable from the time of the launch
event to the time of the capture event. Robust detection keeps the gating of the path constant
during fault detection and thus, does not affect the path timing. Because it avoids any possible
reconvergent timing effects, it is the most desirable type of detection and for that reason is the
approach the ATPG tool tries first. The tool, however, cannot use robust detection on many
paths because of its restrictive nature and if it is unable to create a robust test, it will
automatically try to create a non-robust test. The application places faults detected by robust
detection in the DR (det_robust) fault class.

Figure 7-24 gives an example of robust detection for a rising-edge transition within a simple
path. Notice that, due to the circuitry, the gating value at the second OR gate was able to retain
the proper value for detection during the entire time from launch to capture events.

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Figure 7-24. Robust Detection Example

Initial State
Launch Point
Capture Point
1
0 X
AND 1 1
1 OR X
1 0
0 1 0
OR
1

After Transition
Launch Point
Capture Point
0
1 X
AND 0 0
1 0 OR X
1
1
1
1
OR
0
Gating Value Constant
During Transition

Non-robust detection does not require constant values on the gating inputs used to sensitize the
path. It only requires the proper gating values at the time of the capture event. The ATPG tool
places faults detected by non-robust detection in the DS (det_simulation) fault class.

Figure 7-25 gives an example of non-robust detection for a rising-edge transition within a
simple path.

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Figure 7-25. Non-robust Detection Example

Initial State
Launch Point
Capture Point
1
0X
AND 1 1
1 1 OR X
1
0 1
AND 0
1

After Transition
Launch Point
Capture Point
0
1X
AND 0 0
1 0 OR X
1
1 0
AND 1
1 Gating Value Changed
During Transition

Notice that due to the circuitry, the gating value on the OR gate changed during the 0 to 1
transition placed at the launch point. Thus, the proper gating value was only at the OR gate at
the capture event.

Functional detection further relaxes the requirements on the gating inputs used to sensitize the
path. The gating of the path does not have to be stable as in robust detection, nor does it have to
be sensitizing at the capture event, as required by non-robust detection. Functional detection
requires only that the gating inputs not block propagation of a transition along the path. The tool
places faults detected by functional detection in the det_functional (DF) fault class.

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Figure 7-26 gives an example of functional detection for a rising-edge transition within a simple
path. Notice that, due to the circuitry, the gating (off-path) value on the OR gate is neither
stable, nor sensitizing at the time of the capture event. However, the path input transition still
propagates to the path output.

Figure 7-26. Functional Detection Example

Initial State
Launch Point
Capture Point
0
0X
AND 0 0
1 0 OR X
1
0 1 1 0
AND
1
1

After Transition
Launch Point
Capture Point
1
1 X
AND 1 1
1 1 OR X
1
1 0
AND 0 1
1 1 Gating Value Changed
During Transition

• add_ambiguous_paths — Specifies the number of paths the tool should select when
encountering an ambiguous path.
• analyze_fault — Analyzes a fault, including path delay faults, to determine why it was
not detected.
• delete_fault_sites — Deletes paths from the internal path list.
• read_fault_sites — Loads in a file of path definitions from an external file.
• report_fault_sites — Reports information on paths in the path list.
• report_statistics — Displays simulation statistics, including the number of detected
faults in each fault class.
• set_pathdelay_holdpi — Sets whether non-clock primary inputs can change after the
first pattern force, during ATPG.
• write_fault_sites — Writes information on paths in the path list to an external file.

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The Path Definition File

In an external ASCII file, you must define all paths that you want tested in the test set. For each
path, you must specify:
• Path_name — A unique name you define to identify the path.
• Path_definition — The topology of the path from launch to capture point as defined by
an ordered list of pin pathnames. Each path must be unique.
The ASCII path definition file has several syntax requirements. The tools ignore as a comment
any line that begins with a double slash (//) or pound sign (#). Each statement must be on its
own line. The four types of statements include:

• Path — A required statement that specifies the unique pathname of a path.

• Condition — An optional statement that specifies any conditions necessary for the
launch and capture events. Each condition statement contains two arguments: a full pin
pathname for either an internal or external pin, and a value for that pin. Valid pin values
for condition statements are 0, 1, or Z. Condition statements must occur between the
path statement and the first pin statement for the path.
• Transition_condition — An optional statement that specifies additional transitions
required in the test pattern. Each transition_condition statement contains two arguments:
a full pin pathname for either an internal or external pin and a direction.
Transition_condition statements must occur between the path statement and the first pin
statement for the path.
The direction can be one of the following: rising, falling, same, or opposite. Rising and
falling specify that a rising edge and falling edge, respectively, are required on the
specified pin at the same time as launching a transition into the first pin of the path.
Same specifies for the tool to create a transition in the same direction as the one on the
first pin in the path definition. Opposite creates a transition in the opposite direction.
Figure 7-27 shows an example where a transition_condition statement could be
advantageous.

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Figure 7-27. Example Use of Transition_condition Statement.

0-1
PI or 0-1 0-1
scan cell 0-1 PO or
AND scan cell
1 - 1 (tool’s preference) Capture
Launch
Event 0 - 1 (your preference) Event
(force PI) (measure PO)
d To other circuit
elements requiring
a 0-1 transition

A defined path includes a 2-input AND gate with one input on the path, the other
connected to the output of a scan cell. For a robust test, the AND gate’s off-path or
gating input needs a constant 1. The tool, in exercising its preference for a robust test,
would try to create a pattern that achieved this. Suppose however that you wanted the
circuit elements fed by the scan cell to receive a 0-1 transition. You could add a
transition_condition statement to the path definition, specifying a rising transition for
the scan cell. The path capture point maintains a 0-1 transition, so remains testable with
a non-robust test, and you also get the desired transition for the other circuit elements.
• Pin — A required statement that identifies a pin in the path by its full pin pathname. Pin
statements in a path must be ordered from launch point to capture point. A “+” or “-”
after the pin pathname indicates the inversion of the pin with respect to the launch point.
A “+” indicates no inversion (you want a transition identical to the launch transition on
that pin), while a “-” indicates inversion (you want a transition opposite the launch
transition).

Note
If you use “+” or “-” in any pin statement, you must include a “+” for the launch point.
The polarity of the launch transition must always be “+”.

You must specify a minimum of two pin statements, the first being a valid launch point
(primary input or data output of a state element or RAM) and the last being a valid
capture point (primary output, data or clk input of a state element, or data input of a
RAM). The current pin must have a combinational connectivity path to the previous pin
and the edge parity must be consistent with the path circuitry. If a statement violates
either of these conditions, the tool issues an error. If the path has edge or path ambiguity,
it issues a warning.
Paths can include state elements (through data or clock inputs), but you must explicitly
name the data or clock pins in the path. If you do not, the tool does not recognize the
path and issues a corresponding message.

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• End — A required statement that signals the completion of data for the current path.
Optionally, following the end statement, you can specify the name of the path. However,
if the name does not match the pathname specified with the path statement, the tool
issues an error.
The following shows the path definition syntax:

PATH <pathname> =
CONDition <pin_pathname> <0|1|Z>;
TRANsition_condition <pin_pathname> <Rising|Falling|Same|Opposite>;
PIN <pin_pathname> [+|-];
PIN <pin_pathname> [+|-];
...
PIN <pin_pathname> [+|-];
END [pathname];

The following is an example of a path definition file:

PATH "path0" =
PIN /I$6/Q + ;
PIN /I$35/B0 + ;
PIN /I$35/C0 + ;
PIN /I$1/I$650/IN + ;
PIN /I$1/I$650/OUT - ;
PIN /I$1/I$951/I$1/IN - ;
PIN /I$1/I$951/I$1/OUT + ;
PIN /A_EQ_B + ;
END ;
PATH "path1" =
PIN /I$6/Q + ;
PIN /I$35/B0 + ;
PIN /I$35/C0 + ;
PIN /I$1/I$650/IN + ;
PIN /I$1/I$650/OUT - ;
PIN /I$1/I$684/I1 - ;
PIN /I$1/I$684/OUT - ;
PIN /I$5/D - ;
END ;
PATH "path2" =
PIN /I$5/Q + ;
PIN /I$35/B1 + ;
PIN /I$35/C1 + ;
PIN /I$1/I$649/IN + ;
PIN /I$1/I$649/OUT - ;
PIN /I$1/I$622/I2 - ;
PIN /I$1/I$622/OUT - ;
PIN /A_EQ_B + ;
END ;
PATH "path3" =
PIN /I$5/QB + ;
PIN /I$6/TI + ;
END ;

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You use the read_fault_sites command to read in the path definition file. The tool loads the
paths from this file into an internal path list. You can add to this list by adding paths to a new
file and re-issuing the read_fault_sites command with the new filename.

Path Definition Checking

The ATPG tool checks the points along the defined path for proper connectivity and to
determine if the path is ambiguous. Path ambiguity indicates there are several different paths
from one defined point to the next.
Figure 7-28 indicates a path definition that creates ambiguity.

Figure 7-28. Example of Ambiguous Path Definition

Gate3 Gate5

Gate1 Gate2 Gate6 Gate7

Gate4

Defined Points

In this example, the defined points are an input of Gate2 and an input of Gate7. Two paths exist
between these points, thus creating path ambiguity. When the ATPG tool encounters this
situation, it issues a warning message and selects a path, typically the first fanout of the
ambiguity. If you want the tool to select more than one path, you can specify this with the
add_ambiguous_paths command.

During path checking, the tool can also encounter edge ambiguity. Edge ambiguity occurs when
a gate along the path has the ability to either keep or invert the path edge, depending on the
value of another input of the gate. Figure 7-29 shows a path with edge ambiguity due to the
XOR gate in the path.

Figure 7-29. Example of Ambiguous Path Edges

Path Edges

/
Gate XOR

0/1

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The XOR gate in this path can act as an inverter or buffer of the input path edge, depending on
the value at its other input. Thus, the edge at the output of the XOR is ambiguous. The path
definition file lets you indicate edge relationships of the defined points in the path. You do this
by specifying a “+” or “-” for each defined point, as was previously described in “The Path
Definition File” on page 240.

Basic Procedure for Generating a Path Delay Test Set

Use the following basic procedure to generate a path delay test set:
1. Perform circuit setup tasks.
2. Constrain the scan enable pin to its inactive state. For example:
SETUP> add_input_constraints scan_en -c0

3. (Optional) Turn on output masking.

SETUP> set_output_masks on

4. Add nofaults <x, y, z>

5. Set the sequential depth to two or greater:
SETUP> set_pattern_type -sequential 2

6. Enter analysis system mode. This triggers the tool’s automatic design flattening and
rules checking processes.
7. Set the fault type to path delay:
ANALYSIS> set_fault_type path_delay

8. Write a path definition file with all the paths you want to test. “The Path Definition File”
on page 240 describes this file in detail. If you want, you can do this prior to the session.
You can only add faults based on the paths defined in this file.
9. Load the path definition file (assumed for the purpose of illustration to be named
path_file_1):
ANALYSIS> read_fault_sites path_file_1

10. Specify any ambiguous paths you want the tool to add to its internal path list. The
following example specifies to add all ambiguous paths up to a maximum of 4.
ANALYSIS> add_ambiguous_paths -all -max_paths 4

11. Define faults for the paths in the tool’s internal path list:
ANALYSIS> add_faults -all

This adds a rising edge and falling edge fault to the tool’s path delay fault list for each
defined path.

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12. Perform an analysis on the specified paths and delete those the analysis proves are un-
sensitizable:
ANALYSIS> delete_fault_sites -unsensitizable_paths

13. Run test generation:

ANALYSIS> create_patterns

Path Delay Testing Limitations

Path delay testing does not support the following:
• RAMs Within a Specified Path — A RAM as a launch point is supported only if the
launch point is at the RAM’s output. A RAM as a capture point is supported only if the
capture point is at the RAM’s input.
• Paths Starting at a Combinationally Transparent Latch — A combinationally
transparent latch as a capture point is supported only if the capture point is at the latch’s
input.
• Path Starting and/or Ending at ROM — You should model ROM as a read-only
CRAM primitive (that is, without any _write operation) to enable the tool to support
path delay testing starting and/or ending at ROM.

At-Speed Test Using Named Capture Procedures

To create at-speed test patterns for designs with complicated clocking schemes, you may need
to specify the actual launch and capture clocking sequences. For example, in an LSSD type
design with master and slave clocks, the number and order of clock pulses might need to be
organized in a specific way. You can do this using a named capture procedure in the test
procedure file.
A named capture procedure is an optional procedure, with a unique name, used to define
explicit clock cycles. Named capture procedures can be used for generating stuck-at, path delay,
and broadside transition patterns, but not launch off shift transition patterns. You can create
named capture procedures using the create_capture_procedures command and then write out the
procedures using the write_procfile command. You can also manually create or edit named
capture procedures using an external editor if needed. For information on manually creating and
editing named capture procedures, see the “Rules for Creating and Editing Named Capture
Procedures” section in the Tessent Shell User’s Manual.

When the test procedure file contains named capture procedures, the ATPG tool generates
patterns that conform to the waveforms described by those procedures. Alternatively, you can
use the set_capture_procedures command to disable a subset of the named capture procedures,
and only the enabled subset is used. For example, you might want to exclude named capture
procedures that are unable to detect certain types of faults during test pattern generation.

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You can have multiple named capture procedures within one test procedure file in addition to
the default capture procedure the file typically contains. Each named capture procedure must
reflect clock behavior that the clocking circuitry is actually capable of producing. When you use
a named capture procedure to define a waveform, it is assumed you have expert design
knowledge; the ATPG tool does not verify that the clocking circuitry is capable of delivering
the waveform to the defined internal pins.

The ATPG tool uses either all named capture procedures (the default) or only those named
capture procedures you enable with the set_capture_procedures command. When the test
procedure file does not contain named capture procedures, or you use the “set capture procedure
off -all” command, the tool uses the default capture procedure. However, usually you would not
use the default procedure to generate at-speed tests. The tool does not currently support use of
both named capture procedures and clock procedures in a single ATPG session.

Note
If a DRC error prevents use of a capture procedure, the run will abort.

For more information on named capture procedures, see the “Non-Scan Procedures” section in
the Tessent Shell User’s Manual.

Support for On-Chip Clocks (PLLs)

One way you can use named capture procedures is for the support of on-chip or internal clocks.
These are clocks generated on-chip by a phased-locked loop (PLL) or other clock generating
circuitry as shown in Figure 7-30. In addition, an example timing diagram for this circuit is
shown in Figure 7-31. A PLL can support only certain clock waveforms and named capture
procedures let you specify the allowed set of clock waveforms. In this case, if there are multiple
named capture procedures, the ATPG engine will use these named capture procedures instead
of assuming the default capture behavior.

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Figure 7-30. On-chip Clock Generation

External Signals Internal Signals

Integrated Circuit
clk1
system clk PLL
PLL clk2
begin_ac
Control
scan_en cntrl Design
Core

scan_clk1
scan_clk2

Defining Internal and External Modes

When manually creating or editing named capture procedures, you can use the optional
keyword “mode” with two mode blocks, “internal” and “external” to describe what happens on
the internal and external sides of an on-chip phase-locked loop (PLL) or other on-chip clock-
generating circuitry. You use “mode internal =” and “mode external =” to define mode blocks in
which you put procedures to exercise internal and external signals. You must use the internal
and external modes together, and ensure no cycles are defined outside the mode definitions.

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Figure 7-31. PLL-Generated Clock and Control Signals

240 ns total

system_clk

scan_en

begin_ac

scan_clk1

scan_clk2

clk1

clk2

Slow Fast Fast Slow

80 ns 40 ns 40 ns 80 ns

The internal mode is used to describe what happens on the internal side of the on-chip PLL
control logic, while the external mode is used to describe what happens on the external side of
the on-chip PLL. Figure 7-30 shows how this might look. The internal mode uses the internal
clocks (/pll/clk1 & /pll/clk2) and signals while the external mode uses the external clocks
(system_clk) and signals (begin_ac & scan_en). If any external clocks or signals go to both the
PLL and to other internal chip circuitry (scan_en), you need to specify their behavior in both
modes and they need to match, as shown in the following example (timing is from Figure 7-31):

timeplate tp_cap_clk_slow =
force_pi 0;
pulse /pll/clk1 20 20;
pulse /pll/clk2 40 20;
period 80;
end;
timeplate tp_cap_clk_fast =
force_pi 0;
pulse /pll/clk1 10 10;
pulse /pll/clk2 20 10;
period 40;
end;

timeplate tp_ext =
force_pi 0;
measure_po 10;
force begin_ac 60;
pulse system_clk 0 60;
period 120;

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end;

procedure capture clk1 =

observe_method master;

mode internal =
cycle slow =
timeplate tp_cap_clk_slow;
force system_clk 0;
force scan_clk1 0;
force scan_clk2 0;
force scan_en 0;
force_pi;
force /pll/clk1 0;
force /pll/clk2 0;
pulse /pll/clk1;
end;
// launch cycle
cycle =
timeplate tp_cap_clk_fast;
pulse /pll/clk2;
end;
// capture cycle
cycle =
timeplate tp_cap_clk_fast;
pulse /pll/clk1;
end;
cycle slow =
timeplate tp_cap_clk_slow;
pulse /pll/clk2;
end;
end;

mode external =
timeplate tp_ext;
cycle =
force system_clk 0;
force scan_clk1 0;
force scan_clk2 0;
force scan_en 0;
force_pi;
force begin_ac 1;
pulse system_clk;
end;
cycle =
force begin_ac 0;
pulse system_clk;
end;
end;
end;

For more information about internal and external modes, see the “Rules for Creating and
Editing Named Capture Procedures” section in the Tessent Shell User’s Manual.

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Displaying Named Capture Procedures

When the ATPG tool uses a named capture procedure, it uses a “cyclized” translation of the
internal mode. The tool may merge certain internal mode cycles in order to optimize them, and
it may expand others to ensure correct simulation results. These modifications are internal only;
the tool does not alter the named capture procedure in the test procedure file.
You can use the report_capture_procedures command to display the cyclized procedure
information with annotations that indicate the timing of the cycles and where the at-speed
sequences begin and end. If you want to view the procedures in their unaltered form in the test
procedure file, use the report_procedures command.

After cyclizing the internal mode information, the tool automatically adjusts the sequential
depth to match the number of cycles that resulted from the cyclizing process. Patterns will
automatically reflect any sequential depth adjustment the tool performs.

Figure 7-32 illustrates cycle merging.

Figure 7-32. Cycles Merged for ATPG

cycle A cycle B cycle A

clk

50 ns 50 ns 50 ns

Tool merges cycles A and B into 1 cycle internally to optimize.

cycle C cycle A
clk

100 ns 50 ns

Figure 7-33 illustrates cycle expansion.

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Figure 7-33. Cycles Expanded for ATPG

cycle A
clk1

clk2

0 ns 50 ns
Tool expands cycle A into 2 cycles internally for simulation.

cycle A cycle B
clk1

clk2

0 ns 50 ns

Achieving the Test Coverage Goal Using Named Capture

Procedures
Use the following steps when you know the repeated clock sequences to which the test clocks
will be applied and want to investigate the best sequential depth of each clock domain to
achieve the test coverage goal.
1. Use the create_capture_procedures command to define the minimum clock sequences in
a named capture procedure.
2. Gradually add more capture procedures with higher sequential depth until the test
coverage goal is achieved or the pattern count limit is reached.

Debugging Low Test Coverage Using Pre-Defined Named

Capture Procedures
Use the following steps when you want to know which clock sequences will provide the best
test coverage and then create a named capture procedure from it.
1. Start ATPG from the default capture procedure.
2. Use the create_capture_procedures command to create named capture procedures by
extracting the most often applied clock sequences from the pattern set.
3. Use the write_procfile command to write the named capture procedure into the test
procedure file.

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Note
You may need to manually edit the named capture procedure in the test procedure file to
achieve the functionality you want. For example, you may need to add condition
statements or add external mode definitions. For information on rules to follow when
editing named capture procedures, see the “Rules for Creating and Editing Named
Capture Procedures” section in the Tessent Shell User’s Manual.

Clocking During At-speed Fault Simulation

Not all clocks specified in the capture procedures are applied at-speed. During at-speed fault
simulation, the tool does not activate at-speed related faults when slow clock sequences are fault
simulated, even if a transition occurs in two consecutive cycles. Generally, the clock sequence
defined in a capture procedure can consist of zero or more slow clock sequences, followed by
zero or more at-speed clock sequences, followed by zero or more slow clock sequences.

Internal Signals and Clocks

The tool provides a way for you to specify internal signals and clocks.
For clocks and signals that come out of the PLL or clock generating circuitry that are not
available at the real I/O interface of the design, you use the -Internal switch with the add_clocks
or add_primary_inputs commands to define the internal signals and clocks for use in ATPG.

For example, when setting up for pattern generation for the example circuit shown in Figure 7-
30, you would issue this command to define the internal clocks:

SETUP> add_clocks 0 /pll/clk1 /pll/clk2

The two PLL clocks would then be available to the tool’s ATPG engine for pattern generation.

Saving Internal and External Patterns

By default, the ATPG tool uses only the primary input clocks when creating test patterns.
However, if you use named capture procedures with internal mode clocks and control signals
you define with the “add_clocks or “add_primary_inputs -Internal” command, the tool uses
those internal clocks and signals for pattern generation and simulation. To save the patterns
using the same internal clocks and signals, you must use the -Mode_internal switch with the
write_patterns command. The -Mode_internal switch is the default when saving patterns in
ASCII or binary format.

Note
The -Mode_internal switch is also necessary if you want patterns to include internal pin
events specified in scan procedures (test_setup, shift, load_unload).

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To obtain pattern sets that can run on a tester, you need to write patterns that contain only the
true primary inputs to the chip. These are the clocks and signals used in the external mode of
any named capture procedures, not the internal mode. To accomplish this, you must use the
-Mode_external switch with the write_patterns command. This switch directs the tool to map
the information contained in the internal mode blocks back to the external signals and clocks
that comprise the I/O of the chip. The -Mode_external switch is the default when saving
patterns in a tester format (for example, WGL) and Verilog format.

Note
The -Mode_external switch ignores internal pin events in scan procedures (test_setup,
shift, load_unload).

Mux-DFF Example
In a full scan design, the vast majority of transition faults are between scan cells (or cell to cell)
in the design. There are also some faults between the PI to cells and cells to the PO. Targeting
these latter faults can be more complicated, mostly because running these test patterns on the
tester can be challenging. For example, the tester performance or timing resolution at regular
I/O pins may not be as good as that for clock pins. This section shows a mux-DFF type scan
design example and covers some of the issues regarding creating transition patterns for the
faults in these three areas.
Figure 7-34 shows a conceptual model of an example chip design. There are two clocks in this
mux-DFF design, which increases the possible number of launch and capture combinations in
creating transition patterns. For example, depending on how the design is actually put together,
there might be faults that require these launch and capture combinations: C1-C1, C2-C2,
C1-C2, and C2-C1. The clocks may be either external or are created by some on-chip clock
generator circuitry or PLL.

“Timing for Transition Delay Tests” on page 225 shows the basic waveforms and partial test
procedure files for creating broadside and launch off shift transition patterns. For this example,
named capture procedures are used to specify the timing and sequence of events. The example
focuses on broadside patterns and shows only some of the possible named capture procedures
that might be used in this kind of design.

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Figure 7-34. Mux-DFF Example Design

scan chain
logic logic

PIs POs

logic

C1
C2

A timing diagram for cell to cell broadside transition faults that are launched by clock C1 and
captured by clock C2 is shown in Figure 7-35.

Figure 7-35. Mux-DFF Broadside Timing, Cell to Cell

120 ns 80 ns 40 ns 120 ns
scan_en

scan_clk

C2

C1
shift launch capture load/unload

Following is the capture procedure for a matching test procedure file that uses a named capture
procedure to accomplish the clocking sequence. Other clocking combinations would be handled
with additional named capture procedures that pulse the clocks in the correct sequences.

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set time scale 1.000000 ns ;

timeplate tp1 =
force_pi 0;
measure_po 10;
pulse scan_clk 50 20;
period 120;
end;
timeplate tp2 =
force_pi 0;
pulse c1 10 10;
pulse c2 10 10;
measure_po 30;
period 40;
end;
timeplate tp3 =
force_pi 0;
pulse c1 50 10;
pulse c2 10 10;
period 80;
end;
procedure load_unload =
timeplate tp1;
cycle =
force c1 0;
force c2 0;
force scan_en 1;
end;
apply shift 255;
end;
procedure shift =
timeplate tp1;
cycle =
force_sci;
measure_sco;
pulse scan_clk;
end;
end;
procedure capture launch_c1_cap_c2 =
cycle =
timeplate tp3;
force c1 0;
force c2 0;
force scan_clk 0;
force_pi;//force scan_en to 0
pulse c1;//launch clock
end;
cycle =
timeplate tp2;
pulse c2;//capture clock
end;
end;

Be aware that this is just one example and your implementation may vary depending on your
design and tester. For example, if your design can turn off scan_en quickly and have it settle
before the launch clock is pulsed, you may be able to shorten the launch cycle to use a shorter

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period; that is, the first cycle in the launch_c1_cap_c2 capture procedure could be switched
from using timeplate tp3 to using timeplate tp2.

Another way to make sure scan enable is turned off well before the launch clock is to add a
cycle to the load_unload procedure right after the “apply shift” line. This cycle would only need
to include the statement, “force scan_en 0;”.

Notice that the launch and capture clocks shown in Figure 7-35 pulse in adjacent cycles. The
tool can also use clocks that pulse in non-adjacent cycles, as shown in Figure 7-36, if the
intervening cycles are at-speed cycles.

Figure 7-36. Broadside Timing, Clock Pulses in Non-adjacent cycles

120 ns 40 ns 40 ns 40 ns 120 ns
scan_en
3 at-speed cycles

scan_clk
specified by
“launch_capture_pair c3 c3”
C3

C2
Also a valid capture point
shift launch capture load/unload

To define a pair of nonadjacent clocks for the tool to use as the launch clock and capture clock,
include a “launch_capture_pair” statement at the beginning of the named capture procedure.
Multiple “launch_capture_pair” statements are permitted, but the tool will use just one of the
statements for a given fault. Without this statement, the tool defaults to using adjacent clocks.

When its choice of a launch and capture clock is guided by a launch_capture_pair statement, the
tool may use for launch, the clock specified as the launch clock in the statement or another clock
that is pulsed between the launch and capture clocks specified in the statement. The capture
clock, however, will be the one specified in the statement or another clock that has the same
period as the specified capture clock.

If a named capture procedure for example pulses clocks clk1, clk2 and clk3 in that order in each
of three successive at-speed cycles and the launch_capture_pair {clk1, clk3} is defined, the tool
could use either clk1 or clk2 to launch and clk3 to capture. The idea of the launch and capture
pair is that it allows you to specify the capture clock and the farthest launch clock from the
capture clock. In this example, the {clk1, clk3} pair directs the tool to use clk3 to capture and
the farthest launch clock to be clk1. The tool considers it all right for clk2 to launch since if
{clk1, clk3} is at speed, {clk2, clk3} should be at speed as well.

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For more information on using the “launch_capture_pair” statement, see the

“launch_capture_pair Statement” section in the Tessent Shell User’s Manual.

If you want to try to create transition patterns for faults between the scan cells and the primary
outputs, make sure your tester can accurately measure the PO pins with adequate resolution. In
this scenario, the timing looks similar to that shown in Figure 7-35 except that there is no
capture clock. Figure 7-37 shows the timing diagram for these cell to PO patterns.

Figure 7-37. Mux-DFF Cell to PO Timing

120 ns 80 ns 40 ns 120 ns
scan_en

scan_clk

C1
shift launch meas. load/unload
PO

Following is the additional capture procedure that is required:

procedure capture launch_c1_meas_PO=

cycle =
timeplate tp3;
force_pi; //force scan_en to 0
pulse c1; //launch clock
end;
cycle =
timeplate tp2;
measure_po; //measure PO values
end;
end;

Note
You will need a separate named capture procedure for each clock in the design that can
cause a launch event.

What you specify in named capture procedures is what you get. As you can see in the two
preceding named capture procedures (launch_c1_cap_c2 and launch_c1_meas_PO), both
procedures used two cycles, with timeplate tp3 followed by timeplate tp2. The difference is that
in the first case (cell to cell), the second cycle only performed a pulse of C2 while in the second
case (cell to PO), the second cycle performed a measure_po. The key point to remember is that

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even though both cycles used the same timeplate, they only used a subset of what was specified
in the timeplate.

To create effective transition patterns for faults between the PI and scan cells, you also may
have restrictions due to tester performance and tolerance. One way to create these patterns can
be found in the example timing diagram in Figure 7-38. The corresponding named capture
procedure is shown after the figure.

Figure 7-38. Mux-DFF PI to Cell Timing

120 ns 40 ns 80 ns 120 ns
scan_en

scan_clk

C2

PI
shift setup launch and load/unload
initial capture
value

procedure capture launch_PI_cap_C2 =

cycle =
timeplate tp2;
force_pi; //force initial values
end;
cycle =
timeplate tp3;
force_pi; //force updated values
pulse c2; //capture clock
end;
end;

As before, you would need other named capture procedures for capturing with other clocks in
the design. This example shows the very basic PI to cell situation where you first set up the
initial PI values with a force, then in the next cycle force changed values on the PI and quickly
capture them into the scan cells with a capture clock.

Note
You do not need to perform at-speed testing for all possible faults in the design. You can
eliminate testing things like the boundary scan logic, the memory BIST, and the scan shift
path by using the add_nofaults command.

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Support for Internal Clock Control

You can use clock control definitions in your test procedure file to specify the operation of on-
chip/internal clocks during capture cycles. A clock control definition is one or more blocks in
the test procedure file that define internal clock operation by specifying source clocks and
conditions at scan cell outputs when a clock can be pulsed. ATPG interprets these definitions
and determines which clock to pulse during the capture cycle to detect the most faults.
When a clock control is defined, the clock control bits are included in chain test pattern to turn
off capture clocks when the clock under control has no source clock or when the source clock
defined in the clock under control is a always-pulse clock.

By default in the ATPG tool, one capture cycle is included in chain test patterns if there is no
pulse-always clock, or if the pulse-always clock drives neither observation points nor buses.

You can manually create clock control definitions or use the stil2mgc tool to generate them
automatically from a STIL Procedure File (SPF) that contains a ClockStructures block.

To specify explicit launch/capture sequences or external/internal clock relationship definitions,

you must use Named Capture Procedures (NCPs).

Note that you can turn off clock control using the “set_clock_controls off” command. You
should turn off clock controls only for debug purposes, typically to determine if a fault is
untestable because of the clock control constraints and if the fault is testable when those
constraints are not present. For more information, refer to the “set_clock_controls” command
description in the Tessent Shell Reference Manual.

Per-Cycle Clock Control

Per-cycle clock control allows you to define the internal clock output based on a single capture
cycle using scan cell values. Per-cycle clock control is commonly used for pipeline-based clock
generation where one bit controls the clock in each cycle.
Figure 7-39 shows a simplified per-cycle clock control model.

Figure 7-39. Simplified Per-Cycle Clock Control Model

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Sequence Clock Control

Sequence clock control allows you to define the internal clock output based on a sequence of
capture cycles using scan cell values. Sequence clock control is commonly used for counter-
based clock generation circuitry.
Figure 7-40 shows a simplified sequence clock control model.

Figure 7-40. Simplified Sequence Clock Control Model

Capture Cycle Determination

The actual capture cycle is relative to how many times the source clock pulses between scan
loads as follows.
If the source clock is pulse-always, the specified capture cycle and the actual capture cycle are
always the same because the source clock pulses in every ATPG cycle as shown in Figure 7-41.
No source clock or a pulse-in-capture clock are considered equivalent to a pulse-always source
clock.

If the source clock is defined as anything but pulse-always or pulse-in-capture, the

specified capture cycle is determined by the pulse of the source clock. For example, an internal
clock defined to pulse for cycle 0 may not pulse in cycle 0 but in the cycle that corresponds to
the source clock pulse as shown in Figure 7-41.

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Figure 7-41. Clock Control Capture Cycles

Applying Internal Clock Control

Use this procedure to apply internal clock control for ATPG via the test procedure file.

Restrictions and Limitations

Clock control definitions and Named Capture Procedures (NCPs) cannot both be enabled during
test pattern generation. By default, clock control definitions are disabled when NCPs are
enabled.

Note
Test patterns created with NCPs and test patterns created with the clock control
definitions can be fault simulated simultaneously.

Procedure
1. Depending on your application, either:
o Create clock control definitions in your test procedure file.
OR
o Run the stil2mgc tool on an SPF to generate a test procedure file with clock control
definitions. Refer to the stil2mgc description in the Tessent Shell Reference Manual.

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For more information, see “Clock Control Definition” in the Tessent Shell User’s
Manual.
2. Invoke Tessent Shell, set the context to “patterns -scan,” read in the netlist, and set up
the appropriate parameters for ATPG run.
3. Load the test procedure file created in step 1 and set up for ATPG. For example:
SETUP> add_scan_groups group1 scan_g1.procfile
SETUP> add_scan_chains chain1 group1 indata2 testout2
SETUP> add_scan_chains chain2 group1 indata4 testout4
SETUP> add_clocks 0 clk1

The add_scan_groups command loads the specified test procedure file from setup mode.
You can also use the read_procfile command load the test procedure file from analysis
mode. For more information, see “Test Pattern Formatting and Timing.”
4. Exit setup mode and run DRC. For example:
SETUP> set_system_mode analysis

5. Correct any DRC violations. For information on clock control definition DRCs, see
“Procedure Rules (P Rules)” in the Tessent Shell Reference Manual.
Clock control definitions are enabled by default unless there are NCPs enabled in the
test procedure file. If NCPs exist and are enabled, they override clock control
definitions.
6. Report the clock control configurations. For example:
ANALYSIS> report_clock_controls
CLOCK_CONTROL "/top/core/clk1 (3457)" =
SOURCE_CLOCK "/pll_clk (4)";
ATPG_CYCLE 0 =
CONDITION "/ctl_dff2/ (56)" 1;
END;
ATPG_CYCLE 1 =
CONDITION "/ctl_dff1/ (55)" 1;
END;
END;

Values in parenthesis are tool-assigned gate ID numbers.

7. Generate test patterns. For example:
ANALYSIS> create_patterns

8. Save test patterns. For example:

ANALYSIS> write_patterns ../generated/patterns_edt_p.stil -stil -replace

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Generating Test Patterns for Different Fault Models

and Fault Grading
Use this procedure when you want to create test patterns for path delay, transition, stuck-at, and
bridge fault models and fault grade the test patterns to improve fault coverage.
Note
You can use N-detect for stuck-at and transition patterns. If you use N-detect, replace the
stuck-at and/or transition patterns described in the procedure with the N-detect patterns.

Prerequisites
• A test procedure file must be available. See “Test Procedure File” in the Tessent Shell
User’s Manual.
• A path definition file must be available. See “The Path Definition File” on page 240.
• A bridge definition file must be available. See “Net Pair Identification with Calibre for
Bridge Fault Test Patterns” on page 214.

Procedure
1. Create path delay test patterns for your critical path(s) and save them to a file. Then fault
grade the path delay test patterns for transition fault coverage. Following are example
commands in a dofile.
//-----------------Create path delay patterns--------------------
// Enable two functional pulses (launch and capture).
set_fault_type path_delay
read_fault_sites my_critical_paths
report_fault_sites path0
// Uncomment next 2 lines to display path in DFTVisualizer.
// set_gate_level primitive
// report fault sites path0 -display debug
create_patterns
// Save path delay patterns.
write_patterns pathdelay_pat.bin.gz -binary -replace
//---------------------------------------------------------------
//----------Grade for broadside transition fault coverage--------
// Change the fault model (when you change the fault model, the
// the internal pattern set database is emptied).
set_fault_type transition -no_shift_launch
add_faults -all
// Read the previously saved path delay patterns into the external
// pattern set database; include the -All_patterns switch so the
// patterns are copied to the now empty internal pattern set
// database when they are simulated.
read_patterns pathdelay_pat.bin.gz
// Simulate all the path delay patterns for transition fault
// coverage, copying them into the internal pattern set as they
// are simulated.
simulate_patterns -store_patterns all

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report_statistics
//---------------------------------------------------------------

2. Create additional transition test patterns for any remaining transition faults, and add
these test patterns to the original test pattern set. Then fault grade the enlarged test
pattern set for stuck-at fault coverage. Following are example commands in a dofile.
//---------Create add’l transition fault patterns-------
// Create transition patterns that detect the remaining transition
// faults the path delay patterns did not detect during
// simulation.
create_patterns
order_patterns 3 // optimize the pattern set
// Save original path delay patterns and add’l transition patterns.
write_patterns pathdelay_trans_pat.bin.gz -binary -replace
//---------------------------------------------------------------
//----------Grade for stuck-at fault coverage--------------------
set_fault_type stuck
add_faults -all
// Read in previously saved path delay and transition patterns and
// add them to the internal pattern set when they are simulated.
read_patterns pathdelay_trans_pat.bin.gz -all_patterns
simulate_patterns -store_patterns all
report_statistics
//---------------------------------------------------------------

3. Create additional stuck-at test patterns for any remaining stuck-at faults and add them to
the test pattern set. Then fault grade the enlarged test pattern set for bridge fault
coverage. Following are example commands in a dofile.
//----------Create add’l (top-up) stuck-at patterns--------------
create_patterns
order_patterns 3 // optimize the pattern set
// Save original path delay patterns and transition patterns, plus
// the add’l stuck-at patterns.
write_patterns pathdelay_trans_stuck_pat.bin.gz -binary -replace
//---------------------------------------------------------------
//----------Grade for bridge fault coverage--------------------
set_fault_type bridge
read_fault_sites my_fault_definitions
add_faults -all
// Read in previously saved path delay, transition, and stuck-at
// patterns and add them to the internal pattern set
// when they are simulated.
read_patterns pathdelay_trans_stuck_pat.bin.gz
simulate_patterns -store_patterns all
report_statistics
//---------------------------------------------------------------

4. Create additional bridge test patterns for any remaining bridge faults and add these test
patterns to the test pattern set. Following are example commands in a dofile.
//----------Create add’l bridge patterns--------------
create_patterns
order_patterns 3 // optimize the pattern set
// Save original path delay patterns, transition patterns, stuck-at

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// patterns, plus the add’l bridge patterns.

write_patterns pathdelay_trans_stuck_bridge_pat.bin.gz -binary
-replace
// Close the session and exit.
exit

Using Timing-Aware ATPG

Timing-aware ATPG reads timing information from a Standard Delay Format (SDF) file and
tries to generate patterns that detect transition faults using the longest detection path.

Slack Calculation
Slack is equal to the margin between the path delay and the clock period. Slack within Small
Delay Fault Model represents the smallest delay defect that can be detected.
Figure 7-42 illustrates slack calculations. Assume there are three paths that can detect a fault.
The paths have a 9.5 ns, 7.5 ns and 7.0 ns delay, respectively. The clock period is 10 ns. The
slacks for the paths are calculated as 0.5 ns, 2.5 ns and 3 ns, respectively. For the longest path,
which has a 0.5 ns slack, the smallest delay defect can be detected is 0.5 ns. Similarly for the
path with 2.5 ns, the smallest detectable delay defect is 2.5 ns. You can detect smaller delay
defects by using a path with a smaller slack. Any path that is longer than the clock period is a
false path.

Figure 7-42. Timing Slack

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Delay Test Coverage Calculation

ATPG calculates a metric called Delay Test Coverage to determine the quality of the test
patterns. The delay test coverage is automatically included in the ATPG statistics report when
timing-aware ATPG is enabled.
Delay Test Coverage = (Path Delay that ATPG used)/(Longest Path Delay)*100%

Using the same example paths from Figure 7-42, if ATPG uses path R2, the delay test coverage
is: 7.5 ns / 9.5 ns = 79%. If ATPG used the longest path (R1), the delay test coverage would be
100%.

Undetected faults have a delay test coverage of 0%. DI faults (Detected by Implication) have a
delay test coverage of 100%. Chip-level delay test coverage is calculated by averaging the delay
test coverage values for all faults.

Timing-Aware ATPG vs. Transition ATPG

The following data was gathered from the STARC03 testcase, which STARC and Mentor
Graphics used to evaluate timing-aware ATPG.
Table 7-3 compares transition fault ATPG and timing-aware ATPG. The testcase has the
following characteristics:

• Design Size: 2.4M sim_gate

• Number of FFs: 69,153
• CPU: 2.2Ghz
• 315 sec to read SDF file that has 10,674,239 lines

Table 7-3. Testcase 2 Data

Run Parameters Pattern Transition Delay CPU Time Memory
Count TC TC
Before ATPG 0 0.00% 0.00% 0 2.1G
Transition ATPG 3,668 91.23% 66.13% 4,180 sec 1.0G
(1 detection)
Transition ATPG 7,979 91.27% 68.07% 11,987 sec 1.0G
(7 detections)
Timing-Aware ATPG 3,508 91.19% 67.39% 17,373 sec 2.2G
(SMFD1 = 100%)
Timing-Aware ATPG 8,642 91.26% 76.26% 129,735 sec 2.2G
(SMFD = 50%)
Timing-Aware ATPG 24,493 91.28% 77.71% 178,673 sec 2.2G
(SMFD = 0%)

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1. Slack Margin for Fault Dropping = (Ta-Tms)*100/Ta.

For more information about setting run parameters, see the set_atpg_timing command.

Timing-Aware ATPG Limitations

The following is a list of timing-aware ATPG limitations:
• For limitations regarding the SDF file, see the read_sdf command.
• Launch-off shift is not supported.
• The ATPG run time for timing-aware is about eight times slower than the normal
transition fault ATPG. For more information, see “Timing-Aware ATPG vs. Transition
ATPG”. Targeting critical faults may help.
• A large combinational loop may slow down the analysis to calculate the static slack. It
also makes the actual delay analysis less accurate.
• The transition test coverage on timing-aware ATPG may be lower than the normal
transition fault ATPG. Because timing-aware ATPG tries to detect a fault with a longer
path, it is more likely to hit the abort limit. Use the -Retarget switch with the
create_patterns command to improve transition test coverage.
• False and multicycle paths are defined by the add_false_paths or read_sdc commands.
The slack for false and multicycle paths is rounded to zero, and the delay test coverage is
0%.
• Timing information in Named Capture Procedure are not included in static slack
calculations.
• When saving check point, the SDF database is not stored. You must reload the SDF
data, using the read_sdf command, when using a flattened model.
• Static Compression (compress_patterns command) and pattern ordering (order_patterns
command) cannot be used when -Slack_margin_for_fault_dropping is specified.
• Clock skew is ignored. Clock skew is caused by a delay on clock paths, and timing-
aware ATPG does not use the delay on clock path. Consequently, clock skew (the delay
on clock path) is not accounted for in the timing-calculation.
• The SDF file does not impact the good machine simulation value. The SDF file is used
mainly to guide timing-aware ATPG to detect a fault along a long path as well as for
calculating the delay test coverage. That is, the tool does not extract (or infer) false and
multicycle paths from the delays in the SDF file. You have to provide that information
with an SDC file or add_false_paths command.

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Understanding Inaccuracies in Timing Calculations

When performing timing-aware ATPG based on SDF, the following factors can result in
inaccurate timing calculations:
• Device delay is not supported.
• Conditional delay is not supported. For a given IOPATH or INTERCONNECT pin pair,
the maximal number among all conditional delays defined for this pair will be used by
the tool when calculating static and actual delays.
• Negative delay is supported. However, if path delay is a negative number, the tool forces
the delay value to 0 when calculating delay coverage, path delay, and slack, etc.
• Static timing is calculated in a pessimistic way that does not consider false or multicycle
paths.

Running Timing-Aware ATPG

Use this procedure to create Timing-Aware test patterns with the ATPG tool.

Prerequisites
• Because timing-aware ATPG is built on transition ATPG technology, you must set up
for Transition ATPG first before starting this procedure. See “Creating a Transition
Delay Test Set.”
• SDF file from static timing analysis.

Procedure
1. Load the timing information from an SDF file using the read_sdf command. For
example:
ANALYSIS> read_sdf top_worst.sdf

If you encounter problems reading the SDF file, see “Errors and Warnings While
Reading SDF Files” on page 272.
2. Define clock information using the set_atpg_timing command. You must define the
clock information for all clocks in the design, even for those not used for ATPG (not
used in a named capture procedure). For example:
ANALYSIS> set_atpg_timing -clock clk_in 36000 18000 18000
ANALYSIS> set_atpg_timing -clock default 36000 18000 18000

3. Enable timing-aware ATPG using the set_atpg_timing command. For example:

ANALYSIS> set_atpg_timing on -slack_margin_for_fault_dropping 50%

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If you specify a slack margin for fault dropping, the fault simulation keeps faults for
pattern generation until the threshold is met. During normal transition fault simulation,
faults are dropped as soon as they are detected.
4. Select timing-critical faults using the set_atpg_timing command. For example:
ANALYSIS> set_atpg_timing -timing_critical 90%

5. Run ATPG. For example:

ANALYSIS> create_patterns -retarget

6. Report delay_fault test coverage using the report_statistics command. For example:
ANALYSIS> report_statistics

Example
Figure 7-43 shows a testcase where there are 17 scan flip-flops and 10 combinational gates.
Each gate has 1 ns delay and there is no delay on the scan flip-flops. A slow-to-rise fault is
injected in G4/Y. The test period is 12 ns. The last scan flip-flop (U17) has an OX cell
constraint so that it cannot be used as observation point.

The longest path starts at U1, moving through G1 through G10 and ending at U17. The total
path delay is 10 ns. Because U17 cannot be used as observation point, timing-aware ATPG uses
the path starting at U1, moving through G1 through G9 and ending at U16. The total path delay
is 9 ns.

Therefore static minimum slack is 12 ns - 10 ns = 2 ns, and the best actual slack is 12 ns - 9 ns =
3 ns.

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Figure 7-43. Testcase 1 Logic

Figure 7-44 shows the dofile used for this example. As you can see in the comments, the dofile
goes through the procedure outlined in “Running Timing-Aware ATPG.”

Figure 7-44. Dofile

Figure 7-45 shows the fault statistics report. Timing-aware ATPG used the longest possible
path, which is 9 ns. Static longest path is 10 ns. The delay test coverage is 9 ns / 10 ns = 90%.

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Figure 7-45. Reports

Statistics report
---------------------------------------------
#faults #faults
fault class (coll.) (total)
------------------------ ------- -------
FU (full) 1 1
------------------------ ------- -------
DS (det_simulation) 1 1
------------------------ ------- -------
test_coverage 100.00% 100.00%
fault_coverage 100.00% 100.00%
atpg_effectiveness 100.00% 100.00%
------------------------ ------- -------
Delay_test_coverage 90% 90%
---------------------------------------------
#test_patterns 1
#clock_seqiential_patterns 1
#simulated_patters 32
CPU_time (secs) 0.1
---------------------------------------------

Troubleshooting Topics
The following topics describe common issues related to timing-aware ATPG:
• Reducing run Time for Timing-Aware ATPG
• Errors and Warnings While Reading SDF Files
• Warnings During ATPG
• Actual Slack Smaller Than Tms

Reducing run Time for Timing-Aware ATPG

Timing-aware ATPG takes much longer than the regular transition ATPG. You can make it
faster by targeting for timing-critical faults only.
For more information about why timing-aware ATPG is slower, refer to Table 7-3 on page 266.

You can use the set_atpg_timing command to make a fault list that includes faults with less
slack time than you specified. The fault is put in the fault list if its (Longest delay)/(Test time) is
more than your specified threshold.

For example, assume there are 3 faults and their longest paths are 9.5 ns, 7.5 ns and 6.0 ns
respectively and the test time is 10 ns. The (Longest delay)/(Test time) is calculated 95%, 75%
and 60% respectively. If you set the threshold to 80%, only the first fault is included. If you set
it to 70%, the first and second faults are included.

The following series of commands inject only timing-critical faults with 70% or more.

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ANALYSIS> add_faults -all

ANALYSIS> set_atpg_timing -timing_critical 70%
ANALYSIS> write_faults fault_list1 -timing_critical -replace
ANALYSIS> delete_faults -all
ANALYSIS> read_faults fault_list1

Errors and Warnings While Reading SDF Files

Below is a list of the most common errors and warnings when reading SDF.
• Error: Near line N - The destination pin A/B/C is mapped to gate output

For an interconnect delay, the destination pin has to be a gate input pin. If a gate output
pin is used, the tool issues the error message and ignores the delay.
• Error: Near line N - The pin A/B/C for conditional delay is mapped to
gate output.

The signals used for a condition on a conditional delay must all be gate inputs. If a gate
output is used for the condition, the tool issues the error message and ignores the delay.
For example, there is a component that has two outputs “O1” and “O2”. If a conditional
delay on “O1” is defined by using “O2”, it will produce an error.
• Error: Near line N - Unable to map destination SDF pin.
Ignore INTERCONNECT delay from pin A/B/C to pin D/E/F.

The destination pin “D/E/F” does not have a receiver. It is likely to be a floating net,
where no delay can be added.
• Error: Near line N - Unable to map source SDF pin.
Ignore INTERCONNECT delay from pin A/B/C to pin D/E/F.

The source pin “A/B/C” does not have a driver. It is likely to be a floating net, where no
delay can be added.
• Error: Near line N - Unable to map flatten hierarchical source pin
A/B/C derived from D/E/F.

“D/E/F” is a source pin defined in SDF (either interconnect delay or IO Path delay), and
“A/B/C” is a flattened (cell library based) pin corresponding to “D/E/F,” and there is no
driver found.
• Error: Near line N - Unable to map flatten hierarchical destination pin
A/B/C derived from D/E/F.

“D/E/F” is a destination pin defined in SDF (either internet delay or IO Path delay), and
“A/B/C” is a flattened (cell library based) pin corresponding to “D/E/F,” and there is no
receiver found.

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• Error: Near line N - There is no net path from pin A/B/C to pin D/E/F
(fanin=1). Ignore current net delay.

There is no connection between “A/B/C” (source) and “D/E/F” (destination). The

“fanin=1” means the first input of the gate.
• Warning: Near line N - Negative delay value is declared.

Negative delay is stored in the timing database and the negative value can be seen as the
minimum delay, but the negative number is not used for delay calculation. It will be
treated as a zero.
• Error: Near line N - Flatten hierarchical destination pin "A" is not
included in instance "B".

In this case, “B” is an instance name where both the source and destination pins are
located (for example, an SDF file was loaded with “-instance B” switch). But a flattened
hierarchical destination pin (cell based library) “A” traced from the destination pin is
outside of the instance “B.”
• Error: Near line N - Flatten hierarchical source pin "A" is not
included in instance "B".

In this case, “B” is an instance name where both the source and destination pins are
located (for example, an SDF file was loaded with “-instance B” switch). But a flattened
hierarchical source pin (cell based library) “A” traced from the source pin is outside of
the instance “B.”
• Error: Near line N - Unable to add IOPATH delay from pin A/B/C to pin
D/E/F even when the pins are treated as hierarchical pin. Ignore IOPATH
delay.

There was an error when mapping the hierarchical source “A/B/C” and/or destination
“D/E/F” to their flattened hierarchical pins. The delay is ignored. Most likely, one or
both hierarchical pins are floating.
• Error: Near line N - Unable to add INTERCONNECT delay from pin "A/B/C"
to pin "D/E/F" even when the pins are treated as hierarchical pin.
Ignore INTERCONNECT delay.

There was an error when mapping the hierarchical source “A/B/C” and/or destination
“D/E/F” to their flattened hierarchical pins. The delay is ignored. Most likely, one or
both hierarchical pins are floating.
• Error: Near line N - The conditional delay expression is not supported.
Ignore the delay.

The delay was ignored because the condition was too complex.
• Error: Near line N - Delay from pin A/B/C (fanin=1) to pin D/E/F has
been defined before. Ignore current IOPATH delay.

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Two or more IOPATH delays defined from SDF pin names map to the same flattened
gate pin. The tool will keep the value for the first definition and ignore all the
subsequent duplicates. This error with IOPATH delay is likely an SDF problem. For
interconnect delay, the tool cannot handle multiple definitions. For example, the Verilog
path is: /u1/Z -> net1 -> net2 -> /u2/A. If you define the first interconnect delay /u1/Z ->
net1, the second net1 -> net2, and the third net2 -> /u2/A, then all the three interconnect
delays are mapped to /u1/Z -> /u2/A in the gate level. This will cause an error.
• Error: Near line N - There is no combinational path from pin A/B/C
(fanin=3) to pin D/E/F. Ignore current IOPATH delay.

The tool can only handle the IOPATH delay through combinational gates. Delays
passing through state elements cannot be used.

Warnings During ATPG

You may see the following message when you start ATPG:
Warning: Inconsistent holding PI attribute is set between test generation
and static timing analysis.
Test generation with capture procedures holds PI, but static timing
analysis allows PI change.
The inconsistent settings will impact the timing metrics reported by the
tool.

This is due to the conflicting setting for holding PI and masking PO between ATPG and static
timing analysis.

For ATPG, holding PI and masking PO can be set by:

• Including the “add_input_constraints -hold” and “set_output_masks” commands in your

dofile.
• Including the “force_pi” and “measure_po” statements in a named capture procedure.
For static timing analysis, holding PI and masking PO can be set using the “-hold_pi” and
“-mask_po” switches for the set_atpg_timing command.

The static timing database is created once, before running ATPG, with the information of
holding PI and/or masking PO, whereas the setting can change for ATPG especially when using
NCP (for example, one NCP has force_pi and the other does not).

Actual Slack Smaller Than Tms

This condition occurs when a one-shot circuit is used in the data path. In static analysis, it is
identified as a “static-inactive” (STAT) path but it could be used as a fault detection path.
For more information, refer to the circuit in Figure 7-46.

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Figure 7-46. Glitch Detection Case

A slow-to-rise fault is injected at the input of the inverter (U1). There are two paths to detect it.
One is FF2 through U1 and U2 and the other is FF3 through U1 and U3. In the second path, the
fault is detected as a glitch detection as shown in the waveform.

But in the static analysis (when calculating Tms), this path is blocked because U3/out is
identified as a Tie-0 gate by DRC. Therefore the maximum static delay will be 2 ns (first path).
If the test cycle is 10 ns, the Tms will be 8 ns. And if ATPG uses the second path to detect the
fault, the actual slack will be 7 ns (10-1-2).

You will see the following message during DRC if DRC identifies a Tie-AND or Tie-OR.

// Learned tied gates: #TIED_ANDs=1 #TIED_ORs=1

Following are reports for the example shown in Figure 7-46.

// command: report_faults -delay

// type code pin_pathname
// static_slack actual_min_slack
// ---- ---- ------------ ------------ ----------------
0 DS /U1/A
// 8.0000 7.0000
// command: set_gate_report delay actual_path 0
// command: report_gates /U1/A
// /U1 inv01
// A I (0-1) (3.0000) /sff1/Q
// Y O (1-0) (3.0000) /U2/A /U3/A1
// command: report_gates /U3/A1

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// /U3 and02
// A0 I (0-1) (3.0000) /sff1/Q
// A1 I (1-0) (3.0000) /U1/Y
// Y O (0-0) (-) /sff3/D

// command: set_gate_report delay static_path

// command: report_gates /U3
// /U3 and02
// A0 I (2.0000, 2.0000)/(0.0000, 0.0000) /sff1/Q
// A1 I (2.0000, 2.0000)/(2.0000, 2.0000) /U1/Y
// Y O (STAT) /sff3/D

Generating Patterns for a Boundary Scan

Circuit
The following example shows how to create a test set for an 1EEE 1149.1 (boundary scan)-
based circuit. The following subsections list and explain the dofile and test procedure file.

Dofile and Explanation

The following dofile shows the commands you could use to specify the scan data in the ATPG
tool:
add_clocks 0 tck
add_scan_groups group1 proc_fscan
add_scan_chains chain1 group1 tdi tdo
add_input_constraints tms -c0
add_input_constraints trstz -c1
set_capture_clock TCK -atpg

You must define the tck signal as a clock because it captures data. There is one scan group,
group1, which uses the proc_fscan test procedure file (see page 278). There is one scan chain,
chain1, that belongs to the scan group. The input and output of the scan chain are tdi and tdo,
respectively.

The listed pin constraints only constrain the signals to the specified values during ATPG—not
during the test procedures. Thus, the tool constrains tms to a 0 during ATPG (for proper pattern
generation), but not within the test procedures, where the signal transitions the TAP controller
state machine for testing. The basic scan testing process is:

1. Initialize scan chain.

2. Apply PI values.
3. Measure PO values.
4. Pulse capture clock.
5. Unload scan chain.

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During Step 2, you must constrain tms to 0 so that the Tap controller’s finite state machine
(Figure 7-47) can go to the Shift-DR state when you pulse the capture clock (tck). You
constrain the trstz signal to its off-state for the same reason. If you do not do this, the Tap
controller goes to the Test-Logic-reset_state at the end of the Capture-DR sequence.

The set_capture_clock TCK -ATPG command defines tck as the capture clock and that the
capture clock must be used for each pattern (as the ATPG tool is able to create patterns where
the capture clock never gets pulsed). This ensures that the Capture-DR state properly transitions
to the Shift-DR state.

TAP Controller State Machine

The following shows the finite state machine for the TAP controller of a IEEE 1149.1 circuit.

Figure 7-47. State Diagram of TAP Controller Circuitry

Test-Logic
1 -Reset
0 Data Register Instruction
(Scan & Boundary Scan) Register
run-Test/ Select- Select-
0 Idle 1 DR-Scan 1 IR-Scan 1
0 0

Capture-DR Capture-IR
1 1
0 0

Shift-DR 0 Shift-IR 0
1 1

Exit1-DR Exit1-IR
1 1
0 0

Pause-DR 0 Pause-IR 0
1 1
0 0
Exit2-DR Exit2-IR
1 1

Update-DR Update-IR
1 0 1 0

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The TMS signal controls the state transitions. The rising edge of the TCK clock captures the
TAP controller inputs. You may find this diagram useful when writing your own test procedure
file or trying to understand the example test procedure file that the next subsection shows.

Test Procedure File and Explanation

The test procedure file proc_fscan follows:
set time scale 1 ns;
set strobe_window time 1;
timeplate tp0 =
force_pi 100;
measure_po 200;
pulse TCK 300 100;
period 500;
end;

procedure test_setup =
timeplate tp0;

// Apply reset procedure

// Test cycle one

cycle =
force TMS 1;
force TDI 0;
force TRST 0;
pulse TCK;
end;

// "TMS"=0 change to run-test-idle

// Test cycle two

cycle =
force TMS 0;
force TRST 1;
pulse TCK;
end;

// "TMS"=1 change to select-DR

// Test cycle three

cycle =
force TMS 1;
pulse TCK;
end;

// "TMS"=1 change to select-IR

// Test cycle four

cycle =
force TMS 1;
pulse TCK;
end;

// "TMS"=0 change to capture-IR

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// Test cycle five

cycle =
force TMS 0;
pulse TCK;
end;

// "TMS"=0 change to shift-IR

// Test cycle six

cycle =
force TMS 0;
pulse TCK;
end;

// load MULT_SCAN instruction "1000" in IR

// Test cycle seven

cycle =
force TMS 0;
pulse TCK;
end;

// Test cycle eight

cycle =
force TMS 0;
pulse TCK;
end;

// Test cycle nine

cycle =
force TMS 0;
pulse TCK;
end;

// Last shift in exit-IR Stage

// Test cycle ten

cycle =
force TMS 1;
force TDI 1;
pulse TCK;
end;

// Change to shift-dr stage for shifting in data

// "TMS" = 11100
// "TMS"=1 change to update-IR state
// Test cycle eleven

cycle =
force TMS 1;
force TDI 1;
pulse TCK;
end;

// "TMS"=1 change to select-DR state

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// Test cycle twelve

cycle =
force TMS 1;
pulse TCK;
end;

// "TMS"=0 change to capture-DR state

// Test cycle thirteen

cycle =
force TMS 0;
pulse TCK;
end;

// "TMS"=0 change to shift-DR state

// Test cycle fourteen

cycle =
force TMS 0;
force TEST_MODE 1;
force RESETN 1;
pulse TCK;
end;
end;

procedure shift =
scan_group grp1;
timeplate tp0;
cycle =
force_sci;
measure_sco;
pulse TCK;
end;
end;

procedure load_unload =
scan_group grp1;
timeplate tp0;
cycle =
force TMS 0;
force CLK 0;
end;
apply shift 77;

// "TMS"=1 change to exit-1-DR state

cycle =
force TMS 1;
end;
apply shift 1;

// "TMS"=1 change to update-DR state

cycle =
force TMS 1;
pulse TCK;
end;

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// "TMS"=1 change to select-DR-scan state

cycle =
force TMS 1;
pulse TCK;
end;

// "TMS"=0 change to capture-DR state

cycle =
force TMS 0;
pulse TCK;
end;
end;

Upon completion of the test_setup procedure, the tap controller is in the shift-DR state in
preparation for loading the scan chain(s). It is then placed back into the shift-DR state for the
next scan cycle. This is achieved by the following:

• The items that result in the correct behavior are the pin constraint on tms of C1 and the
fact that the capture clock has been specified as TCK.
• At the end of the load_unload procedure, the tool asserts the pin constraint on TMS,
which forces tms to 0.
• The capture clock (TCK) occurs for the cycle and this results in the tap controller
cycling from the run-test-idle to the Select-DR-Scan state.
• The load_unload procedure is again applied. This will start the next load/unloading the
scan chain.
The first procedure in the test procedure file is test_setup. This procedure begins by resetting
the test circuitry by forcing trstz to 0. The next set of actions moves the state machine to the
Shift-IR state to load the instruction register with the internal scan instruction code (1000) for
the MULT_SCAN instruction. This is accomplished by shifting in 3 bits of data (tdi=0 for three
cycles) with tms=0, and the 4th bit (tdi=1 for one cycle) when tms=1 (at the transition to the
Exit1-IR state). The next move is to sequence the TAP to the Shift-DR state to prepare for
internal scan testing.

The second procedure in the test procedure file is shift. This procedure forces the scan inputs,
measures the scan outputs, and pulses the clock. Because the output data transitions on the
falling edge of tck, the measure_sco command at time 0 occurs as tck is falling. The result is a
rules violation unless you increase the period of the shift procedure so tck has adequate time to
transition to 0 before repeating the shift. The load_unload procedure, which is next in the file,
calls the shift procedure.

The basic flow of the load_unload procedure is to:

1. Force circuit stability (all clocks off, etc.).

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2. Apply the shift procedure n-1 times with tms=0

3. Apply the shift procedure one more time with tms=1
4. Set the TAP controller to the Capture-DR state.
The load_unload procedure inactivates the reset mechanisms, because you cannot assume they
hold their values from the test_setup procedure. It then applies the shift procedure 77 times
with tms=0 and once more with tms=1 (one shift for each of the 77 scan registers within the
design). The procedure then sequences through the states to return to the Capture-DR state. You
must also set tck to 0 to meet the requirement that all clocks be off at the end of the procedure.

Using the MacroTest Capability

MacroTest is a utility that helps automate the testing of embedded logic and memories (macros)
by automatically translating user-defined patterns for the macros into scan patterns. Because it
enables you to apply your macro test vectors in the embedded environment, MacroTest
improves overall IC test quality. It is particularly useful for testing small RAMs and embedded
memories but can also be used for a disjoint set of internal sites or a single block of hardware
represented by an instance in HDL.
This is illustrated conceptually in Figure 7-48.

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Figure 7-48. Conceptual View of MacroTest

Scan Patterns Scan Patterns

MacroTest
0 1

1 1 0 0
0 0
1 0 1
1 1
0
1 Macro 1
0 Logic 0 0 Logic 1
1 1
0
1 1

0 0

1 0
010001
01110101
01000010
Macro 10011100
10111000
Test Vectors
11110011
(user defined)

MacroTest provides the following capabilities and features:

• Supports user-selected scan observation points

• Supports synchronous memories; for example, supports positive (or negative) edge-
triggered memories embedded between positive (or negative) edge-triggered scan chains
• Enables you to test multiple macros in parallel
• Analyzes single macros, and reports patterns that are logically inconsistent with the
surrounding hardware
• Allows you to define macro output values that do not require observation
• Has no impact on area or performance

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The MacroTest Process Flow

To use MacroTest effectively, you need to be familiar with two commands:
• set_macrotest_options — Modifies two rules of the DRCs to allow otherwise illegal
circuits to be processed by MacroTest. Black box (un-modelled) macros may require
this command.
• macrotest — Runs the MacroTest utility to read functional patterns you provide and
convert them into scan-based manufacturing test patterns.
The MacroTest flow requires a set of patterns and MacroTest. The patterns are a sequence of
tests (inputs and expected outputs) that you develop to test the macro. For a memory, this is a
sequence of writes and reads. You may need to take embedding restrictions into account as you
develop your patterns. Next, you set up and run MacroTest to convert these cycle-based patterns
into scan-based test patterns. The converted patterns, when applied to the chip, reproduce your
input sequence at the macro’s inputs through the intervening logic. The converted patterns also
ensure that the macro’s output sequence is as you specified in your set of patterns.

Note
You can generate a wide range of pattern sets: From simple patterns that verify basic
functionality, to complex, modified March algorithms that exercise every address
location multiple times. Some embeddings (the logic surrounding the macro) do not
allow arbitrary sequences, however.

Figure 7-49 shows the basic flow for creating scan-based test patterns with MacroTest.

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Figure 7-49. Basic Scan Pattern Creation Flow with MacroTest

Simulate macro
stand-alone

Capture port For memories, use

patterns to a file Perl or Awk to
create pattern file

Use the ATPG tool to

read the top-level
design

Setup Mode
set_macrotest_options
(if necessary)

Analysis Mode
Macrotest

write_patterns

Note
The patterns produced by MacroTest cannot be read back into the ATPG tool. This is
because the simulation and assumptions about the original macro patterns are no longer
valid and the original macro patterns are not preserved in the MacroTest patterns.

Tessent Diagnosis, a Mentor Graphics test failure diagnosis tool, also cannot read a
pattern set that includes MacroTest patterns. If Tessent Diagnosis is used in your
manufacturing test flow, save MacroTest patterns separately from your other patterns.
This will enable a test engineer to remove them from the set of patterns applied on ATE
before attempting to read that set into Tessent Diagnosis for diagnosis.

When you run the macrotest command, MacroTest reads your pattern file and begins analyzing
the patterns. For each pattern, the tool searches back from each of the macro’s inputs to find a
scan flip-flop or primary input. Likewise, the tool analyzes observation points for the macro’s
output ports. When it has justified and recorded all macro input values and output values,
MacroTest moves on to the next pattern and repeats the process until it has converted all the
patterns. The default MacroTest effort exhaustively tries to convert all patterns. If successful,

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then the set of scan test patterns MacroTest creates will detect any defect inside the macro that
changes any macro output from the expected value.

Note
If you add_faults prior to running MacroTest, the ATPG tool automatically fault
simulates the patterns as they are created. This is time consuming, but is retained for
backward compatibility. It is advised that you generate and save macrotest patterns in a
separate run from normal ATPG and faultsim and that you not issue the add_faults
command in the MacroTest run.

The patterns you supply to MacroTest must be consistent with the macro surroundings
(embedding) to assure success. In addition, the macro must meet certain design requirements.
The following sections detail these requirements, describe how and when to use MacroTest, and
conclude with some examples.

Qualifying Macros for MacroTest

If a design meets the following three criteria, then you can use MacroTest to convert a sequence
of functional cycles (that describe I/O behavior at the macro boundary) into a sequence of scan
patterns:
1. The design has at least one combinational observation path for each macro output pin
that requires observation (usually all outputs).
2. All I/O of the RAM/macro block to be controlled or observed are unidirectional.
3. The macro/block can hold its state while the scan chain shifts, if the test patterns require
that the state be held across patterns. This is the case for a March algorithm, for
example.
If you write data to a RAM macro (RAM), for example, then later read the data from the RAM,
typically you will need to use one scan pattern to do the write, and a different scan pattern to do
the read. Each scan pattern has a load/unload that shifts the scan chain, and you must ensure that
the DFT was inserted, if necessary, to allow the scan chain to be shifted without writing into the
RAM. If the shift clock can also cause the RAM to write and there is no way to protect the
RAM, then it is very likely that the RAM contents will be destroyed during shift; the data
written in the early pattern will not be preserved for reading during the latter pattern. Only if it is
truly possible to do a write followed by a read, all in one scan pattern, then you may be able to
use MacroTest even with an unprotected RAM.

Because converting such a multicycle pattern is a sequential ATPG search problem, success is
not guaranteed even if success is possible. Therefore, you should try to convert a few patterns
before you depend on MacroTest to be able to successfully convert a given embedded macro.
This is a good idea even for combinational conversions.

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If you intend to convert a sequence of functional cycles to a sequence of scan patterns, you can
insert the DFT to protect the RAM during shift: The RAM should have a write enable that is PI-
controllable throughout test mode to prevent destroying the state of the RAM. This ensures the
tool can create a state inside the macro and retain the state during the scan loading of the next
functional cycle (the next scan pattern after conversion by MacroTest).

The easiest case to identify is where the ATPG tool issues a message saying it can use the RAM
test mode, RAM_SEQUENTIAL. This message occurs because the tool can independently
operate the scan chains and the RAM. The tool can operate the scan chain without changing the
state of the macro as well as operate the macro without changing the state loaded into the scan
chain. This allows the most flexibility for ATPG, but the most DFT also.

However, there are cases where the tool can operate the scan chain without disturbing the
macro, while the opposite is not true. If the scan cells are affected or updated when the macro is
operated (usually because a single clock captures values into the scan chain and is also an input
into the macro), the tool cannot use RAM_SEQUENTIAL mode. Instead, the tool can use a
sequential MacroTest pattern (multiple cycles per scan load), or it can use multiple single cycle
patterns if the user’s patterns keep the write enable or write clock turned off during shift.

For example, suppose a RAM has a write enable that comes from a PI in test mode. This makes
it possible to retain written values in the RAM during shift. However, it also has a single edge-
triggered read control signal (no separate read enable) so the RAM’s outputs change any time
the address lines change followed by a pulse of the read clock/strobe. The read clock is a shared
clock and is also used as the scan clock to shift the scan chains (composed of MUX scan cells).
In this case, it is not possible to load the scan chains without changing the read values on the
output of the macro. For this example, you will need to describe a sequential read operation to
MacroTest. This can be a two-cycle operation. In the first cycle, MacroTest pulses the read
clock. In the second cycle, MacroTest observes and captures the macro outputs into the
downstream scan cells. This works because there is no intervening scan shift to change the
values on the macro’s output pins. If a PI-controllable read enable existed, or if you used a non-
shift clock (clocked scan and LSSD have separate shift and capture clocks), an intervening scan
load could occur between the pulse of the read clock and the capture of the output data. This is
possible because the macro read port does not have to be clocked while shifting the scan chain.

When to Use MacroTest

MacroTest is primarily used to test small memories (register file, cache, FIFO, and so on).
Although the ATPG tool can test the faults at the boundaries of such devices, and can propagate
the fault effects through them (using the _ram or _cram primitives), it does not attempt to create
a set of patterns to test them internally. This is consistent with how it treats all primitives.
Because memory primitives are far more complex than a typical primitive (such as a NAND
gate), you may prefer to augment ATPG tool patterns with patterns that you create to test the
internals of the more complex memory primitives. Such complex primitives are usually
packaged as models in the ATPG library, or as HDL modules that are given the generic name
“macro”.

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Note
Although the ATPG library has specific higher level collections of models called macros,
MacroTest is not limited to testing these macros. It can test library models and HDL
modules as well.

Here, the term “macro” simply means some block of logic, or even a distributed set of lines that
you want to control and observe. You must provide the input values and expected output values
for the macro. Typically you are given, or must create, a set of tests. You can then simulate
these tests in some time-based simulator, and use the results predicted by that simulator as the
expected outputs of the macro. For memories, you can almost always create both the inputs and
expected outputs without any time-based simulation. For example, you might create a test that
writes a value, V, to each address. It is trivial to predict that when subsequent memory reads
occur, the expected output value will be V.

MacroTest converts these functional patterns to scan patterns that can test the device after it is
embedded in systems (where its inputs and outputs are not directly accessible, and so the tests
cannot be directly applied and observed). For example, a single macro input enable might be the
output of two enables which are ANDed outside the macro. The tests must be converted so that
the inputs of the AND are values which cause the AND’s output to have the correct value at the
single macro enable input (the value specified by the user as the macro input value). MacroTest
converts the tests (provided in a file) and provides the inputs to the macro as specified in the
file, and then observes the outputs of the macro specified in the file. If a particular macro output
is specified as having an expected 0 (or 1) output, and this output is a data input to a MUX
between the macro output and the scan chain, the select input of that MUX must have the
appropriate value to propagate the macro’s output value to the scan chain for observation.
MacroTest automatically selects the path(s) from the macro output(s) to the scan chain(s), and
delivers the values necessary for observation, such as the MUX select input value in the
example above.

Often, each row of a MacroTest file converts to a single 1-system cycle scan test (sometimes
called a basic scan pattern in the ATPG tool). A scan chain load, PI assertion, output measure,
clock pulse, and scan chain unload result for each row of the file if you specify such patterns. To
specify a write with no expected known outputs, specify the values to apply at the inputs to the
device and give X output values (don't care or don't measure). To specify a read with expected
known outputs, specify both the inputs to apply, and the outputs that are expected (as a result of
those and all prior inputs applied in the file so far). For example, an address and read enable
would have specified inputs, whereas the data inputs could be X (don’t care) for a memory read.

Mentor Graphics highly recommends that you not over-specify patterns. It may be impossible,
due to the surrounding logic, to justify all inputs otherwise. For example, if the memory has a
write clock and write enable, and is embedded in a way that the write enable is independent but
the clock is shared with other memories, it is best to turn off the write using the write enable,
and leave the clock X so it can be asserted or de-asserted as needed. If the clock is turned off
instead of the write enable, and the clock is shared with the scan chain, it is not possible to pulse
the shared clock to capture and observe the outputs during a memory read. If instead, the write

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enable is shared and the memory has its own clock (not likely, but used for illustration), then it
is best to turn off the write with the clock and leave the shared write enable X.

Realize that although the scan tests produced appear to be independent tests, the tool assumes
that the sequence being converted has dependencies from one cycle to the next. Thus, the scan
patterns have dependencies from one scan test to the next. Because this is atypical, the tool
marks MacroTest patterns as such, and you must save such MacroTest patterns using the
write_patterns command. The MacroTest patterns cannot be reordered or reduced using
compress_patterns; reading back MacroTest patterns is not allowed for that reason. You must
preserve the sequence of MacroTest patterns as a complete, ordered set, all the way to the tester,
if the assumption of cycle-to-cycle dependencies in the original functional sequence is correct.

To illustrate, if you write a value to an address, and then read the value in a subsequent scan
pattern, this will work as long as you preserve the original pattern sequence. If the patterns are
reordered, and the read occurs before the write, the patterns will then mismatch during
simulation or fail on the tester. The reason is that the reordered scan patterns try to read the data
before it has been written. This is untrue of all other ATPG tool patterns. They are independent
and can be reordered (for example, to allow pattern compaction to reduce test set size).
Macrotest patterns are never reordered or reduced, and the number of input patterns directly
determines the number of output patterns.

Defining the Macro Boundary

The macro boundary is typically defined by its instance name with the macrotest command. If
no instance name is given, then the macro boundary is defined by a list of hierarchical pin
names (one per macro pin) given in the header of the MacroTest patterns file.

Defining a Macro Boundary by Instance Name

The macro is a particular instance, almost always represented by a top-level model in the ATPG
library. More than one instance may occur in the netlist, but each instance has a unique name
that identifies it. Therefore, the instance name is all that is needed to define the macro boundary.
The definition of the instance/macro is accessed to determine the pin order as defined in the port
list of the definition. MacroTest expects that pin order to be used in the file specifying the I/O
(input and expected output) values for the macro (the tests). For example, the command:

macrotest regfile_8 file_with_tests

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would specify for MacroTest to find the instance “regfile_8”, look up its model definition, and
record the name and position of each pin in the port list. Given that the netlist is written in
Verilog, with the command:

regfile_definition_name regfile_8 (net1, net2, ... );

the portlist of regfile_definition_name (not the instance port list “net1, net2, …”) is used to get
the pin names, directions, and the ordering expected in the test file, file_with_tests. If the library
definition is:

model "regfile_definition_name"
("Dout_0", "Dout_1", Addr_0", "Addr_1", "Write_enable", ...)
( input ("Addr_0") () ... output ("Dout_0") () ... )

then MacroTest knows to expect the output value Dout_0 as the first value (character)
mentioned in each row (test) of the file, file_with_tests. The output Dout_1 should be the 2nd
pin, input pin Addr_0 should be the 3rd pin value encountered, etc. If it is inconvenient to use
this ordering, the ordering can be changed at the top of the test file, file_with_tests. This can be
done using the following syntax:

macro_inputs Addr_0 Addr_1

macro_output Dout_1
macro_inputs Write_enable
...
end

which would cause MacroTest to expect the value for input Addr_0 to be the first value in each
test, followed by the value for input Addr_1, the expected output value for Dout_1, the input
value for Write_enable, and so on.

Note
Only the pin names need be specified, because the instance name “regfile_8” was given
for the macrotest command.

Defining a Macro Boundary Without Using an Instance

Name
If an instance name is not given with the macrotest command, then you must provide an entire
hierarchical path/pin name for each pin of the macro. This is given in the header of the
MacroTest patterns file. There must be one name per data bit in the data (test values) section
which follows the header. For example:
macro_inputs regfile_8/Addr_0regfile_8/Addr_1
macro_outputregfile_8/Dout_1
macro_inputsregfile_8/write_enable
...
end

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The above example defines the same macro boundary as was previously defined for regfile_8
using only pin names to illustrate the format. Because the macro is a single instance, this would
not normally be done, because the instance name is repeated for each pin. However, you can use
this entire pathname form to define a distributed macro that covers pieces of different instances.
This more general form of boundary definition allows a macro to be any set of pins at any
level(s) of hierarchy down to the top library model. If you use names which are inside a model
in the library, the pin pathname must exist in the flattened data structures. (In other words, it
must be inside a model where all instances have names, and it must be a fault site, because these
are the requirements for a name inside a model to be preserved in the tool).

This full path/pin name form of “macro boundary” definition is a way to treat any set of
pins/wires in the design as points to be controlled, and any set of pins/wires in the design as
points to be observed. For example, some pin might be defined as a macro_input which is then
given {0,1} values for some patterns, but X for others. In some sense, this “macro input” can be
thought of as a programmable ATPG constraint (see add_atpg_constraints), whose value can be
changed on a pattern by pattern basis. There is no requirement that inputs be connected to
outputs. It would even be possible to define a distributed macro such that the “output” is really
the input to an inverter, and the “input” is really the output of the same inverter. If the user
specified that the input = 0, and the expected output = 1, MacroTest would ensure that the
macro “input” was 0 (so the inverter output is 0, and its input is 1), and would sensitize the input
of the inverter to some scan cell in a scan chain. Although this is indeed strange, it is included to
emphasize the point that full path/pin forms of macro boundary definition are completely
flexible and unrelated to netlist boundaries or connectivity. Any set of connected or disjoint
points can be inputs and/or outputs.

Reporting and Specifying Observation Sites

You can report the set of possible observation sites using the macrotest command switch,
-Report_observation_candidates. This switch reports, for each macro output, the reachable scan
cells and whether the scan cell is already known to be unable to capture/observe. Usually, all
reachable scan cells can capture, so all are reported as possible observation sites. The report
gives the full instance name of the scan cell’s memory element, and its gate id (which follows
the name and is surrounded by parentheses).
Although rarely done, you can specify for one macro output at a time exactly which of those
reported scan cells is to be used to observe that particular macro output pin. Any subset can be
so specified. For example, if you want to force macro output pin Dout_1 to be observed at one
of its reported observation sites, such as “/top/middle/bottom/ (13125)”, then you can specify
this as follows:

macro_output regfile_8/Dout_1
observe_at13125

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Note
There can be only one macro_output statement on the line above the observe_at directive.
Also, you must specify only one observe_at site, which is always associated with the
single macro_output line that precedes it. If a macro_input line immediately precedes the
observe_at line, MacroTest will issue an error message and exit.

The preceding example uses the gate id (number in parentheses in the -Report output) to specify
the scan cell DFF to observe at, but you can also use the instance pathname. Instances inside
models may not have unique names, so the gate id is always an unambiguous way to specify
exactly where to observe. If you use the full name and the name does not exactly match, the tool
selects the closest match from the reported candidate observation sites. The tool also warns you
that an exact match did not occur and specifies the observation site that it selected.

Defining a Macro Boundary With Trailing Edge Inputs

MacroTest treats macros as black boxes, even if modelled, so do not assume that this
information will be gathered using connectivity. Assuming nothing is known about the macro’s
internals, Macrotest forces the user-specified expected outputs onto the macro outputs for each
pattern. This allows black-boxed macros to be used, or you to create models for normal ATPG
using the _cram primitive, but treat the macro as a black box for internal testing. A _cram
primitive may be adequate for passing data through a RAM, for example, but not for modelling
it for internal faults. Macrotest trusts the output values you provide regardless of what would
normally be calculated in the tool, allowing you to specify outputs for these and other situations.
Due to its black box treatment of even modelled RAMs/macros, MacroTest must sometimes get
additional information from you. Macrotest assumes that all macro inputs capture on the leading
edge of any clock that reaches them. So, for a negative pulse, MacroTest assumes that the
leading (falling) edge causes the write into the macro, whereas for a positive pulse, MacroTest
assumes that the leading (rising) edge causes the write. If these assumptions are not true, you
must specify which data or address inputs (if such pins occur) are latched into the macro on a
trailing edge.

Occasionally, a circuit uses leading DFF updates followed by trailing edge writes to the
memory driven by those DFFs. For trailing edge macro inputs, you must indicate that the
leading edge assumption does not hold for any input pin value that must be presented to the
macro for processing on the trailing edge. For a macro which models a RAM with a trailing
edge write, you must specify this fact for the write address and data inputs to the macro which
are associated with the falling edge write. To specify the trailing edge input, you must use a
boundary description which lists the macro’s pins (you cannot use the instance name only
form).

Regardless of whether you use just pin names or full path/pin names, you can replace
“macro_inputs” with “te_macro_inputs” to indicate that the inputs that follow must have their
values available for the trailing edge of the shared clock. This allows MacroTest to ensure that
the values arrive at the macro input in time for the trailing edge, and also that the values are not

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overwritten by any leading edge DFF or latch updates. If a leading edge DFF drives the trailing
edge macro input pin, the value needed at the macro input will be obtained from the D input side
of the DFF rather than its Q output. The leading edge will make Q=D at the DFF, and then that
new value will propagate to the macro input and be waiting for the trailing edge to use. Without
the user specification as a trailing edge input, MacroTest would obtain the needed input value
from the Q output of the DFF. This is because MacroTest would assume that the leading edge of
the clock would write to the macro before the leading edge DFF could update and propagate the
new value to the macro input.

It is not necessary to specify leading edge macro inputs because this is the default behavior. It is
also unnecessary to indicate leading or trailing edges for macro outputs. You can control the
cycle in which macro outputs are captured. This ensures that the tool correctly handles any
combination of macro outputs and capturing scan cells as long as all scan cells are of the same
polarity (all leading edge capture/observe or all trailing edge capture/observe).

In the rare case that a particular macro output could be captured into either a leading or a trailing
edge scan cell, you must specify which you prefer by using the -Le_observation_only switch or
-Te_observation_only switch with the macrotest command for that macro. For more
information on these switches, see “Example 3 — Using Leading Edge & Trailing Edge
Observation Only” and the macrotest description in the Tessent Shell Reference Manual.

An example of the TE macro input declaration follows:

macro_input clock
te_macro_inputs Addr_0 Addr_1 // TE write address inputs
macro_output Dout_1
...
end

Defining Test Values

The test file may consist of the following:
• Comments (a line starting with “//” or #)
• Blank lines
• An optional pin reordering section (which must come before any values) that begins
with “MACRO_INPuts” or “MACRO_OUTputs” and ends with “END”
• The tests (one cycle per row of the file)
Normal (nonpulseable) input pin values include {0,1,X,Z}. Some macro inputs may be driven
by PIs declared as pulseable pins (add_clocks, add_read_controls, and add_write_controls
specify these pins in the tool). These pins can have values from {P,N} where P designates a
positive pulse and N designates a negative pulse. Although you can specify a P or N on any pin,
the tool issues a warning if it cannot verify that the pin connects to a pulseable primary input
(PI). If the tool can pulse the control and cause a pulse at that macro pin, then the pulse will
occur. If it cannot, the pulse will not occur. Users are warned if they specify the wrong polarity

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of pulse (an N, for example, when there is a direct, non-inverting connection to a clock PI that
has been specified with an off value of 0, which means that it can only be pulsed positively). A
P would need to be specified in such a case, and some macro inputs would probably have to be
specified as te_macro_inputs since the N was probably used due to a negative edge macro. P
and N denote the actual pulse, not the triggering edge of the macro. It is the embedding that
determines whether a P or N can be produced.

Note
It is the declaration of the PI pin driving the macro input, not any declaration of the macro
input itself, which determines whether a pin can be pulsed in the tool.

Normal observable output values include {L,H}, which are analogous to {0,1}. L represents
output 0, and H represents output 1. You can give X as an output value to indicate Don't
Compare, and F for a Floating output (output Z). Neither a Z nor an X output value will be
observed. Occasionally an output cannot be observed, but must be known in order to prevent
bus contention or to allow observation of some other macro output.

If you provide a file with these characters, a check is done to ensure that an input pin gets an
input value, and an output pin gets an output value. If an “L” is specified in an input pin
position, for example, an error message is issued. This helps detect ordering mismatches
between the port list and the test file. If you prefer to use 0 and 1 for both inputs and outputs,
then use the -No_l_h switch with the macrotest command:

macrotest regfile_8 file_with_tests -no_l_h

Assuming that the -L_h default is used, the following might be the testfile contents for our
example register file, if the default port list pin order is used.

// Tests for regfile_definition_name.

//
// W
// r
// i
// t
// e
// _
// DD AA e
// oo dd n
// uu dd a
// tt rr b
// __ __ l
// 01 01 e

XX 00 0
XX 00 1
HH 00 0

The example file above has only comments and data; spaces are used to separate the data into
fields for convenience. Each row must have exactly as many value characters as pins mentioned
in the original port list of the definition, or the exact number of pins in the header, if pins were

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specified there. Pins can be left off of an instance if macro_inputs and macro_outputs are
specified in the header, so the header names are counted and that count is used unless the
instance name only form of macro boundary definition is used (no header names exist).

To specify less than all pins of an instance, omit the pins from the header when reordering the
pins. The omitted pins are ignored for purposes of MacroTest. If the correct number of values
do not exist on every row, an error occurs and a message is issued.

The following is an example where the address lines are exchanged, and only Dout_0 is to be
tested:

// Tests for regfile_definition_name testing only Dout_0

macro_output Dout_0
macro_inputs Addr_1 Addr_0 write_enable ..
...
end
// W
// r
// i
// t
// e
// _
// D AA e
// o dd n
// u dd a
// t rr b
// _ __ l
// 0 10 e

X 00 0
X 00 1
H 00 0

It is not necessary to have all macro_inputs together. You can repeat the direction designators as
necessary:

macro_input write_enable
macro_output Dout_0
macro_inputs Addr_1 Addr_0
macro_outputs Dout_1 ...
...
end

Recommendations for Using MacroTest

When using MacroTest, Mentor Graphics recommends that you begin early in the process. This
is because the environment surrounding a regfile or memory may prevent the successful
delivery of the original user specified tests, and Design-for-Test hardware may have to be added
to allow the tests to be delivered, or the tests may have to be changed to match the surroundings
so that the conversion can occur successfully.

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For example, if the write enable line outside the macro is the complement of the read enable line
(perhaps due to a line which drives the read enable directly and also fans out to an inverter
which drives the write enable), and you specify that both the read enable and write enable pins
should be 0 for some test, then MacroTest will be unable to deliver both values. It stops and
reports the line of the test file, as well as the input pins and values that cannot be delivered. If
you change the enable values in the MacroTest patterns file to always be complementary,
MacroTest would then succeed. Alternatively, if you add a MUX to make the enable inputs
independently controllable in test mode and keep the original MacroTest patterns unchanged,
MacroTest would use the MUX to control one of the inputs to succeed at delivering the
complementary values.

Once MacroTest is successful, you should simulate the resulting MacroTest patterns in a time-
based simulator. This verifies that the conversion was correct, and that no timing problems
exist. The tool does not simulate the internals of primitives, and therefore relies on the fact that
the inputs produced the expected outputs given in the test file. This final simulation ensures that
no errors exist due to modeling or simulation details that might differ from one simulator to the
next. Normal ATPG tool considerations hold, and it is suggested that DRC violations be treated
as they would be treated for a stuck-at fault ATPG run.

To prepare to MacroTest an empty (TieX) macro that needs to be driven by a write control (to
allow pulsing of that input pin on the black box), issue the set_macrotest_options command.
This command prevents a G5 DRC violation and allows you to proceed. Also, if a transparent
latch (TLA) on the control side of an empty macro is unobservable due to the macro, the
set_macrotest_options command prevents it from becoming a TieX, as would normally occur.
Once it becomes a TieX, it is not possible for MacroTest to justify macro values back through
the latch. If in doubt, when preparing to MacroTest any black box, issue the
set_macrotest_options command before exiting setup mode. No errors will occur because of
this, even if none of the conditions requiring the command exist.

ATPG commands and options apply within MacroTest, including cell constraints, ATPG
constraints, clock restrictions (it only pulses one clock per cycle), and others. If MacroTest fails
and reports that it aborted, you can use the set_abort_limit command to get MacroTest to work
harder, which may allow MacroTest to succeed. Mentor Graphics recommends that you set a
moderate abort limit for a normal MacroTest run, then increase the limit if MacroTest fails and
issues a message saying that a higher abort limit might help.

ATPG effort should match the simulation checks for bus contention to prevent MacroTest
patterns from being rejected by simulation. Therefore, if you specify set_contention_check On,
you should use the -Atpg option. Normally, if you use set_contention_check Capture_clock,
you should use the -Catpg option instead. Currently, MacroTest does not support the -Catpg
option, so this is not advised. Using the set_decision_order Random is strongly discouraged. It
can mislead the search and diagnosis in MacroTest.

In a MacroTest run, as each row is converted to a test, that test is stored internally (similar to a
normal ATPG run). You can save the patterns to write out the tests in a desired format (perhaps
Verilog to allow simulation and WGL for a tester). The tool supports the same formats for

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MacroTest patterns as for patterns generated by a normal ATPG run. However, because
MacroTest patterns cannot be reordered, and because the expected macro output values are not
saved with the patterns, it is not possible to read macrotest patterns back into the ATPG tool.
You should generate Macrotest patterns, then save them in all desired formats.

Note
The macro_output node in the netlist must not be tied to Z (floating).

MacroTest Examples
The following are examples of using MacroTest.

Example 1 — Basic 1-Cycle Patterns

Verilog Contents:

RAM mem1 (.Dout ({ Dout[7],Dout[6],Dout[5],Dout[4],Dout[3],

Dout[2], Dout[1], Dout[0]}) ,
.RdAddr ({ RdAddr[1] , RdAddr[0] }) ,
.RdEn ( RdEn ) ,
.Din ({ Din[7] , Din[6] , Din[5] , Din[4] , Din[3] ,
Din[2] , Din[1] , Din[0] }) ,
.WrAddr ({ WrAddr[1] , WrAddr[0] }) , .WrEn ( WrEn ));

ATPG Library Contents:

model RAM (Dout, RdAddr, RdEn, Din, WrAddr, WrEn) (

input (RdAddr,WrAddr) (array = 1 : 0;)
input (RdEn,WrEn) ()
input (Din) (array = 7 : 0;)
output (Dout) (
array = 7 : 0;
data_size = 8;
address_size = 2;
read_write_conflict = XW;
primitive = _cram(,,
_write {,,} (WrEn,,WrAddr,Din),
_read {,,,} (,RdEn,,RdAddr,Dout)
);
)
)

Note
Vectors are treated as expanded scalars.

Because Dout is declared as “array 7:0”, the string “Dout” in the port list is equivalent to
“Dout<7> Dout<6> Dout<5> Dout<4> Dout<3> Dout<2> Dout<1> Dout<0>”. If the

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declaration of Dout had been Dout “array 0:7”, then the string “Dout” would be the reverse of
the above expansion. Vectors are always allowed in the model definitions. Currently, vectors
are not allowed in the macrotest input patterns file, so if you redefine the pin order in the header
of that file, scalars must be used. Either “Dout<7>”, “Dout(7)”, or “Dout[7]” can be used to
match a bit of a vector.

Dofile Contents:

set_system_mode analysis
macrotest mem1 ram_patts2.pat
write_patterns results/pattern2.f -replace

Test File Input (ram_patts2.pat) Contents:

// model RAM (Dout, RdAddr, RdEn, Din, WrAddr, WrEn) (

// input (RdAddr,WrAddr) (array = 1 : 0;)
// input (RdEn,WrEn) ()
// input (Din) (array = 7 : 0;)
//
// output (Dout) (
// array = 7 : 0;
// data_size = 8;
// address_size = 2;
// .....
// Write V1 (data vector 1) to address 0. Data Outputs
// and Read Address are Don’t Cares.
XXXXXXXX XX 0 10101010 00 P
// Read V1 from address 0. Data Inputs and Write Address
// are Don’t Cares.
HLHLHLHL 00 1 XXXXXXXX XX 0
XXXXXXXX XX 0 0x010101 01 P // Write V2 to address 1.
LXLHLHLH 01 1 xxxxxxxx xx 0 // Read V2 from address 1.

Converted Test File Output (results/pattern2.f) Contents:

... skipping some header information ....

SETUP =
declare input bus "PI" = "/clk", "/Datsel",
"/scanen_early", "/scan_in1", "/scan_en",
.... skipping some declarations ....

declare output bus "PO" = "/scan_out1";

.... skipping some declarations ....

CHAIN_TEST =
pattern = 0;
apply "grp1_load" 0 =
chain "chain1" = "0011001100110011001100";
end;
apply "grp1_unload" 1 =
chain "chain1" = "0011001100110011001100";
end;
end;

SCAN_TEST =

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pattern = 0 macrotest ;
apply "grp1_load" 0 =
chain "chain1" = "0110101010000000000000";
end;
force "PI" "001X0XXXXXXXX" 1;
pulse "/scanen_early" 2;
measure "PO" "1" 3;
pulse "/clk" 4;
apply "grp1_unload" 5 =
chain "chain1" = "XXXXXXXXXXXXXXXXXXXXXX";
end;

pattern = 1 macrotest ;
apply "grp1_load" 0 =
chain "chain1" = "1000000000000000000000";
end;
force "PI" "001X0XXXXXXXX" 1;
measure "PO" "1" 2;
pulse "/clk" 3;
apply "grp1_unload" 4=
chain "chain1" = "XXXXXXXXXXXXXX10101010";
end;
... skipping some output ...

SCAN_CELLS =
scan_group "grp1" =
scan_chain "chain1" =
scan_cell = 0 MASTER FFFF "/rden_reg/ffdpb0"...
scan_cell = 1 MASTER FFFF "/wren_reg/ffdpb0"...
scan_cell = 2 MASTER FFFF "/datreg1/ffdpb7"...
... skipping some scan cells ...
scan_cell = 20 MASTER FFFF "/doutreg1/ffdpb1"...
scan_cell = 21 MASTER FFFF "/doutreg1/ffdpb0"...
end;
end;
end;

Example 2— Synchronous Memories (1- & 2-Cycle Patterns)

Verilog Contents:

For this example, the RAM is as before, except a single clock is connected to an edge-triggered
read and edge-triggered write pin of the macro to be tested. It is also the clock going to the
MUX scan chain. There is also a separate write enable. As a result, it is possible to write using a
one-cycle pattern, and then to preserve the data written during shift by turning the write enable
off in the shift procedure. However, for this example, a read must be done in two cycles—one to
pulse the RAM’s read enable and make the data come out of the RAM, and another to capture
that data into the scan chain before shifting changes the RAM’s output values. There is no
independent read enable to protect the outputs during shift, so they must be captured before
shifting, necessitating a 2 cycle read/observe.

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ATPG Library Contents:

model RAM (Dout, RdAddr, RdClk, Din, WrAddr, WrEn, WrClk) (

input (RdAddr,WrAddr) (array = 1 : 0;)
input (RdClk,WrEn, WrClk) ()
input (Din) (array = 7 : 0;)
output (Dout) (
array = 7 : 0;
data_size = 8;
edge_trigger = rw;
address_size = 2;
read_write_conflict = XW;
primitive = _cram(,,
_write {,,} (WrClk,WrEn,WrAddr,Din),
_read {,,,} (,RdClk,,RdAddr,Dout)
);
)
)

Note that because the clock is shared, it is important to only specify one of the macro values for
RdClk or WrClk, or to make them consistent. X means “Don’t Care” on macro inputs, so it will
be used to specify one of the two values in all patterns to ensure that any external embedding
can be achieved. It is easier to not over-specify MacroTest patterns, which allows using the
patterns without having to discover the dependencies and change the patterns.

Dofile Contents:

set_system_mode analysis
macrotest mem1 ram_patts2.pat
write_patterns results/pattern2.f -replace

Test File Input (ram_patts2.pat) Contents:

// model RAM (Dout, RdAddr, RdClk, Din, WrAddr, WrEn, WrClk) (

// input (RdAddr,WrAddr) (array = 1 : 0;)
// input (RdClk,WrEn, WrClk) ()
// input (Din) (array = 7 : 0;)
//
// output (Dout) (
// array = 7 : 0;
// data_size = 8;
// edge_trigger = rw;
// .....
// Write V1 (data vector 1) to address 0.
XXXXXXXX XX X 10101010 00 1 P
// Read V1 from address 0 -- next 2 rows (1 row per cycle).
XXXXXXXX 00 P XXXXXXXX XX 0 X + // + indicates another cycle.
HLHLHLHL XX X XXXXXXXX XX 0 X // Values observed this cycle.
XXXXXXXX XX X 01010101 01 1 P // Write V2 to address 1.
xxxxxxxx 01 P xxxxxxxx xx 0 X + // Read V2,address 1,cycle 1.
LHLHLHLH XX X XXXXXXXX XX 0 X // Read V2,address 1,cycle 2.

Converted Test File Output (results/pattern2.f) Contents:

... skipping some header information ....

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SETUP =
declare input bus "PI" = "/clk", "/Datsel",
"/scanen_early", "/scan_in1", "/scan_en",
.... skipping some declarations ....

declare output bus "PO" = "/scan_out1";

.... skipping some declarations ....

CHAIN_TEST =
pattern = 0;
apply "grp1_load" 0 =
chain "chain1" = "0011001100110011001100";
end;
apply "grp1_unload" 1 =
chain "chain1" = "0011001100110011001100";
end;
end;

SCAN_TEST =

pattern = 0 macrotest ;
apply "grp1_load" 0 =
chain "chain1" = "0110101010000000000000";
end;
force "PI" "001X0XXXXXXXX" 1;
pulse "/scanen_early" 2;
measure "PO" "1" 3;
pulse "/clk" 4;
apply "grp1_unload" 5 =
chain "chain1" = "XXXXXXXXXXXXXXXXXXXXXX";
end;

pattern = 1 macrotest ;
apply "grp1_load" 0 =
chain "chain1" = "1000000000000000000000";
end;
force "PI" "001X0XXXXXXXX" 1;
pulse "/clk" 2;
force "PI" "001X0XXXXXXXX" 3;
measure "PO" "1" 4;
pulse "/clk" 5;
apply "grp1_unload" 6=
chain "chain1" = "XXXXXXXXXXXXXX10101010";
end;
... skipping some output ...

Example 3 — Using Leading Edge & Trailing Edge Observation Only

Assume that a clock with an off value of 0 (positive pulse) is connected through buffers to a
rising edge read input of a macro, and also to both rising and falling edge D flip-flops. Either of
the flip-flops can capture the macro’s output values for observation. If you specify that the
outputs should be captured in the same cycle as the read pulse, then this will definitely occur if
you invoke MacroTest with the -Te_observation_only switch because only the trailing edge
(TE) flip-flops will be selected for observation. The rising edge of the clock triggers the macro’s

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read, the values propagate to the scan cells in that same cycle, and then the falling edge of the
clock captures those values in the TE scan cells.

On the other hand, if you invoke MacroTest with the -Le_observation_only switch and indicate
in the MacroTest patterns that the macro’s outputs should be observed in the cycle after pulsing
the read pin on the macro, the rising edge of one cycle would cause the read of the macro, and
then the rising edge on the next cycle would capture into the TE scan cells.

These two command switches (-Te_observation_only and -Le_observation_only) ensure that

MacroTest behaves in a manner that is compatible with the particular macro and its embedding.
In typical cases, only one kind of scan cell is available for observation and the MacroTest
patterns file would, of course, need to be compatible. These options are only needed if both
polarities of scan cells are possible observation sites for the same macro output pin.

For additional information on the use of these switches, refer to the macrotest description in the
Tessent Shell Reference Manual.

Verifying Test Patterns

After testing the functionality of the circuit with a simulator, and generating the test vectors
with the ATPG tool, you should run the test vectors in a timing-based simulator and compare
the results with predicted behavior from the ATPG tools. This run will point out any
functionality discrepancies between the two tools, and also show timing differences that may
cause different results. The following subsections further discuss the verification you should
perform.

Simulating the Design with Timing

At this point in the design process, you should run a full timing verification to ensure a match
between the results of golden simulation and ATPG. This verification is especially crucial for
designs containing asynchronous circuitry.
You should have already saved the generated test patterns with the write_patterns command.
The tool saved the patterns in parallel unless you used the -Serial switch to save the patterns in
series. You can reduce the size of a serial pattern file by using the -Sample switch; the tool then
saves samples of patterns for each pattern type, rather than the entire pattern set (except
MacroTest patterns, which are not sampled nor included in the sampled pattern file). This is
useful when you are simulating serial patterns because the size of the sampled pattern file is
reduced and thus, the time it takes to simulate the sampled patterns is also reduced.

Note
Using the -Start and -End switches will limit file size as well, but the portion of internal
patterns saved will not provide a very reliable indication of pattern characteristics when
simulated. Sampled patterns will more closely approximate the results you would obtain
from the entire pattern set.

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The Verilog Test Bench

If you selected -Verilog as the format in which to save the patterns, the application
automatically creates a test bench that you can use in a timing-based simulator such as
ModelSim to verify that the tool-generated vectors behave as predicted by the ATPG tools.
For example, assume you saved the patterns generated as follows:

ANALYSIS> write_patterns pat_parallel.v -verilog -replace

The tool writes the test patterns out in one or more pattern files and an enhanced Verilog test
bench file that instantiates the top level of the design. These files contain procedures to apply
the test patterns and compare expected output with simulated output.

After compiling the patterns, the scan-inserted netlist, and an appropriate simulation library,
you simulate the patterns in a Verilog simulator. If there are no miscompares between the
ATPG tool’s expected values and the values produced by the simulator, a message reports that
there is “no error between simulated and expected patterns.” If any of the values do not match, a
simulation mismatch has occurred and must be corrected before you can use the patterns on a
tester.

Verilog Plusargs
The default Verilog test bench supports the following Verilog plusargs:
• STARTPAT — Sets the starting pattern number for the simulation.
• ENDPAT — Sets the ending pattern number for the simulation.
• CHAINTEST — Makes the STARTPAT and ENDPAT plusargs apply to the chain test
patterns instead of scan test.
• END_AFTER_SETUP — Causes the test bench to simulate only the test_setup vectors
and then finish without simulating any of the other patterns.
• SKIP_SETUP — Causes the test bench to skip simulating the test_setup vectors and
start simulation with the first pattern (either chain or scan test, whichever is present).
• CONFIG — Specifies a name of the .cfg file, which controls which .vec files are
simulated. For information about using this plusarg, refer to “CONFIG Usage” on
page 304.

STARTPAT, ENDPAT, and CHAINTEST Usage

By default (CHAINTEST=0), STARTPAT and ENDPAT specify the starting and ending scan
test patterns. When CHAINTEST=1, STARTPAT and ENDPAT specify the starting and ending
chain test patterns. So, specifying a STARTPAT of 0 without specifying CHAINTEST=1 skips
the chain test patterns and simulates only the scan test patterns. Specifying an ENDPAT of a
certain number and specifying CHAINTEST=1 causes simulation up to that number of chain
test patterns and no simulation of scan test patterns. Specifying a STARTPAT and

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CHAINTEST=1 causes simulation starting from the chain test specified, and also simulation of
all scan test patterns. What is not supported is the ability to start simulation at a certain chain
test pattern and then end at a certain scan test pattern.

CONFIG Usage
If you have multiple test benches for the same design but each for different pattern sets, you
need compile only one of the test benches and then specify the .cfg files from the other pattern
sets in order to simulate those patterns. From the same design and setup information, you can
create patterns within the tool and then write these out as a Verilog test bench. You can then
create different pattern sets for the same design and setup information (that is, don't change
clocks, scan chains, or edt logic) and then also write out as Verilog test benches. You can save
the first test bench and compile in ModelSim, and then delete the test bench and .name files for
all of the other pattern sets, saving only the .cfg and .vec files. Then, using the one test bench,
you can use the CONFIG plusarg to specify which pattern set to simulate and use the other
plusargs to control the range of patterns to simulate.

Example
As an example of how to use plusargs, the following command line invokes ModelSim and
simulates four scan test patterns:

vsim <testbench_top> -c -do "run -all" +STARTPAT=5 +ENDPAT=8

Parallel Versus Serial Patterns

Be sure to simulate parallel patterns and at least a few serial patterns. Parallel patterns simulate
relatively quickly, but do not detect problems that occur when data is shifted through the scan
chains. One such problem, for example, is data shifting through two cells on one clock cycle
due to clock skew. Serial patterns can detect such problems. Another reason to simulate a few
serial patterns is that correct loading of shadow or copy cells depends on shift activity. Because
parallel patterns lack the requisite shift activity to load shadow cells correctly, you may get
simulation mismatches with parallel patterns that disappear when you use serial patterns.
Therefore, always simulate at least the chain test or a few serial patterns in addition to the
parallel patterns.
For a detailed description of the differences between serial and parallel patterns, refer to the first
two subsections under “Pattern Formatting Issues” on page 370. See also “Sampling to Reduce
Serial Loading Simulation Time” on page 371 for information on creating a subset of sampled
serial patterns. Serial patterns take much longer to simulate than parallel patterns (due to the
time required to serially load and unload the scan chains), so typically only a subset of serial
patterns is simulated.

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Potential Causes of Simulation Mismatches

Simulation mismatches can have any number of causes; consequently, the most challenging part
of troubleshooting them is knowing where to start. Because a lot of information is available,
your first step should be to determine the likeliest potential source of the mismatch.
Note
Before troubleshooting simulation mismatches, be sure that the chain test has run without
error. You should resolve any problem with the scan chain before investigating
simulation mismatches.

Figure 7-50 is a suggested flow to help you investigate causes of simulation mismatches.

Figure 7-50. Mismatch Diagnosis Guidelines

Start

When, Where, and How Many Mismatches?

Serial Clock Skew

Chain Test Y Problem
Fails? (page 318)

All
Scan Tests Y Timing Violations
Fail?

Library Problems

Parallel
Patterns Fail, DRC Issues
Shadow Cells Y
Serial Pass?
N

If you are viewing this document online, you can click on the links in the figure to see more
complete descriptions of issues often at the root of particular mismatch failures. These issues
are discussed in the following sections:

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• When, Where, and How Many Mismatches?

• DRC Issues
• Shadow Cells
• Library Problems
• Timing Violations
• Analyzing the Simulation Data
• Analyzing Patterns
• Checking for Clock-Skew Problems with Mux-DFF Designs

When, Where, and How Many Mismatches?

If DRC violations do not seem to be a problem, you need to take a closer look at the mismatches
and check the following:
• Are the mismatches reported on primary outputs (POs), scan cells or both?
Mismatches on scan cells can be related to capture ability and timing problems on the
scan cells. For mismatches on primary outputs, the issue is more likely to be related to
an incorrect value being loaded into the scan cells.
• Are the mismatches reported on just a few or most of the patterns? Mismatches on a
few patterns indicates a problem that is unique to certain patterns, while mismatches on
most patterns indicate a more generalized problem.
• Are the mismatches observed on just a few pins/cells or most pins/cells? Mismatches
on a few pins/cells indicates a problem related to a few specific instances or one part of
the logic, while mismatches on most patterns indicate that something more general is
causing the problem.
• Do both the serial and the parallel test bench fail or just one of them? A problem in
the serial test bench only, indicates that the mismatch is related to shifting of the scan
chains (for example, data shifting through two cells on one clock cycle due to clock
skew). The problem with shadows mentioned in the preceding section, causes the serial
test bench to pass and the parallel test bench to fail.
• Does the chain test fail? As described above, serial pattern failure can be related to
shifting of the scan chain. If this is true, the chain test (which simply shifts data from
scan in to scan out without capturing functional data) also fails.
• Do only certain pattern types fail? If only ram sequential patterns fail, the problem is
most certainly related to the RAMs (for instance incorrect modeling). If only
clock_sequential patterns fail, the problem is probably related to non-scan flip-flops and
latches.

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DRC Issues
The DRC violations that are most likely to cause simulation mismatches are C6 and T24.
For details on these violations, refer to “Design Rule Checking” in the Tessent Shell Reference
Manual and SupportNet KnowledgeBase TechNotes describing each of these violations. For
most DRC-related violations, you should be able to see mismatches on the same flip-flops
where the DRC violations occurred.

You can avoid mismatches caused by the C6 violation by enabling the set_clock_off_simulation
command.

Shadow Cells
Another common problem is shadow cells. Such cells do not cause DRC violations, but the tool
issues the following message when going into analysis mode:
// 1 external shadows that use shift clocking have been identified.

A shadow flip-flop is a non-scan flip-flop that has the D input connected to the Q output of a
scan flip-flop. Under certain circumstances, such shadow cells are not loaded correctly in the
parallel test bench. If you see the above message, it indicates that you have shadow cells in your
design and that they may be the cause of a reported mismatch. For more information about
shadow cells and simulation mismatches, consult the online SupportNet KnowledgeBase. Refer
to “Mentor Graphics Support” on page 472 for information about SupportNet.

Library Problems
A simulation mismatch can be related to an incorrect library model; for example, if the reset
input of a flip-flop is modeled as active high in the analysis model used by the tool, and as
active low in the Verilog model used by the simulator. The likelihood of such problems depends
on the library. If the library has been used successfully for several other designs, the mismatch
probably is caused by something else. On the other hand, a newly developed, not thoroughly
verified library could easily cause problems. For regular combinational and sequential elements,
this causes mismatches for all patterns, while for instances such as RAMs, mismatches only
occur for a few patterns (such as RAM sequential patterns).
Another library-related issue is the behavior of multi-driven nets and the fault effect of bus
contention on tristate nets. The ATPG tool is conservative by default, so non-equal values on
the inputs to non-tristate multi-driven nets, for example, always results in an X on the net. For
additional information, see the set_net_resolution and set_net_dominance commands.

Timing Violations
Setup and hold violations during simulation of the test bench can indicate timing-related
mismatches. In some cases, you see such violations on the same scan cell that has reported

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mismatches; in other cases, the problem might be more complex. For instance, during loading
of a scan cell, you may observe a violation as a mismatch on the cell(s) and PO(s) that the
violating cell propagates to. Another common problem is clock skew.
This is discussed in “Checking for Clock-Skew Problems with Mux-DFF Designs” on
page 318.

Another common timing related issue is that the timeplate and/or test procedure file has not
expanded. By default, the test procedure and timeplate files have one “time unit” between each
event. When you create test benches using the -Timingfile switch with the write_patterns
command, the time unit expands to 1000 ns in the Verilog test benches. When you use the
default -Procfile switch and a test procedure file with the write_patterns command, each time
unit in the timeplate is translated to 1 ns. This can easily cause mismatches.

Analyzing the Simulation Data

If you still have unresolved mismatches after performing the preceding checks, examine the
simulation data thoroughly and compare the values observed in the simulator with the values
expected by the tool. The process you would use is very similar in the Verilog test benches.

Resolving Mismatches Using Simulation Data

When simulated values do not match the values expected by the ATPG tool, the enhanced
Verilog parallel test bench reports the time, pattern number, and scan cell or primary output
where each mismatch occurred. The serial test bench reports only output and time, so it is more
challenging to find the scan cell where the incorrect value has been captured.
Based on the time and scan cell where the mismatch occurred, you can generate waveforms or
dumps that display the values just prior to the mismatch. You can then compare these values to
the values that the tool expected. With this information, you can trace back in the design (in
both the ATPG tool and the simulator) to see where the mismatch originates.

When comparing Verilog simulation data to ATPG tool data, it is helpful to use the
SIM_VECTYPE_SIGNAL keyword in the parameter file. When this keyword is used, the
Verilog test bench will include additional keywords that makes it easier for you to understand
the sequence of events in the test bench, and also how to compare data between the ATPG tool
and the simulator. In the example simulation transcript in Figure 7-51, a mismatch is reported
for patterns 1 and 6. For pattern 6, the mismatch is reported at time 1170.

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Figure 7-51. Simulation Transcript

Note that the waveforms include the values for the flip-flop with mismatches, as well as the
signals mgcdft_shift, mgc_dft_launch_capture, and _pattern_count. Note that the
_pattern_count variable will increment just prior to the capture procedure. That means that
when _pattern_count variable is 6 and mgcdft_shift is 1, data is shifted out for pattern 6 (and
shifted in for pattern 7). By using these signals as a guide, you can see that the time of the
mismatch is during the shift procedure after pattern 6. Note that the capture for pattern 6 occurs
between time 1080 and 1120, when the mgcdft_launch_capture signal is high.

To see the corresponding data in the ATPG tool, you can use the set_gate_report
<pattern_index> command to see data for pattern 6. The data shown in the ATPG tool
corresponds to the state just before the capture clock pulse. In this example, for pattern 6, the

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ATPG tool shows data corresponding to the simulation values just prior to the clock pulse at
time 1100 in the waveform. A detailed example showing this process for a Verilog test bench is
contained in AppNote 3002, available on http://supportnet.mentor.com.

Analyzing Simulation Mismatches

By default, the simulation VCD file, test patterns, and design data are all analyzed and the
mismatch sources are identified. Once the analysis is complete, you can use DFTVisualizer to
graphically display the overlapping data and pinpoint the source of each mismatch.
The ATPG simulation mismatch analysis functionality is enhanced to optimize the debugging
of large (100K gates and more) designs and to support all simulators and distributed processing.
Once analysis is complete, DFTVisualizer graphically displays the source of the mismatches for
easy identification.

Understanding the Simulation Mismatch Analysis Flow

In general, the simulation mismatch analysis flow includes the following stages:
• Stage 1 — ATPG
• Stage 2 — Verilog test bench simulation
• Stage 3 — Debug test bench generation
• Stage 4 — Debug test bench simulation
• Stage 5 — Mismatch source identification
You can analyze simulation mismatches using the following methods:

• Automatically Analyzing Simulation Mismatches — Using this method, the tool runs
the entire flow, from Stage 1 — ATPG through Stage 5 — Mismatch source
identification using a single command invocation— see Figure 7-52.
• Manually Analyzing Simulation Mismatches — Using this method, you can run
mismatch analysis flow in steps. For example, you correct known issues in your failure
file and, instead of re-running Stage 2 — Verilog test bench simulation, you can proceed
to Stage 3 — Debug test bench generation using the modified failure file.
Both the automatic and manual flows follow the identical stages: the key difference is the
automatic flow runs the entire flow while the manual flow allows you to use the flow in steps.
The following sections cover the simulation mismatch analysis flow in detail.

Note
This procedure does not support debugging MacroTest or chain test patterns.

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Stage 1 — ATPG
Generate the flattened model, test patterns, and Verilog test bench (mentor_default.v). The test
patterns must be saved in a format that can be read back into the ATPG tool (binary, ASCII,
STIL, or WGL), and the test bench must be able to generate a failure file.

Stage 2 — Verilog test bench simulation

Simulating the Verilog test bench generates a failure file (mentor_default.v.fail). If no
simulation mismatches are found, the automatic simulation mismatch analysis stops.

Stage 3 — Debug test bench generation

Generate a test bench for just the failing patterns (mentor_default.v_vcdtb.v) by using the ATPG
tool to read the flattened netlist and read in the test patterns generated in Stage 1 and the failure
file from Stage 2. The test bench is set up to output a simulation results VCD file.

Stage 4 — Debug test bench simulation

In the same ATPG tool session, simulate the debug test bench generated in Stage 3. This
simulation produces the VCD file (mentor_default.v_debug.vcd).

Stage 5 — Mismatch source identification

Load the VCD file and trace each mismatch to its source. Then report the mismatch sources.

Note
All generated files are placed in the work_dft_debug directory inside your working
directory. This directory is created if it does not already exist.

Automatically Analyzing Simulation Mismatches

The following procedure shows how to use the tool to automatically analyze simulation
mismatches.
Figure 7-52 shows the automatic simulation mismatch analysis flow. Using this procedure, you
invoke the tool using the analyze_simulation_mismatches command with the -auto switch, and
the tool automatically runs the all stages of the flow. This procedure supports using a third-party
simulator and distributing processing to a remote server.

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Figure 7-52. Automatic Simulation Mismatch Analysis Flow

Prerequisites
• (Optional) If you are using an external third-party simulator, you must create a script to
invoke, setup, and run the simulator. See “Example” on page 318.
• (Optional) If you want to distribute the simulation part of the analysis, you must have
access to a remote server that can be accessed through rsh. For more information, see the
analyze_simulation_mismatches description in the Tessent Shell Reference Manual.

Procedure
Note
All generated files are placed in the work_dft_debug directory inside your working
directory. This directory is created if it does not already exist.

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1. Use the ATPG tool to read a design netlist or flat model. For example:
$ tessent -shell
SETUP> set_context patterns -scan
SETUP> read_verilog data/design.v

2. Specify the scan data for the scan cells in the design and switch to analysis system mode.
For example:
SETUP> add_scan_groups ...
SETUP> add_scan_chains ...
SETUP> add_clocks ...
SETUP> set_system_mode analysis

3. Specify the source test patterns for the design. For example:
ANALYSIS> read_patterns pats/testpat.bin

4. If you are using a third-party simulator, specify your simulator invoke script. For
example:
ANALYSIS> set_external_simulator -simulation_script runsim

For more information, see Example and the set_external_simulator description in the
Tessent Shell Reference Manual.
5. Run the automatic mismatch analysis. For example:
ANALYSIS> analyze_simulation_mismatches -auto -external_patterns

By default, the analysis runs on the local server. To run the simulation portion of the
analysis on a remote server, use the -host option. For example:
ANALYSIS> analyze_simulation_mismatches -auto -external_patterns -host abc_test

For more information, see the analyze_simulation_mismatches description in the

Tessent Shell Reference Manual.
By default, the analysis compares the specified failure file to the current test pattern
source to verify that both are generated from the same version of the design. If files do
not match, an error displays and the process aborts.
Once the test patterns and failure file pass the verification, a test bench is created
specifically for the mismatches in the failure file and simulated. The simulation results
are compared with the test patterns and design data to determine the source of simulation
mismatches listed in the failure file.
6. Open DFTVisualizer to view and further debug the mismatches. For example:
ANALYSIS> open_visualizer

7. Click Debug Simulation Mismatches. The Select a Mismatch ID dialog box displays.

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8. Select the ID for the mismatch to debug, and click Analyze.

DFTVisualizer displays and highlights overlapping design, simulation, and test pattern
data for the selected simulation mismatch.

Tip: Optionally, you can use the report_mismatch_sources command to replace steps 6
through 8 and automatically display the mismatches in DFTVisualizer.

If you issue report_mismatch_sources with the -display option, the Wave window
displays the following signals in addition to the instances associated with the displayed
mismatch source:

mgcdft_test_setup, mgcdft_load_unload, mgcdft_single_shift, mgcdft_shift_extra,

mgcdft_launch_capture

You can use the waveform of these signals to easily determine the active test procedure in
any displayed cycle. The signal value will be 1 to designate the current active procedure.

Note
In the Debug window, pin names assigned to the value “.” indicate that the VCD debug
test bench did not capture the VCD value for that location. It is not possible to capture
VCD values for every node in the design due to very large file sizes and run time; the test
bench only captures values in the combination cone behind the failure location, which in
most cases provides sufficient information to explain the mismatch.

For a complete list of possible pin name values resulting from simulation mismatches, see
the report_mismatch_sources description in the Tessent Shell Reference Manual.

Manually Analyzing Simulation Mismatches

Use this procedure to manually debug simulation mismatches.
With this procedure, you manually perform the simulation mismatch analysis using the
analyze_simulation_mismatches command. This procedure supports using a third-party
simulator and distributing processing to a remote server.

Figure 7-53 shows the manual simulation mismatch analysis flow.

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Figure 7-53. Manual Simulation Mismatch Analysis Flow

Prerequisites
• (Optional) If you are using an external third-party simulator, you must create a script to
invoke, setup, and run the simulator. See “Example” on page 318.
• (Optional) If you want to distribute the simulation part of the analysis, you must have
access to a remote server that can be accessed through rsh. For more information, see the
analyze_simulation_mismatches description in the Tessent Shell Reference Manual.
• A design netlist or flat model and the associated test patterns are available.
• Test patterns must have been verified and mismatches exist.
• In order to generate a failure file for a manually generated test bench, you must set the
“SIM_DIAG_FILE” parameter file keyword to 2 or 1(default) prior to ATPG.

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• For a manual simulation, you must set the “_write_DIAG_file” parameter to 1 in the
Verilog test bench to generate the failure file. This is done automatically if you set the
SIM_DIAG_FILE parameter file keyword to 2 prior to ATPG.

Procedure
Note
All generated files are placed in the work_dft_debug directory inside your working
directory. This directory is created if it does not already exist.

1. Use the ATPG tool to read a design netlist or flat model. For example:
$ tessent -shell
SETUP> set_context patterns -scan
SETUP> read_verilog data/design.v

2. Specify the scan data for the scan cells in the design and switch to analysis system mode.
For example:
SETUP> add_scan_groups ...
SETUP> add_scan_chains ...
SETUP> add_clocks ...
SETUP> set_system_mode analysis

3. Specify the source test patterns for the design. For example:
> read_patterns pats/testpat.bin

4. If you are using a third-party simulator, specify your simulator invoke script. For
example:
> set_external_simulator -simulation_script runsim

For more information, see Example and the set_external_simulator description in the
Tessent Shell Reference Manual.
5. Run the mismatch analysis. For example:
> analyze_simulation_mismatches -external_patterns

By default, the analysis runs on the local server. To run the simulation portion of the
analysis on a remote server, use the -host option. For example:
ANALYSIS> analyze_simulation_mismatches -external_patterns -host abc_test

For more information, see the analyze_simulation_mismatches description in the

Tessent Shell Reference Manual.

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By default, the analysis compares the specified failure file to the current test pattern
source to verify that both are generated from the same version of the design. If files do
not match, an error displays and the process aborts.
Once the test patterns and failure file pass the verification, a test bench is created
specifically for the mismatches in the failure file and simulated. The simulation results
are compared with the test patterns and design data to determine the source of simulation
mismatches listed in the failure file.
6. Simulate the Verilog test bench using ModelSim or a third-party simulator to create a
failure file—see the set_external_simulator command for details.
You must also create a script to set up and run an external simulator for the subsequent
steps of this procedure.
7. After simulation, perform a simulation mismatch analysis using the
analyze_simulation_mismatches command with the -FAilure_file switch and argument
to generate a test bench for just the failing patterns. For example:
ANALYSIS> analyze_simulation_mismatches -failure_file pat.fs.v.fail

You must use the flattened netlist and test patterns from the initial ATPG session.
This step creates the test bench for just the failing patterns (mentor_default.v_vcdtb.v).
The test bench is set up to output the simulation results to a VCD file.
8. In the same ATPG tool session, simulate the test bench for the failing patterns using the
analyze_simulation_mismatches command with the -TEstbench_for_vcd switch
argument. For example:
ANALYSIS> analyze_simulation_mismatches -testbench_for_vcd mentor_default.v_vcdtb.v

This step produces a VCD file (mentor_default.v_debug.vcd).

9. After simulation, load the VCD file. For example:
ANALYSIS> analyze_simulation_mismatches -vcd_file mentor_default.v_debug.vcd

10. Report the mismatch sources using the report_mismatch_sources command.

You can also view and further debug the mismatches using DFTVisualizer.
11. Open DFTVisualizer to view and further debug the mismatches. For example:
ANALYSIS> open_visualizer

12. Click Debug Simulation Mismatches. The Select a Mismatch ID dialog box displays.
13. Select the ID for the mismatch to debug, and click Analyze.
DFTVisualizer displays and highlights overlapping design, simulation, and test pattern
data for the selected simulation mismatch.

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Example
The following example shows a simulation script for ModelSim where the design netlist and
model library have been pre-compiled into the my_work directory. The script will be used to
compile and simulate the test bench that was saved using the write_patterns command as well as
the debug test bench created by the tool. Prior to execution of the script, the ${1} variable will
be replaced by the tool with the name of the test bench being compiled and ${2} will be
replaced by the test bench top level being simulated.

#! /bin/csh -f
vlog ${1} -work my_work
vsim ${2} -c -lib my_work -do "run -all" -sdfmax \
/${2}/design_inst=~/design/good.sdf

If there are simulation mismatches in the first execution of the saved test bench, the
analyze_simulation_mismatches command creates a new version of the test bench for the
analysis. To create a simulation script that modifies this test bench, such as adding fixed force
statements, you must substitute the variable $ENTITY for the test bench name.

For example: force -freeze sim:$entity/instance/pin0 1

The script must be executable otherwise the following error is returned:

analyze_simulation_mismatches -simulation_script vsim_scr

sh: line 1: ./vsim_scr: Permission denied
// Error: Error when running script: vsim_scr
work_dft_debug/mentor_default.v circle_mentor_default_v_ctl > /dev/null
// No mismatch source is found.

To correct this error, use the following Linux command in the tool before running the
analyze_simulation_mismatches command:

!chmod +x vsim_scr

Analyzing Patterns
Sometimes, you can find additional information that is difficult to access in the Verilog test
benches in other pattern formats. When comparing different pattern formats, it is useful to know
that the pattern numbering is the same in all formats. In other words, pattern #37 in the ASCII
pattern file corresponds to pattern #37 in the WGL or Verilog format.
Each of the pattern formats is described in detail in the section, “Saving Patterns in Basic Test
Data Formats,” beginning on page 372.

Checking for Clock-Skew Problems with Mux-DFF

Designs
If you have mux-DFF scan circuitry in your design, you should be aware of, and thus test for, a
common timing problem involving clock skew.

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Figure 7-54 depicts the possible clock-skew problem with the mux-DFF architecture.

Figure 7-54. Clock-Skew Example

data Combinational Logic

mux
delay setup
MUX MUX
sc_in
DFF DFF

sc_en

clk
clk delay

You can run into problems if the clock delay due to routing, modeled by the buffer, is greater
than the mux delay minus the flip-flop setup time. In this situation, the data does not get
captured correctly from the previous cell in the scan chain and therefore, the scan chain does not
shift data properly.

To detect this problem, you should run both critical timing analysis and functional simulation of
the scan load/unload procedure. You can use ModelSim or another HDL simulator for the
functional simulation, and a static timing analyzer such as SST Velocity for the timing analysis.
Refer to the ModelSim SE/EE User’s Manual or the SST Velocity User’s Manual for details on
performing timing verification.

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 Chapter 8
Multiprocessing for ATPG and Simulation

This chapter explains multiprocessing functionality for accelerating ATPG and simulation.
Multiprocessing is a combination of distributed processing and multithreading. Multiprocessing
functionality allows you to create slave processes with multiple threads to efficiently use
additional processors.

Definition of Multiprocessing Terms

This manual uses the following terminology:
• Distribution or Distributed Processing — The dividing up and simultaneous
execution of processing tasks on multiple machines.
• Multithreading — The dividing up and simultaneous execution of processing tasks
within one process running on one machine. This method minimizes memory use per
thread by sharing the design information across all threads running on the same
machine.
• Multiprocessing — A general term that can refer either to distribution or
multithreading, or to a combination of both.
• Manual Mode — Starting additional processors by explicitly specifying the host(s) on
which the processors run.
• Process — The instance of the executable running on the master or slave host,
regardless of the number of threads in use.
• Processor — A resource that executes a thread, which is added with the add_processors
command or by starting the tool.
• Grid Mode — Starting additional processors using the grid engine. You have limited
control in specifying the host upon which the job is started because that is handled by the
grid engine, subject to any constraints you place on the grid job submission.
• Thread — The smallest independent unit of a process. A process can consist of multiple
threads, all of which share the same allocated memory and other resources on the same
host.
• Worker Thread — Threads other than the main thread.

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Using Multiprocessing to Reduce Runtime

Using Multiprocessing to Reduce Runtime

During ATPG and fault simulation, the tool can optionally use multiprocessing when executing
the following commands:
• compress_patterns (fault simulation)
• create_patterns (ATPG)
• identify_redundant_faults (ATPG)
• order_patterns (fault simulation)
• simulate_patterns (fault simulation)
For large designs, distributing portions of these commands’ processing load using
multiprocessing can reduce their runtime, sometimes significantly. The reduction in runtime
depends on several factors and is not directly proportional to the number of additional threads
doing the processing. The particular design, the tool setups you use, and the type of patterns you
generate, for example, determine the kinds of processes the tool must run and the proportion of
them it can distribute to additional threads. Generally, the ATPG runtime improvement will be
greater for transition patterns than for stuck-at.

Slave processors or threads can be on the machine on which the tool is running or on remote
machines, wherever you can access additional processors on your network. ATPG results
(coverage and patterns) using multiprocessing are the same as without multiprocessing,
regardless of the number of processors.

Multiprocessing Requirements
To enable the tool to establish and maintain communication with multiple processors on
multiple host machines and run slave processing jobs on them, you need to inform the tool of
the network names of available machines (manual specification) or direct the tool to use an
automated job scheduler to select machines for you. The tool supports Load Sharing Function
(LSF), Sun Grid Engine (SGE), or custom job schedulers. The following prerequisites must be
satisfied for whichever you use:
• Manual Specification (does not require a job scheduler)
You can specify hosts manually, without using the SGE, LSF or a custom job scheduler.
The master host must be able to create processes on the slave host via the rsh or ssh
shell command.
o rsh — This requires that the network allow connection via rsh, and that your .rhosts
file allow rsh access from the master host without specifying a password. This is the
default.

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Note
The .rhosts file on host machines must have read permission set for user. Write and
execute permission can optionally be set for user, but must not be set for other and group.

rsh access is not required for the tool to create additional processes on the master
host.
o ssh — This requires that the network allow connection via ssh. To enable use of ssh,
issue a set_multiprocessing_options command within the tool to set the
multiprocessing “remote_shell” variable to ssh. Do this prior to issuing an
add_processors command.
Master and slave machines must be correctly specified in the global DNS name server
for reliable network operation, and you will need to know either the network name or IP
address of each remote machine you plan to use. Consult the System Administrator at
your site for additional information.
• Job Scheduler — You must have available at your site at least one of the following
methods of network job scheduling. Whichever you use, it must allow the master
process (the process started when you invoke the tool) to create slave processes on
different host machines.
o Load Sharing Function (LSF) — To use LSF, ensure your environment supports
use of the LSF scheduler before you invoke the tool. For example, the LSF_BINDIR
environment variable must be set appropriately, in addition to other requirements.
An appropriate setup can often be performed by sourcing a configuration file
supplied with the scheduler installation.
o Sun Grid Engine (SGE) — To use SGE, ensure your environment supports use of
the SGE scheduler before you invoke the tool. For example, the SGE_ROOT
environment variable must be set appropriately, in addition to other requirements.
An appropriate setup can often be performed by sourcing a configuration file
supplied with the scheduler installation.
o Custom Job Scheduling — For the tool to use a custom job scheduler, you need to
inform the tool of the command used at your site to launch the custom job scheduler.
You do this by issuing a set_multiprocessing_options command within the tool to set
the “generic_scheduler” variable to the appropriate site-specific command.
• Job Scheduling Options — You can control certain aspects of the job scheduling
process with the switches you set with the set_multiprocessing_options command. For
complete information about these switches, refer to the set_multiprocessing_options
description in the Tessent Shell Reference Manual.
• Tessent Software Tree Installation — It must be possible to execute the Mentor
Graphics Tessent executables (via fully specified paths) from any new processes.

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Procedures for Multiprocessing

• Tool Versions — All multiprocessing hosts must run the same version of the tool. This
is not an issue when using a single executable for all slave hosts, but depending on the
installation, may become an issue when the path to the executable points to a different
physical disk location on a slave host than on the master host. The tool installation tree
must be accessible via the same path on all slave hosts.

Procedures for Multiprocessing

The following describes how to use multiprocessing functionality:
• Using Multiprocessing Overview
• Disabling Multithreading Functionality
• Troubleshooting SSH Environment and Passphrase Errors
• Adding Threads to the Master
• Adding Processors in Manual Mode
• Adding Processors in Grid Mode
• Adding Processors to the LSF Grid
• Deleting Processors Added in Manual Mode
• Deleting Processors Added in Grid Mode

Using Multiprocessing Overview

The following describes the basic procedure for using multiprocessing for ATPG and fault
simulation.
These steps assume you have satisfied the “Multiprocessing Requirements” on page 322.

Procedure
1. Use the set_multiprocessing_options command to specify the values of any
multiprocessing variables, such as the job scheduler you will be using or the remote
shell. Manual specification requires this step only if you want the tool to use ssh.
Specific LSF and SGE options will depend on how SGE and LSF are configured at your
site. It is a good idea to consult your System Administrator, as you may not need any
options.
2. Use the add_processors command to define host machines and the number of processors
that the tool can use for slave processes. For example:
SETUP> add_processors lsf:4 machineB:2

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specifies for the tool to distribute the processing load among any four available LSF
processors and two processors on the machine named “machineB.” Note that with the
default settings there would be one slave with four threads from LSF, and one slave with
two threads from machineB. And without multithreading enabled, there would be four
separate slaves from LSF, and two from machineB.

Note
If there are no slave processes or additional threads running, the session consumes just
one license. However, when you initiate additional processors using the add_processors
command, the tool acquires additional licenses for these processors, with each additional
license allowing up to four additional processors. The previous example adds six
processors, so two additional licenses would be used.

3. Perform any other tool setups you need for ATPG or fault simulation, then issue the
command to be multiprocessed. The tool displays a message indicating the design is
being sent to slave processors and then let you know when the slave processors start
participating.
The following is a list of related commands:

• add_processors — Runs multiple processors in parallel on multiple machines to reduce

ATPG or fault simulation runtime.
• delete_processors — Removes processors previously defined using the add_processors
command.
• get_multiprocessing_option — Returns the value of a single specified variable
previously set with the set_multiprocessing_options command. This is an introspection
command that returns a Tcl result.
• report_multiprocessing_options — Displays the values of all variables used when
executing multiprocessing commands for ATPG or fault simulation.
• report_processors — Displays information about the processors used for the current
multiprocessing environment.
• set_multiprocessing_options — Sets the values of one or more multiprocessing
variables.

Troubleshooting SSH Environment and Passphrase

Errors
Use the information in this section to resolve problems that can occur if you attempt to use the
SSH protocol when your environment has not set up the secure shell agent and password.

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Error: Cannot change from rsh to ssh in the same tool session
SETUP> report_processors
Error: Cannot change from rsh to ssh in the same tool session.

This error message indicates the set_multiprocessing_options command was not used to set the
remote_shell variable to ssh prior to use of the report_processors or add_processors command.

Error: Not running a secure shell agent (SSH_AUTH_SOCK does not exist,
use ssh-agent)
SETUP> set_multiprocessing_options -remote_shell ssh
Permission denied (publickey,password,keyboard-interactive).
// Error: Not running a secure shell agent ( SSH_AUTH_SOCK does not
// exist, use ssh-agent ).

The SSH functionality checks for the presence of two environment variables,
SSH_AGENT_PID and SSH_AUTH_SOCK, that describe your secure shell session. If not
present, it indicates the required ssh-agent daemon is not running.

To fix this problem, suspend the tool session with Control-Z and run the ssh-agent shell
program. This will start the agent and echo to the screen the required settings for the
environment variables. For example:

ssh-agent
setenv SSH_AUTH_SOCK /tmp/ssh-yXm13171/agent.13171;
setenv SSH_AGENT_PID 13172;

Note
Ensure you remove the trailing semicolons (;) if you copy and paste the ssh-agent output
from the shell environment when you resume the tool session.

You can then resume the suspended session and set the environment variables:

SETUP> setenv SSH_AUTH_SOCK /tmp/ssh-yXm13171/agent.13171

SETUP> setenv SSH_AGENT_PID 13172

Then attempt to set remote_shell again.

Error: ssh-agent requires passphrase (use ssh-add)

SETUP> set_multiprocessing_options -remote_shell ssh
Permission denied (publickey,password,keyboard-interactive).
// Error: ssh-agent requires passphrase (use ssh-add).
// Note: SSH protocol not engaged.

This error message indicates the SSH agent is running but has not been told the passphrase to
allow SSH operations.

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To fix this problem, suspend the tool session with Control-Z and run the ssh-add shell program:

SETUP> ^Z
Stopped (user)
% ssh-add
Could not open a connection to your authentication agent.
% setenv SSH_AUTH_SOCK /tmp/ssh-yXm13171/agent.13171
% setenv SSH_AGENT_PID 13172
% ssh-add
Enter passphrase for /user/.ssh/id_dsa:

Enter a passphrase:

Enter passphrase for /user/.ssh/id_dsa: xxxxxxxxxx

Identity added: /user/.ssh/id_dsa (/user/royb/.ssh/id_dsa)
%

Then resume the session and attempt to set remote_shell again:

SETUP> set_multiprocessing_options -remote_shell ssh

// Note: SSH protocol is engaged.

Disabling Multithreading Functionality

Multithreading functionality is enabled by default. However, if the tool reads a flat model
produced by the pre-v2013.2 simulation kernel, the tool is able to use multithreading
functionality only for ATPG and not simulation. In this case, the tool uses only one thread per
host to simulate the patterns generated.
You can disable multithreading functionality with the following command:

> set_multiprocessing_options -multithreading off

Adding Threads to the Master

If you add processors to the master host with multithreading enabled, the master process
increases its thread count instead of adding additional slaves. The command transcript lists how
many threads were started.
> add_processors localhost:4 ; # adds 4 threads to the master process for a total thread count of 5
// Adding 4 threads to birdeye (master)
// Master with 5 threads running.

> report_processors
// hosts threads arch CPU(s) %idle free RAM process size
// ---------------- ------- ------ ----------- ----- --------- ------------
// birdeye (master) 5 x86-64 8 x 2.9 GHz 89% 113.91 MB 232.56 MB
// master with 5 threads running.

Note that you can specify the master host by name or IP address.

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Adding Processors in Manual Mode

With multithreading enabled, the add_processors command can start only one process per slave
host, and create additional processors (to master or slave) as additional threads.
The following example starts a distributed slave process on the host “odin” that runs with 4 total
threads.

> add_processors odin:4

// Adding 4 threads to odin (new slave)
// Master with 1 thread and 1 slave with 4 threads running (5 total threads).

In the following example, the first command starts a distributed slave process on the host “odin”
that runs with 2 threads. The second command increases the number of threads in the slave
process from 2 to 4 rather than starting a new slave with 2 threads.

# odin is not the master host

> add_processors odin:2
// Adding 2 threads to odin (new slave)
// Master with 1 thread and 1 slave with 2 threads running (3 total threads).

> add_processors odin:2

// Adding 2 threads to odin (existing slave now using 4 threads)
// Master with 1 thread and 1 slave with 4 threads running (5 total threads).

Note that if you don’t specify a number after the host name, the default is 1.

If you add more threads to a host than it has CPU cores, the command adds the threads but
issues a warning that the maximum number of available CPUs has been exceeded. If you
specify the string “maxcpu” instead of a number, the tool fills up the process with threads until
the number of threads equals the number of processors available to the host. If the host already
uses the maxcpu number of threads, the tool issues a warning that no more processors were
added. (Note that this is true for the master host as well as slave hosts.)

For example:

# Assume odin has 8 processors and is not the master host

> add_processors odin:2
// Adding 2 threads to odin (new slave)
// Master with 1 thread and 1 slave with 2 threads running (3 total threads).

> add_processors odin:maxcpu

// Adding 6 threads to odin (existing slave now using 8 threads)
// Master with 1 thread and 1 slave with 8 threads running (9 total threads).

The “add_processors maxcpu” command always adds enough threads to reach a total of
“maxcpu” threads for a host (that is, the second add_processors command in the example above
would always add enough threads to total 8 on the host).

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The following is a more detailed example of adding processors in manual mode:

# 3 different hosts used: thor, odin, and localhost,

# each with 16 CPUs

# First show the options used (defaults)

> report_multiprocessing_options
// Multiprocessing options:
// Option Type Value Description
// --------------------------- ------ ----- -------------------------------------
// generic_delete string generic job scheduler delete
// generic_scheduler string generic job scheduler
// license_timeout number 5 # mins to acquire license ( 0: infinite )
// lsf_options string options for LSF job scheduler
// multithreading on/off on turn on/off multithreading flow
// processors_per_grid_request number -1 # processors grouped for one grid request
// (default: 1 for SGE and GENERIC, 4 for LSF requests)
// remote_shell string rsh rsh or ssh remote_shell setting
// result_time_limit float 45 time limit (min) used to detect
// non-responsive slaves.
// scheduler_timeout number 10 # mins for job scheduler
// sge_options string options for SGE job scheduler

# add a slave process with 2 threads on odin

> add_processors odin:2
// Adding 2 threads to odin (new slave)
// Master with 1 thread and 1 slave with 2 threads running (3 total threads).

# add a slave process with 1 thread on thor

> add_processors thor
// Adding 1 thread to thor (new slave)
// Master with 1 thread and 2 slaves with 3 threads running (4 total threads).

# add 4 additional threads to the master

> add_processors localhost:4
// Adding 4 threads to masterhost (master)
// Master with 5 threads running.

> add_processors odin:maxcpu thor:maxcpu

// Adding 6 threads to odin (existing slave now using 8 threads)
// Adding 7 threads to thor (existing slave now using 8 threads)
// Master with 5 threads and 2 slaves with 16 threads running
// (21 total threads).

# do it again just to get a warning that no more threads are added

> add_processors odin:maxcpu thor:maxcpu
// Warning: Max number of processors on odin already in use (8 exist).
// No processors added.
// Warning: Max number of processors on thor already in use (8 exist).
// No processors added.

> report_processors
// hosts threads arch CPU(s) %idle free RAM process size
// -------------------------- ------- ------ ----------- ----- ----------- ------------
// masterhost (master) 5 x86-64 1 x 2.6 GHz 99% 59.28 MB 256.79 MB
// odin 8 x86-64 8 x 2.9 GHz 86% 6525.36 MB 154.53 MB
// thor 8 x86-64 8 x 2.8 GHz 100% 58782.55 MB 153.59 MB
// master with 5 threads and 2 slaves with 16 threads running.

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Adding Processors in Grid Mode

As in manual mode, the tool doesn't start a new slave process on a host given by the grid engine
if there is already a running slave on that machine. Instead, the thread count of the existing slave
(or master, if the grid request returns the master host) is increased.
Requesting one processor at a time from the grid engine is inefficient in many cases, as the tool
has to request enough memory for a full slave process with every grid request. Therefore, the
tool groups a number of processors together into one grid request for a host with enough
processors and the memory needed by one slave with the specified number of threads.

So, if the processors_per_grid_request variable is set to 4, then an “add_processors sge:8”

command results in two grid requests for 4 processors (slots) each.

The threads started are associated with the requested grid resource, and the resource is freed
back to the grid only if all threads are removed.

The following is an example of adding processors in grid mode:

# group 4 processors per grid request

> set_multiprocessing_options -processors_per_grid_request 4
…
# add 20 processors via LSF in 5 groups of 4 processors each
> add_processors lsf:20
# this may result in a situation like this:
> report_processors
// hosts threads arch CPU(s) %idle free RAM process size
// ----------------------- ------- ------ ------------ ----- ----------- ------------
// masterhost (master) 5 x86-64 16 x 2.8 GHz 100% 4655.45 MB 189.70 MB
// kraken 4 x86-64 8 x 2.8 GHz 100% 49828.50 MB 151.86 MB
// brighty 8 x86-64 8 x 2.9 GHz 100% 9461.83 MB 152.66 MB
// joker111 4 x86-64 16 x 2.8 GHz 100% 41892.00 MB 166.05 MB
// master with 5 threads and 3 slaves with 16 threads running.

Note that for LSF, the processors_per_grid is set to 4, and the requested processors_per_grid is
automatically provided to the LSF scheduler. However, for SGE or the generic scheduler, the
default is set to 1 because bundling of slot requests is site-specific. You have to configure the
options specific to that grid system before using this option on non-LSF grids. For more
information about how to do this, refer to the set_multiprocessing_options description in the
Tessent Shell Reference Manual.

Note
Take care to exclude any unsupported platforms that exist in your grid environment by
specifying appropriate grid job submission constraints. For a list of supported platforms,
refer to “Supported Hardware and Operating Systems” in Managing Mentor Graphics
Tessent Software.

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Adding Processors to the LSF Grid

In addition to the general features described in the previous section, the tool provides some
extra assistance for submitting jobs to the LSF grid.
Whereas SGE has predefined machine types and architectures for which the tool can constrain
grid requests to hosts that are supported by the tool, LSF requires individual grid installations to
classify hosts using site-specific “model” and “type” designations. If you specify “type”
constraints in the lsf_options variable (with “set_multiprocessing_options -lsf_options”), then
the tool uses those constraints. Otherwise, the tool attempts to determine which type/model
combinations it can use as described below.

By default, the tool uses both LSF scheduler learning and LSF heuristics learning to determine
suitable type/model combinations when submitting jobs to the grid. You can enable and disable
these two features using the set_multiprocessing_options switches: -lsf_learning and
-lsf_heuristics. When adding processors to the LSF grid, the sequence of events is as follows:

1. When LSF scheduler learning is enabled, the tool uses the LSF system to determine
what machines and type/model combinations are available. It then runs the lsrun
command to log onto likely machines to determine which are compatible with the tool.
LSF scheduler learning cannot succeed if your LSF system is configured to prevent use
of lsrun.

Tip: If your system does not support lsrun, you may want to disable LSF scheduler
learning to avoid the resulting overhead and warning messages.

2. If LSF scheduler learning is disabled or fails, the tool then uses LSF heuristics learning
unless you have disabled this feature with “-lsf_heuristics off.” LSF heuristics learning
is an attempt to automatically choose machines on the LSF grid that are compatible with
the tool. It attempts to determine tool compatibility based on the names of the types and
models available and successfully identify some type/model combinations that the tool
can use; however, this may also result in specifying incompatible combinations or miss
some compatible combinations.

Tip: If you are sure that your slave requests will always result in the tool obtaining
compatible hosts (for example, because you specified an LSF host queue with
-lsf_options that contains only compatible hosts), then you may want to disable LSF
heuristics learning to avoid learning potentially incorrect or incomplete results.

3. If LSF heuristics learning is disabled or fails, then the tool submits the job anyway
without specifying any type/model restrictions.

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Deleting Processors Added in Manual Mode

The delete_processors command decreases the number of running threads or stops an entire
slave process if the number of threads equals zero (or less) after deducting the number of
processors to delete from the number of running threads.
If you don’t specify a number, the command removes all threads and the slave process on the
related host. If you don’t specify a number or a host, the command removes all slaves and their
threads, as well as all additional threads on the master.

If you issue a delete_processors command for the master, the tool has to make sure that the
master is kept running with at least one thread. If a thread is deleted from its process, the process
immediately frees all the related thread data.

Deleting Processors Added in Grid Mode

The delete_processors command releases grid slots only if all threads belonging to a grid
request were deleted. For example:
# (assume that sge_options have already been configured properly.)
> set_multiprocessing_options -processors_per_grid_request 4
# add 20 processors to SGE
> add_processors sge:20
# this may end up in a situation like this:
# each slave has at least 4 threads
> report_processors
# hosts threads arch CPU(s) %idle free RAM process size
# ----------------------- ------- ------ ------------ ----- ----------- ------------
# localhost (master) 5 x86-64 16 x 2.8 GHz 100% 4655.45 MB 189.70 MB
# kraken 4 x86-64 8 x 2.8 GHz 100% 49828.50 MB 151.86 MB
# brighty 8 x86-64 8 x 2.9 GHz 100% 9461.83 MB 152.66 MB
# joker111 4 x86-64 16 x 2.8 GHz 100% 41892.00 MB 166.05 MB
# master with 5 threads and 3 slaves with 16 threads running.

# delete 4 threads from brighty, which frees exactly one grid slot
# (since 4 processors were requested with each grid request)
> delete_processors brighty:4
# delete 2 threads on kraken, which does not free any grid resource
> delete_processors kraken:2
# delete everything from kraken, all grid resources (1 request) are freed
> delete_processors kraken
# What is left?
> report_processors
# hosts threads arch CPU(s) %idle free RAM process size
# ----------------------- ------- ------ ------------ ----- ----------- ------------
# localhost (master) 5 x86-64 16 x 2.8 GHz 100% 4655.45 MB 189.70 MB
# brighty 4 x86-64 8 x 2.9 GHz 100% 9461.83 MB 152.66 MB
# joker111 4 x86-64 16 x 2.8 GHz 100% 41892.00 MB 166.05 MB
# master with 5 threads and 2 slaves with 8 threads running.

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 Chapter 9
Scan Pattern Retargeting

This chapter describes scan pattern retargeting functionality in the Tessent Shell tool.
Scan pattern retargeting improves efficiency and productivity by enabling you to generate core-
level test patterns and retarget them for reuse at the top level. Patterns for multiple cores can be
merged and applied simultaneously at the chip level. This capability can be used for cores that
include any configuration the tool supports for ATPG; this includes multiple EDT blocks and/or
uncompressed chains, pipeline stages, low power, and cores with varying shift lengths.

The scan pattern retargeting capability enables you to:

• Design and develop cores in isolation from the rest of the design.
• Generate and verify test patterns of the cores at the core level.
• Generate test patterns for identical cores just once.
• Divide and conquer a large design by testing groups of cores.
• Have pipeline stages and/or inversion between the core boundary and chip level.
• Broadcast stimuli to multiple instances of the same core.
• Merge patterns generated for multiple cores and apply them simultaneously.
• Automatically trace core-level scan pins to the chip level to extract connections,
pipelining, and inversion outside the cores.
• Perform reverse mapping of silicon failures from the chip level to the core level to allow
diagnosis to be performed at the core level.

Caution
You should verify the chip-level patterns through simulation since the tool's DRCs may
not detect every setup error or timing issue.

If you would like to use Tessent Diagnosis to diagnose your retargeted patterns, refer to
“Reverse Mapping Top-Level Failures to the Core” in the Tessent Diagnosis User’s Manual.

Tools and Licensing

Scan pattern retargeting requires a TestKompress license. Reverse mapping of chip-level
failures, as well as diagnosis of those failures, requires a Tessent Diagnosis license.

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Scan Pattern Retargeting Process Overview

Scan pattern retargeting and the generation of core-level retargetable patterns must be done
within the Tessent Shell tool. Instructions for invoking Tessent Shell are provided in the
following section.

For complete information about using Tessent Shell, refer to the Tessent Shell User’s Manual.

Scan Pattern Retargeting Process Overview

This section provides an overview of the scan retargeting process.
An example of the scan pattern retargeting process is presented in “Retargeting Example” on
page 354.

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Scan Pattern Retargeting Process

The scan pattern retargeting process is as follows:
1. Generate patterns for all cores. This process is illustrated in Figure 9-1. For more
information, see “Generating Patterns at the Core Level” on page 348.”

Figure 9-1. Core-level Pattern Generation Process

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2. Retarget core-level test patterns to the chip level. This process is illustrated in
Figure 9-2. For more information, see “Retargeting Patterns in Internal Mode” on
page 349.”

Figure 9-2. Scan Pattern Retargeting - Internal Mode

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3. Generate patterns for the top-level chip logic only. This process is illustrated in
Figure 9-3. For more information, see “Generating Patterns in External Mode” on
page 353.

Figure 9-3. Pattern Generation - External Mode

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Core-Level Pattern Generation

Core-Level Pattern Generation

In order to generate retargetable patterns for a core, the core must already have wrapper chains
inserted into it and it must be configured to its internal test mode. Otherwise, you can lose
significant coverage.
To enable core-level generation of retargetable patterns, you must execute “set_current_mode
[mode_name] -type internal” in setup mode.

When later retargeting core-level patterns to the top level, the tool maps only scan data from the
core level to the top level. The tool does not map core-level PIs and POs to the top level. Nor
does the tool map core-level capture cycle events from the core level to the top level. Therefore,
primary inputs must be constrained during capture, outputs must be masked, and clocking must
meet certain requirements, as explained next.

During core-level pattern generation, the tool automatically adds X constraints on most input
ports and adds output masks on all output ports so that generated patterns can be mapped to the
chip level for retargeting. Pulse-always clocks, pulse-in-capture clocks, and constrained pins are
not forced to X. The tool also omits internal primary inputs (added with add_clocks or
“add_primary_inputs -internal”) since those cut points are for modeling and are usually
controlled by Named Capture Procedures (NCPs) or clock control definitions.

You can exclude additional primary inputs from X constraints; this is necessary if the pins are
controlled by an NCP that models chip-level constraints such as a clock controller located
outside of the core. All inputs must be constrained or controlled (e.g., using all enabled NCPs).

Note
Even scan_enable, like any other input, must either be constrained to 0 or forced using an
NCP.

If a pin is not a pulse-always or pulse-in-capture clock, is not constrained, and is excluded from
isolation constraints (that is, the tool did not constrain it to X), the create_patterns command
checks that the pin is controlled. If the pin is not controlled, the tool issues an R7 DRC
violation.

Be careful when you apply the is_excluded_from_isolation_constraints attribute to a port, and

never ignore an R7 DRC violation. Doing this can result in ATPG controlling the unconstrained
port to any arbitrary value which can vary from pattern to pattern, even though pattern
retargeting only retargets the scan data and does NOT retarget values on primary inputs from
the core level to the top.

Core-level test patterns are saved in ASCII format when using the write_patterns command.
Only patterns saved in ASCII format can be retargeted. You can also use the write_patterns
command, as in the normal ATPG flow, to generate simulation test benches or write the patterns
in other formats.

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Core-Level Pattern Generation

You can use existing commands to write out core-level fault list and fault detection information.

Caution
When generating the EDT IP, if output channel pipeline stages will be added later, you
must specify “set_edt_pins -change_edge_at_compactor_output trailing_edge” to ensure
that the compactor output changes consistently on the trailing edge of the EDT clock.
Output channel pipeline stages should then start with leading-edge sequential elements.

Clocking Architecture
The scan pattern retargeting functionality imposes the following restrictions on clock
architecture:

• The tool does not perform per-pattern capture cycle mapping. The capture clock
conditions at the core boundary must be identical for every pattern in the pattern set.
• The recommended methodology is to have a programmable clock chopper inside each
core that is programmed by scan cell values. The scan cells used to program the clock
chopper must be inside the core. This provides ATPG with the flexibility to generate the
required clocking sequences, yet ensures that the clock conditions at the boundary of the
cores are the same (pulse-always clocks, or clocks that are pulsed only during setup
and/or shift but constrained off during capture).
• If the clock controller is outside the core, it must have been programmed statically
during test_setup to deliver the same clock sequence during the capture phase of every
pattern. During core-level ATPG, the clocking at the core boundary must be enforced by
defining an NCP. In addition, any unconstrained pins on the core boundary must have
the is_excluded_from_isolation_constraints attribute set; otherwise, the tool will
constrain them to X.
• No support is provided if the clocking on the boundary of the core is different for each
pattern, such as when the capture clocks are programmed using a scan chain outside the
core, or when the capture clocks are driven directly by chip-level pins.

The Tessent Core Description File

The pattern generation step produces a Tessent core description (TCD) file. The TCD file
contains the information the tool needs to map the core-level patterns to the next level of
hierarchy. This includes information such as description of the EDT hardware and scan chains.

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Scan Pattern Retargeting of Test Patterns

Scan Pattern Retargeting of Test Patterns

To retarget core-level test patterns, Tessent Shell maps the scan path pins from the cores to the
top level, adjusts the core-level patterns based on the additional retargeting pipeline stages
learned between the core and chip level (to ensure that all patterns from different cores have the
same shift length via padding), and maps them to the chip-level pins they connect to.
Note that retargeting of core-level patterns does not require you to provide a full netlist for
every core. Alternatively, you can specify a core as either a graybox or blackbox in the
following ways:

• If you have generated a graybox and want to handle the core as a graybox model, use the
read_verilog command to read the Verilog model that was written out from the graybox
generation step. Although the wrapper scan chains in the graybox model are not needed
for the retargeting mode, feedthroughs are preserved in the graybox model. If
feedthroughs are used for connecting cores to the chip level and blackboxing the cores
would break that path from the core boundary to the top, it is recommended to use the
graybox model.
• If you have the full core netlist in a separate file from the top-level module(s) and just
want to read it in and preserve the boundary but discard the contents, use the
read_verilog <design> -blackbox command. This method can only be used if there are
no feedthroughs or control logic within the core that are needed for the current mode of
operation.
• If you want to read the full design and blackbox specific core instances during design
flattening, use the add_black_box command. This method can only be used if there are
no feedthroughs or control logic within the core that are needed for the current mode of
operation.
For more information, see “Scan Pattern Retargeting Without a Netlist” on page 343.

Chip-Level Test Procedures

You must provide the chip-level test_setup procedure needed for the retargeting step. You must
also either provide the chip-level load_unload and shift procedures, or allow the tool to create
them automatically by mapping and merging the core-level load_unload and shift procedures as
described in “Test Procedure Retargeting For Scan Pattern Retargeting” on page 341.

External_capture Procedure
You specify the capture cycle in an external_capture procedure, using the
set_external_capture_options -capture_procedure switch. The -capture_procedure switch
specifies the external_capture procedure to be used. The tool uses this external capture
procedure for all capture cycles between each scan load, even when the pattern is a multi-load
pattern. An example external_capture procedure is shown here:

procedure external_capture ext_procedure_1 =

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timeplate tmp_1;
cycle =
force_pi;
pulse clk_trigger 1;
pulse reference_clock;
end;
cycle =
pulse reference_clock;
end;
end;

The external_capture procedure has the same syntax as an NCP, except it does not have internal
and external modes. In comparison to an NCP, the external_capture procedure has the following
restrictions:

• Can only have one force_pi statement.

• Cannot have any measure_po statements.
• Cannot have any load_cycles as it is used between each scan load of a pattern.
• Cannot contain any events on internal signals.
Pin constraints defined at the top level are used in this generic capture sequence. No pattern-
specific capture information is mapped from the patterns.

Test Procedure Retargeting For Scan Pattern

Retargeting
The automatic mapping (transfer) of test procedure information from the core level to the top
level of the design occurs when the tool transitions to analysis mode.
load_unload and shift Procedures
When core-level patterns are retargeted to the chip-level, chip-level load_unload and shift test
procedures are needed. If chip-level load_unload and shift test procedures do not exist, the tool
automatically generates them based on the information in the core-level TCD files. This
retargeting step merges all of the core test procedures (if there is more than one core), and maps
them to the top-level test procedures.

Clocks and Pin Constraints

By default, the tool maps all clocks and pin constraints that are specified at the core-level to the
top. Clock types are mapped based on the core-level specification. Note, the tool maps any
permanently tied constraints (CT) in those files as constraints that can be overridden (C).

Even though clocks and pin constraints are automatically mapped to the top level by default,
you can still define them at the top, or drive the core-level input constraints through logic such
as a TAP.

Any clocks that are mapped to the top are removed when the tool exits analysis mode.

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test_setup and test_end Procedures

You will typically need to provide top-level test_setup and test_end procedures to configure
instruments such as PLLs, OCCs, EDT IPs, channel multiplexer access, and so on in preparation
for scan test. These instruments may be at the top level and/or within the cores. The test_setup
and test_end procedures can consist of a cyclized sequence of events, and/or IJTAG iCall
statements.

If any of the instruments used for scan test are inside a core, a test_setup and/or test_end
procedure at the core level must have been used to initialize them. If IJTAG was used within a
core-level test_setup or test_end procedure, this information is retained in the TCD file. When
an instance of this core is added for mapping, the iCalls that were used at the core level are
automatically mapped to the top level and added to the top-level test_setup or test_end
procedure. In other words, if you use IJTAG to set up the core at the core level, the tool
automatically maps those sequences to the top-level test_setup and test_end procedures without
you doing anything. Of course, the top-level test_setup and test_end procedures may have
additional events to configure top-level instruments.

In order for the tool to perform automatic IJTAG mapping, you must have extracted the chip-
level ICL using the extract_icl command. You must also have sourced the core-level PDL. This
step-by-step procedure is described in the “Step-by-Step ICL Extraction Procedure” section in
the Tessent IJTAG User’s Manual.

Several R DRCs validate the presence of both ICL and PDL. Note that the tool does not store
ICL and PDL in the core-level TCD file; it only stores references to their usage in the test_setup
procedure.

Note
You can disable load_unload and shift procedure retargeting by executing the
“set_procedure_retargeting_options -scan off” command. If test procedure retargeting is
disabled and chip-level load_unload and shift test procedures are missing, the tool
generates an error.

You can disable the automated mapping of core-level iCalls stored in the TCD file to the
top level by executing the “set_procedure_retargeting_options -ijtag off” command. (This
will not have any impact on iCalls you explicitly added into the top-level setup
procedures either explicitly or indirectly using the set_test_setup_icall command.) If
IJTAG retargeting is disabled, you must provide the needed test_setup and test_end
procedures.

Constraint of the Top-Level Port During the Load_Unload Test Procedure

You can set a constant value for a top-level port during the simulation of the load_unload test
procedure, when needed, to enable tracing of signals from the core level to the chip level. You
can do this by setting the value of the constraint_value_during_load_unload attribute using the
set_attribute_value command.

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If, for example, scan_enable must be 1 to trace signals needed for load_unload and shift
procedure retargeting, you would need to define this load_unload constraint in the tool for test
procedure retargeting to work; especially, because this value will likely conflict with the 0 input
constraint typically added to scan_enable for capture.

For more information see the set_attribute_value command and the Port data model attributes in
the Tessent Shell Reference Manual.

Scan Pattern Retargeting Without a Netlist

Before describing the process of scan pattern retargeting without a netlist, it is important to
clarify that the retargeting flow consists of two distinct steps.
These two steps are:

• Extraction — This step is the process of determining the connectivity from a lower-level
wrapped core to the current top level of hierarchy. It validates that the core is embedded
properly with respect to the signals and constrained values necessary for retargeting (e.g.
scan chains/channels, clocks, and constrained inputs). This step is also where we
perform optional mapping of the test procedures. Once the extraction step is complete, a
TCD file can be written out that captures the information needed for retargeting
core-level patterns to the current design level.
• Retargeting — This is the step in which the core-level patterns that have been read in are
then mapped to the top-level pins, and multiple pattern sets are merged together for
simultaneous application.
These steps are discrete and you can perform them separately in different sessions or together.

Figure 9-2 shows retargeting as a single pass procedure but you can separate these two steps out
of the process that is illustrated. The extraction step occurs when you execute the
“set_system_mode analysis” command and perform DRC. All connectivity between the core
and the top is extracted and validated. After extraction, you use the write_core_description
command to write out the top-level TCD file which includes the extracted connectivity and
top-level scan procedures. This top-level TCD file includes all of the core-level TCD
information in this mode so that it is self-contained. You can stop at this point, and not read and
write the core-level retargetable patterns, and you will have completed the extraction step only.
At this point, you can write out the top-level TCD file. If, instead, you continue and read and
write the core-level retargetable patterns before writing out the top-level TCD file, you will
have completed the pattern retargeting step.

When performing the extraction step, you are not required to provide a full netlist for every
core; instead, you can specify a core as either a graybox or blackbox. The tool retrieves all of the
information necessary for retargeting its patterns from the core-level TCD file. The only reasons
that you would not want to blackbox a core are:

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• If you have a core-level test_setup procedure. The R5 DRC (test_setup validation)

cannot be performed if the setup logic in the core is not present during extraction DRC.
• If the logic in the core is involved in the top-level procedure simulation and access from
the top to the core. For example, if the core includes feed-throughs of currently active
cores, or includes the TAP for the design which is needed for simulating the test_setup
procedure.
You are able to perform pattern retargeting without performing extraction again because you
can utilize the previously generated information in the top-level TCD file. The top-level TCD
file includes all of the information needed to perform retargeting including the core-level
descriptions, connectivity of the scan pins, and top-level scan procedures used and validated
during extraction. You only need to read in the top-level TCD file and the core-level patterns to
perform pattern retargeting. This enables efficient retargeting of new core-level patterns if the
design and procedures have not changed since DRC and core connectivity extraction were
performed; if this is not the case, you will need to rerun extraction again.

If anything changes with the setup of the core, such as procedures or constraints, you must
re-run extraction and DRC because the previously extracted top-level TCD may no longer be
valid.

Retargeting Support For Patterns Generated at the Top

Level
Scan patterns are typically retargeted with respect to lower-level cores instantiated in the
design; in this case, the core is a lower-level core. However, in some cases, you can test part of
the top-level logic in parallel with lower-level cores by treating this partial view of the top level
as a core instance; in this case, the core is actually the top level.
During pattern retargeting, you use the add_core_instances -current_design command and
option to add the top-level logic as a core instance. This option allows you to add the core
instance with respect to the current design; this core instance then becomes the top level. The
-current_design option also allows patterns for the current design to be merged with patterns of
other core instances in the hierarchy as shown in “Example 2” on page 346.

A core can have multiple modes. A mode of a core is a specific configuration or view of that
design with a specific set of procedures and scan definitions. Two modes of a design can be
mutually exclusive (such as the internal and external test of a wrapper core), or independent of
each other (such as when we generate patterns for non-overlapping parts of a core and then
merge those patterns as in Example 1).

Example 1
The following example demonstrates the use of the -current_design switch to merge two modes
of a core; in this case, the “core” is the top level. In the example, patterns are generated
separately for two sub-blocks of a design from the top-level ports. Sub-Block1 is tested as mode
M1 of the top level, and Sub-Block2 is tested as mode M2 of the top level. The result, shown in

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Figure 9-4, is the merging of the two pattern sets originally generated for each individual mode
into a single pattern set applied at the top level.

The first dofile reads in the top-level design and writes out the patterns and TCD files for mode
M1 targeting Sub-Block1:

// Invoke tool to generate patterns for each mode of top_level

set_context pattern -scan
read_verilog top_level.v // Read netlist of top_level

// Mode M1 consists of the Sub-Block1 design; the user applies constraints to access Sub-Block1
// and can optionally blackbox Sub-Block2 to reduce memory consumption
set_current_mode M1 -type internal
...
set_system_mode analysis
...
create_patterns
write_patterns top_levelM1.ascii
write_core_description top_levelM1.tcd

The second dofile reads in the design and writes out the patterns and TCD files for mode M2
targeting Sub-Block2:

// Mode M2 consists of the Sub-Block2 design; the user applies constraints to access Sub-Block2
and can optionally blackbox Sub-Block1 to reduce memory consumption
set_current_mode M2 -type internal
...
set_system_mode analysis
...
create_patterns
write_patterns top_levelM2.ascii
write_core_description top_levelM2.tcd

The third dofile merges the two separately generated top-level pattern sets (modes M1 and M2)
into a single pattern set:

// Invoke the tool again to retarget patterns to top level

set_context patterns –scan_retargeting
...
read_core_descriptions top_levelM1.tcd
read_core_descriptions top_levelM2.tcd
add_core_instance –core top_level –mode M1 –current_design
add_core_instance –core top_level –mode M2 –current_design
set_system_mode analysis
// Mode extracted from ASCII file
read_patterns top_levelM1.ascii
read_patterns top_levelM2.ascii
write_patterns …

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Figure 9-4. Merging Patterns for Multiple Views of a Core at the Top Level

Example 2
The following example demonstrates the use of the add_core_instances -current_design switch
to merge mode M1 of the current design and mode internal of CoreB. In the example, one
pattern set is generated at the top level (mode M1 of the top level) to test Sub-Block1, and a
second pattern set is generated at the core level of CoreB. The result, as shown in Figure 9-5, is
the integration of the two pattern sets, originally generated for mode M1 of the top level and
mode internal of CoreB, into the top level.

The first dofile reads in the top level design and writes out the patterns and TCD files for mode
M1 targeting Sub-Block1.

// Invoke tool to generate patterns for mode M1 of the top level

set_context pattern -scan
read_verilog top_level.v // Read netlist of top_level

// Mode M1 consists of the Sub-Block1 design; the user applies constraints to access
// Sub-Block1 and can optionally blackbox any other blocks not targeted to reduce memory
// consumption
set_current_mode M1 -type internal
...
set_system_mode analysis
...
create_patterns
write_patterns top_levelM1.ascii
write_core_description top_levelM1.tcd

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The second dofile reads in the CoreB design and writes out the patterns and TCD files.

set_system_mode setup
delete_design
read_verilog CoreB.v
set_current_mode internal -type internal
...
set_system_mode analysis
create_patterns
write_patterns CoreB.ascii
write_core_description CoreB.tcd

The third dofile merges the top level pattern set (mode M1) with the retargeted CoreB patterns.

// Invoke tool to retarget patterns to top level

set_context patterns –scan_retargeting
...
read_core_descriptions top_levelM1.tcd
read_core_descriptions CoreB.tcd
add_core_instance -core top_level -mode M1 -current_design
add_core_instance -core CoreB -mode internal -instances CoreB_i
...
set_system_mode analysis
read_patterns top_levelM1.ascii
read_patterns CoreB.ascii
write_patterns ….

Figure 9-5. Merging Patterns for Top Level and Core at the Top Level

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Generating Patterns at the Core Level

Generating Patterns at the Core Level

This procedure describes how to generate core-level test patterns.

Prerequisites
• Core-level netlist
• Cell library

Procedure
1. Enable ATPG using the “set_context patterns -scan” command.
2. Read the core-level netlist and cell library.
3. Configure the core for internal mode.
4. Generate all pattern types.
5. Address all test coverage issues.
6. Verify the test patterns (test benches, STA).
7. Write the retargetable core test patterns using the write_patterns command.
8. Write the core fault lists using the write_faults command.
9. Write the TCD file using the write_core_description command.
10. Write the flat model for diagnosis using the write_flat_model command.

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Retargeting Patterns in Internal Mode

This procedure describes how to retarget core-level test patterns to the chip level.

Prerequisites
• TCD file for each of the core types.
• Retargetable patterns resulting from core-level pattern generation.
• Top-level netlist (with cores optionally blackboxed or grayboxed).
• Top-level test procedures.

Procedure
1. Enable the retargeting of patterns using the “set_context patterns -scan_retargeting”
command.

Note
A full netlist for cores whose patterns are being retargeted is not required. Retargeting
only requires a graybox or blackbox model of the cores to both retarget the patterns and to
generate chip-level serial and parallel simulation test benches.

2. Read the TCD file for each of the core types.

3. Bind each core description to the core instances it represents in the design using the
add_core_instances command.
4. Read a dofile and test procedure file to configure core access and to configure the cores
whose patterns are being retargeted into their internal mode.
5. Run DRCs when “set_system_mode analysis” is invoked, including extracting scan
connections from the core instances to the top level and validating the embedding of the
cores.
6. Read in the retargetable patterns for each core type using the read_patterns command.
7. Write the retargeted patterns using the write_patterns command. The tool automatically
performs pin mapping (between core and chip-level) and pattern merging as it writes the
chip-level patterns.
8. Write the top-level TCD file. This file includes the core-level core description
information as well as scan connectivity information from the cores to the top. This
information is later used for reverse mapping of silicon failures back to the core level for
diagnosis, or for pattern retargeting without a netlist.

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Note
The top-level TCD file also enables generation of top-level patterns without a netlist. For
more information, see “Retargeting of Patterns at the Chip Level Without a Netlist.”

Chip-level patterns cannot be read back into Tessent Shell since the internal clocking
information is lost. However this is not needed since diagnosis is done at the core-level. The
generated simulation test bench can be simulated for validation. In addition, the design
objective of this functionality is to eliminate the need to load the full design into the tool.

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Extracting Connectivity Between the Top Level and the Core Boundary

Extracting Connectivity Between the Top Level

and the Core Boundary
In the previous “Retargeting Patterns in Internal Mode” section, you perform the extraction and
retargeting steps together.
If you want to separate the extraction and retargeting steps as described in “Scan Pattern
Retargeting Without a Netlist” on page 343, you can use this procedure to extract the
connectivity from the core level to the top level and create the top-level TCD file. You can then
use the procedure in “Retargeting Patterns Without a Top-Level Netlist” on page 352 to
perform retargeting.

Prerequisites
• Core-level netlists. The core can optionally be blackboxed or grayboxed.
• Core-level TCD files
• Top-level netlist

Procedure
1. Enable the retargeting of patterns using the “set_context patterns -scan_retargeting”
command.

Note
A full netlist for cores whose patterns are being retargeted is not required. Retargeting
only requires a graybox or blackbox model of the cores to both retarget the patterns and to
generate chip-level serial and parallel simulation test benches.

2. Read the TCD file for each of the core types.

3. Bind each core description to the core instances it represents in the design using the
add_core_instances command.
4. Read a dofile and test procedure file to configure core access and to configure the cores
whose patterns are being retargeted into their internal mode.
5. Perform extraction using the “set_system_mode analysis” command. This command
runs DRCs, extracts scan connections from the core instances to the top level and
validates the embedding of the cores.
6. Write the top-level TCD file using the “write_core_description command. This file
includes the core-level core description information as well as scan connectivity
information from the cores to the top. This information is later used for reverse mapping
of silicon failures back to the core level for diagnosis, or for pattern retargeting without a
netlist.

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Results
Top-level TCD file.

Related Topics
“Scan Pattern Retargeting Without a Netlist” on page 343

Retargeting Patterns Without a Top-Level

Netlist
You can retarget core-level patterns at the top level without a netlist by using the information
contained in the top-level TCD file.

Prerequisites
• Top-level TCD file. The top-level TCD file results from performing the extraction step
described in “Extracting Connectivity Between the Top Level and the Core Boundary”
on page 351.
• Retargetable patterns resulting from core-level pattern generation.

Procedure
1. Enable the retargeting of patterns using the “set_context patterns -scan_retargeting”
command.
2. Read the top-level TCD file using the read_core_descriptions command. In the absence
of a netlist, the tool recreates what is needed for the design using this file.
3. Change to analysis mode using “set_system_mode analysis”. This command
automatically sets the current design, adds the cores, sets pin constraints, and provides
any other information needed for the design.
4. Read in the retargetable patterns for each core type using the read_patterns command.
5. Report core instances using the report_core_instances command.
6. Retarget and merge patterns as they are written out with respect to the chip-level
boundary using the write_patterns command.

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Generating Patterns in External Mode

This procedure describes how to generate patterns in external mode.

Prerequisites
• Complete top-level netlist or top-level netlist with cores grayboxed.
• Core-level fault lists.
• Top-level dofile.
• Top-level test procedure file.

Procedure
1. Enable the generation of patterns using the “set_context patterns -scan” command.
2. Read a complete top-level netlist, or one where each core is replaced by a light-weight
graybox model, and the cell library.
3. Read the core-level fault lists from each core’s internal mode pattern set. If the netlist is
not complete due to the use of the graybox models, add the -graybox switch to the
“read_faults fault_file -instance instances -merge” command.
4. Read a dofile and test procedure file to configure cores to external mode.
5. Run DRCs when “set_system_mode analysis” is invoked.
6. Create patterns for testing interconnect and top-level glue logic.
7. Write the generated patterns using the write_patterns command.
8. Report the full chip test coverage.

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Retargeting Example
This example demonstrates the retargeting use model.
This example is based on the design shown in Figure 9-6. The design has three cores of two
types: two identical instances of the CPU core and one instance of the NB core. TOP is the chip
level of the design. In the example, the input is broadcast to identical core instances.

Figure 9-6. Retargeting of Core-level Patterns

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Core-level Pattern Generation for CPU Cores

Core-level pattern generation for the CPU core is shown here:
# Set the context for ATPG
set_context patterns -scan

# Read cell library (ATPG library file)

read_cell_library atpglib

# Read the core netlist

read_verilog CPU.v

# Specify to generate patterns that are retargetable for the internal mode
# of the core. Because the mode name was not specified, the default is the
# same as the mode type “internal”.
set_current_mode -type internal

# Specify pins the tool should not constrain to X because they will be
# explicitly controlled by a Named Capture Procedure that models a
# chip-level clock controller driving core-level clocks
set_attribute_value {clk1 clk2} \
-name is_excluded_from_isolation_constraints

# Add commands to set up design for ATPG. To avoid coverage loss due to
# auto isolation constraints on scan_enable signals, constrain scan_enable
# signals to a non-X state
...
add_input_constraint input_scan_enable -C1
add_input_constraint output_scan_enable -C0
add_input_constraint core_scan_enable -C0

# Change to analysis mode

set_system_mode analysis

# Generate TCD file

write_core_description CPU.tcd -replace

# Specify the fault model type and generate patterns

set_fault_type stuck
create_patterns

# Write ASCII patterns for subsequent retargeting; must be ASCII format

write_patterns CPU_stuck.retpat.gz -replace

# Write fault list

write_faults CPU_stuck.faults.gz -replace

# Write flat model for diagnosis

write_flat_model CPU_stuck.flat_model.gz -replace

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Core-level Pattern Generation for NB Core

Core-level pattern generation for the NB core is the same as for the CPU core:
# Set the context for ATPG
set_context patterns -scan

# Read cell library (ATPG library file)

read_cell_library atpglib

# Read core netlist

read_verilog NB.v

# Specify to generate patterns that are retargetable for the internal mode
# of the core. Because the mode name was not specified, the default is the
# same as the mode type “internal”.
set_current_mode -type internal
...

Retargeting of Patterns at the Chip Level

The steps for scan pattern retargeting are shown here. The tool performs design rule checking as
it changes to analysis mode.
# Specify that this is the scan pattern retargeting phase
set_context patterns -scan_retargeting

# Read cell library (ATPG library file)

read_cell_library atpglib

# Read all netlists and the current design. You can replace the core
# netlists with graybox models, or blackbox them to reduce memory usage
# and run time. You can blackbox them using the read_verilog -blackbox,
# or add_black_box commands.
read_verilog TOP.v
read_verilog CPU.v -blackbox
read_verilog NB.v -blackbox
set_current_design TOP

# Specify the top-level test procedure file

set_procfile_name TOP.testproc

# Read all TCD files

read_core_descriptions CPU.tcd
read_core_descriptions NB.tcd

# Bind core descriptions to cores

add_core_instances -core CPU -modules CPU
add_core_instances -core NB -instances /nb_i

# Change to analysis mode

set_system_mode analysis

# Read core (retargetable) patterns

read_patterns CPU_stuck.retpat.gz
read_patterns NB_stuck.retpat.gz

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# Specify generic capture window

set_external_capture_options -capture_procedure \
<external_capture_procedure_name>

# Report core instances, then retarget and merge patterns as they are
# written out with respect to the chip-level boundary
report_core_instances
write_patterns TOP_stuck.stil -stil -replace

# Write top-level core description. Used for reverse mapping of silicon

# failures back to the core level for diagnosis, or for pattern
# retargeting without a netlist
write_core_description TOP_stuck.tcd -replace

Retargeting of Patterns at the Chip Level Without a Netlist

When a logic change is made to a core that requires the core-level patterns to be regenerated, if
the logic change does not affect the ports of the core or the test setup conditions, you can
retarget those patterns without loading the chip-level netlist. Instead, you can use the top-level
TCD file that was written out during the original retargeting run. This allows you to avoid the
runtime required to load in a chip-level netlist and perform the extraction again.
The steps for scan pattern retargeting without a netlist are shown here. Prior to performing the
retargeting in this example, you must have already run the extraction step to generate the
top-level TCD file.

Note
The extraction step is identical to the steps performed in “Retargeting of Patterns at the
Chip Level” on page 356 except that the reading and writing of patterns is omitted.

In the absence of a netlist, the tool recreates what is needed for the design from the top-level
TCD file. The tool performs design rule checking as it changes to analysis mode.

# Specify that this is the scan pattern retargeting phase

set_context patterns -scan_retargeting

# Read the top-level TCD file that resulted from the extraction step
read_core_descriptions TOP_stuck.tcd

# Changing to analysis mode automatically sets the current design, adds

# the cores,sets pin constraints and any other necessary information for
# the design
set_system_mode analysis

# Read core (retargetable) patterns

read_patterns CPU_stuck.retpat.gz
read_patterns NB_stuck.retpat.gz

# Report core instances (created automatically), then retarget and merge

# patterns as they are written out with respect to the chip-level boundary
report_core_instances

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Core-in-Core Pattern Retargeting

write_patterns TOP_stuck.stil -stil -replace

Core-in-Core Pattern Retargeting

Core-in-core pattern retargeting is the process of retargeting patterns through multiple levels of
embedded wrapped cores to the top level of the design. That is, if the top design contains one or
more cores and any one of those cores contains any number of nested cores, you can retarget
patterns from the lowest level to the top level of the design.
The core-in-core pattern retargeting process provides two different options: you can use flat
extraction that retargets patterns from the child to the chip level in one step, or you can perform
extraction in multiple steps (one for each intermediate (typically wrapped) core level).

Flat Extraction Flow — The flat extraction flow requires you to provide the blackbox,
graybox, or full netlist for the core being retargeted and the full netlist for all levels of hierarchy
above it. While the flat extraction flow may require less file management (fewer TCD files to
track) and fewer extraction steps, it does mean loading large netlists that need more memory. It
also is less consistent with a hierarchical methodology in which everything is completed at each
(typically) wrapped level of hierarchy.

Multiple Extraction Step Flow — Instead of the flat extraction flow, you can retarget patterns
by performing multiple extraction steps (one for each intermediate (typically) wrapped level).
For example, consider a design in which there is a low-level wrapped core, a child core,
instantiated inside another wrapped core, a parent core, which is then instantiated in the top
level. You perform extraction from the child to the parent at the parent level, store the
associated TCD file, perform extraction from the parent level to the top, and then perform
pattern retargeting. In addition to requiring multiple extraction steps, this flow requires more
file management. However, the multiple extraction step flow provides the following benefits:

• Allows you to, optionally, perform verification at each level as you traverse the design.
• Compatible with a hierarchical flow methodology because at each level you extract
everything that is needed for the parent level including the TCD file which allows you to
forget about what is inside the parent.
• More efficient because it does not require you to provide the full content of your netlist
from the child to the top level. You need only load a blackbox (or optionally graybox)
for the core being extracted and the netlist of the level to which you are extracting.

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Overview
You perform core-in-core pattern retargeting by, first, doing the incremental extraction step for
each wrapped level in the design. Then, the actual retargeting is performed once at the chip level
at which time the core-level patterns are retargeted to the chip level.
Because your design may contain multiple levels between the child and chip level, the process
can be extended to multiple levels. Your design may have multiple parent cores at the same
level and/or multiple child cores at the same level.

In describing the process and steps required for generating and retargeting patterns from the
child level to the chip level, this section uses three design levels, as shown in Figure 9-7, and
refers to them as the “chip level”, “parent level”, and “child level”. However, as mentioned
before, this is a recursive flow that can be expanded to any number of levels.

For example, the parent core shown in Figure 9-7 could have multiple child cores inside it, or
the chip could have multiple parent cores inside it.

Figure 9-7. Three Levels of Wrapped Cores

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Core-in-Core Pattern Retargeting

The core-in-core scan pattern retargeting process begins at the lowest level of design hierarchy
and traverses up the design one level at a time as shown in Figure 9-8:

Figure 9-8. Core-in-Core Retargeting Process Overview

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Generation of Child Core Internal Mode Patterns

You generate child core internal mode patterns by performing the following steps at the child
core(s) level.
This is illustrated in step 1 of Figure 9-8.

1. Read full child netlist.

2. Set child mode to internal.
3. Generate child internal mode patterns.
4. Write out the child patterns.
5. Write out the child TCD file (write_core_description).
6. Optionally, write out the flat model for diagnosis.
7. Write out the fault list.
8. Verify scan patterns at the child level.

Retargeting of Child Core Internal Mode Patterns

You retarget child core internal mode patterns by first performing intermediate extraction steps
prior to the retargeting step.
This is illustrated in steps 2 and 3 of Figure 9-8. The extraction of the child core(s) to the parent
level is performed in the following steps:

1. Read the child blackbox.

2. Read the full parent netlist.
3. Configure the parent core to provide access to the child core and configure the child core
as it was configured when its patterns were generated.
4. Read the child TCD file.
5. Extract the child core connectivity and verify the child core embedding. Extraction is
performed as part of DRC when you transition to analysis mode (“set_system_mode
analysis”).
6. Write out the parent TCD file (the TCD file will include all child internal mode TCD
info and new parent TCD info).
7. Optionally, retarget child core patterns to parent level for verification of core embedding
only.
Once you get to the top level (below which the highest wrapped level is instantiated), you only
need to incrementally extract connectivity from the parent level to the top. The connectivity

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Core-in-Core Pattern Retargeting

from the child to the parent was previously extracted. Following this incremental extraction
step, pattern retargeting can be performed directly from the child level to the top. This is
described in the following steps, and also illustrated in step 4 of Figure 9-8:

1. Read the parent blackbox.

2. Read in the full chip netlist.
3. Configure the chip to provide scan access from the top to the parent.
4. Configure the parent as it was configured in the previous procedure which configured
the child for internal test and provided scan access from the parent to the child.
5. Read the parent TCD file from step 6 (which includes the extracted child TCD file).
6. Add the parent core as a core instance using add_core_instance.
7. Extract the parent core connectivity and verify the parent core embedding. Extraction is
performed as part of DRC when you transition to analysis mode (“set_system_mode
analysis”).
8. Write the chip-level TCD file. (You will need this file if you will be performing pattern
retargeting without a netlist as described in “Retargeting Patterns Without a Top-Level
Netlist” on page 352.)
9. Read the child core internal mode patterns.
10. Write the retargetable patterns.

Generation of Parent Internal Mode Patterns

You can generate parent-level internal mode patterns, by performing the following steps at the
parent level.
This is illustrated in step 1 of Figure 9-8.

1. Read the child graybox.

2. Read the full parent netlist.
3. Set the child to external mode.
4. Set the parent to internal mode.
5. Write the parent TCD file.
6. Generate parent internal mode patterns.
7. Write the parent internal mode patterns.
8. Optionally, write out the flat model for diagnosis.
9. Write out the fault list.

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10. Verify the scan patterns.

Retargeting of Parent Internal Mode Patterns

Because you are now want to retarget the patterns that were generated at the level of the parent
to the top, you perform extraction and retargeting of the parent internal mode patterns in one
step without the intermediate extraction step.
This is illustrated by step 4 of Figure 9-8,

You do this by performing the following steps at the chip level:

1. Read the parent blackbox.

2. Read in the full chip netlist.
3. Configure the chip to parent internal mode.
4. Read the parent TCD file from step 5.
5. Add the parent core as a core instance using add_core_instance.
6. Extract the parent core connectivity to the top level and verify embedding of the parent
core. Extraction is performed as part of DRC when you transition to analysis mode
(“set_system_mode analysis”).
7. Write the top-level TCD file.
8. Retarget the parent core internal mode patterns to the chip level.

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Limitations

Limitations
The limitations of the scan pattern retargeting functionality are listed here.
• DRCs exist to validate that the design setup at the top level is consistent with the setup
that was used for core-level ATPG. But this validation is not complete. For example, the
capture cycle clocking is not validated. Consequently, you should perform chip-level
serial simulation of a small number of scan patterns to verify the setup is correct.
• Core-level ATPG has limited flat model support. You can save a flat model in analysis
mode for performing diagnosis or rerunning ATPG with that same configuration, but
you must perform the initial design setup (setup mode) and design rule checking on a
Verilog design and not a flat model. Note, you cannot access the
is_excluded_from_isolation_constraints port attribute when running on a flat model.
• During the retargeting phase, flat models are not supported. A Verilog design must be
read in. A flat model cannot be read in or written out because it lacks information
necessary for retargeting.
• Scan pattern retargeting does not support multiple scan groups.
• When generating retargetable core-level patterns, native launch-off-shift is not
supported. Native launch-off-shift is one of two methods for generating launch-off-shift
patterns.
Native launch-off-shift — With this method, the transition tested is triggered by the last
shift applied by the shift procedure. The capture occurs when ATPG generates a single
cycle test. This method is not supported. When generating retargetable patterns for the
transition fault type, launch-off-shift will be automatically disabled as if you had
invoked the “set_fault_type transition” command with the
“-no_shift_launch” option.
Pseudo launch-off-shift — This method, which is supported, is modeled within the
capture cycles of ATPG. The patterns typically include two cycles. During the first
capture cycle, the design is kept in shift mode. During the second cycle, the scan enable
is de-asserted and the capture is performed. This method is more commonly used
because it allows the tool to perform shift and capture at-speed using PLL clocks.
• The shadow_control, shadow_observe, and master_observe procedures are not
supported. You should not use these procedures during core-level ATPG or in the top-
level test procedure file.
• Do not use STARTPAT and ENDPAT when simulating retargeted patterns that contain
multi-load core patterns.When multi-load patterns from a core are retargeted to top-level
chip patterns, the top-level pattern count does not align with the original core-level
patterns. Each retargeted load is marked as a pattern. When saving these retargeted
patterns in a Verilog testbench, if you used the STARTPAT and ENDPAT plusargs,
they refer to the top-level pattern numbers, and it might be possible to erroneously start a
simulation in the middle of a multi-load core pattern.

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Test Pattern Formatting and Timing

This chapter explains test pattern timing and procedure files.

Figure 10-1 shows a basic process flow for defining test pattern timing.

Figure 10-1. Defining Basic Timing Process Flow

Test
Procedure Internal Test
File Pattern Set

ATPG Flow

1. Use “write_procfile -Full” to

generate complete procedure file

2. Examine procedure file, modifying

with new timing if necessary

3. Use read_procfile to load in

new procedure file
4. Issue write_patterns command

Tester Format
Patterns
with Timing

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Test Pattern Timing Overview

Test Pattern Timing Overview

Test procedure files contain both scan and non-scan procedures. All timing for all pattern
information, both scan and non-scan, is defined in this procedure file.
While the ATPG process itself does not require test procedure files to contain real timing
information, automatic test equipment (ATE) and some simulators do require this information.
Therefore, you must modify the test procedure files you use for ATPG to include real timing
information. “General Timing Issues” on page 366 discusses how you add timing information
to existing test procedures.

After creating real timing for the test procedures, you are ready to save the patterns. You use the
write_patterns command with the proper format to create a test pattern set with timing
information. For more information, refer to “Saving Timing Patterns” on page 369.

Test procedures contain groups of statements that define scan-related events. See “Test
Procedure File” on the Tessent Shell User’s Manual.

Timing Terminology
The following list defines some timing-related terms:
• Non-return Timing — Primary inputs that change, at most, once during a test cycle.
• Offset — The timeframe in a test cycle in which pin values change.
• Period — The duration of pin timing—one or more test cycles.
• Return Timing — Primary inputs, typically clocks, that pulse high or low during every
test cycle. Return timing indicates that the pin starts at one logic level, changes, and
returns to the original logic level before the cycle ends.
• Suppressible Return Timing — Primary inputs that can exhibit return timing during a
test cycle, although not necessarily.

General Timing Issues

ATEs require test data in a cycle-based format. The patterns you apply to such equipment must
specify the waveforms of each input, output, or bidirectional pin, for each test cycle.
Within a test cycle, a device under test must abide by the following restrictions:

• At most, each non-clock input pin changes once in a test cycle. However, different input
pins can change at different times.
• Each clock input pin is at its off-state at both the start and end of a test cycle.

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Generating a Procedure File

• At most, each clock input pin changes twice in a test cycle. However, different clock
pins can change at different times.
• Each output pin has only one expected value during a test cycle. However, the
equipment can measure different output pin values at different times.
• A bidirectional pin acts as either an input or an output, but not both, during a single test
cycle.
To avoid adverse timing problems, the following timing requirements satisfy some ATE timing
constraints:

• Unused Outputs — By default, test procedures without measure events (all procedures
except shift) strobe unused outputs at a time of cycle/2, and end the strobe at 3*cycle/4.
The shift procedure strobes unused outputs at the same time as the scan output pin.
• Unused Inputs — By default, all unused input pins in a test procedure have a force
offset of 0.
• Unused Clock Pins — By default, unused clock pins in a test procedure have an offset
of cycle/4 and a width of cycle/2, where cycle is the duration of each cycle in the test
procedure.
• Pattern Loading and Unloading — During the load_unload procedure, when one
pattern loads, the result from the previous pattern unloads. When the tool loads the first
pattern, the unload values are X. After the tool loads the last pattern, it loads a pattern of
X’s so it can simultaneously unload the values resulting from the final pattern.
• Events Between Loading and Unloading (“patterns -scan” context only) — If other
events occur between the current unloading and the next loading, in order to load and
unload the scan chain simultaneously, The tool performs the events in the following
order:
a. Observe Procedure Only — The tool performs the observe procedure before
loading and unloading.
b. Initial Force Only — The tool performs the initial force before loading and
unloading.
c. Both Observe Procedure and Initial Force — The tool performs the observe
procedures followed by the initial force before loading and unloading.

Generating a Procedure File

The basic process flow for defining test pattern timing is as follows:
1. Use the “write_procfile -Full” command and switch to generate a complete procedure
file.
2. Examine the procedure file, modify timeplates with new timing if necessary.

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Defining and Modifying Timeplates

3. Use the read_procfile command to load in the revised procedure file.

4. Issue the write_patterns command.
The “Test Procedure File” section of the Tessent Shell User’s Manual gives an in depth
description of how to create a procedure file.

There are three ways to load existing procedure file information into the tool:

• During SETUP mode, use the add_scan_groups procedure_filename command. Any

timing information in these procedure files will be used when write_patterns is issued if
no other timing information or procedure information is loaded.
• Use the read_procfile command. This is only valid when not in SETUP mode. Using this
command loads a new procedure file that will overwrite or merge with the procedure
and timing data already loaded. This new data is now in effect for all subsequent
write_patterns commands.
• If you specify a new procedure file on the write_patterns command line, the timing
information in that procedure file will be used for that write_patterns command only,
and then the previous information will be restored.

Defining and Modifying Timeplates

This section gives an overview of the test procedure file timeplate syntax, to facilitate Step 2 in
the process flow listed previously.
For a more detailed overview of timeplates, see the “Timeplate Definition” section of the
Tessent Shell User’s Manual.

After you have used the “write_procfile -full” command to generate a procedure file, you can
examine the procedure file, modifying timeplates with new timing if necessary. Any timing
changes to the existing timeplates, cannot change the event order of the timeplate used for scan
procedures. The times may change, but the event order must be maintained.

The following example shows the contents of a timeplate, where there are two timing edges
happening at time 20, and both are listed as timing edge 4. These can be skewed, but they
cannot cross any other timing edge. The timing edges must stay in the order listed in the
comments:

force_pi 0; // timing edge 1

bidi_force_pi 12; // timing edge 3
measure_po 31; // timing edge 7
bidi_measure_po 32; // timing edge 8
force InPin 9; // timing edge 2
measure OutPin 35; // timing edge 9
pulse Clk1 20 5; // timing edge 4 & 5, respectively
pulse Clk2 20 10; // timing edge 4 & 6, respectively
period 50; // all timing edges have to happen in period

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Saving Timing Patterns

The test procedure file syntax and format is explained in “Timeplate Definition” in the Tessent
Shell User’s Manual. Keep in mind that the timeplate definition describes a single tester cycle
and specifies where in that cycle all event edges are placed. You must define all timeplates
before they are referenced. A procedure file must have at least one timeplate definition. All
clocks must be defined in the timeplate definition.

Saving Timing Patterns

You can write the patterns generated during the ATPG process both for timing simulation and
use on the ATE. Once you create the proper timing information in a test procedure file, the tool
uses an internal test pattern data formatter to generate the patterns in the following formats:
• Text format (ASCII)
• Binary format
• Wave Generation Language (WGL)
• Standard Test Interface Language (STIL)
• Verilog
• Texas Instruments Test Description Language (TDL 91)
• Fujitsu Test data Description Language (FTDL-E)
• Mitsubishi Test Description Language (MITDL)
• Toshiba Standard Tester interface Language 2 (TSTL2)

Features of the Formatter

The main features of the test pattern data formatter include:
• Generating basic test pattern data formats. text, Verilog, and WGL (ASCII and binary).
• Generating ASIC Vendor test data formats: TDL 91, FTDL-E, MITDL, and TSTL2.
• Supporting parallel load of scan cells (in Verilog format).
• Reading in external input patterns and output responses, and directly translating to one
of the formats.
• Reading in external input patterns, performing good or faulty machine simulation to
generate output responses, and then translating to any of the formats.
• Writing out just a subset of patterns in any test data format.
• Facilitating failure analysis by having the test data files cross-reference information
between tester cycle numbers and pattern numbers.
• Supporting differential scan input pins for each simulation data format.

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Pattern Formatting Issues

The following subsections describe issues you should understand regarding the test pattern
formatter and pattern saving process.

Serial Versus Parallel Scan Chain Loading

When you simulate test patterns, most of the time is spent loading and unloading the scan
chains, as opposed to actually simulating the circuit response to a test pattern. You can use
either serial or parallel loading, and each affects the total simulation time differently.
The primary advantage of simulating serial loading is that it emulates how patterns are loaded
on the tester. You thus obtain a very realistic indication of circuit operation. The disadvantage is
that for each pattern, you must clock the scan chain registers at least as many times as you have
scan cells in the longest chain. For large designs, simulating serial loading takes an extremely
long time to process a full set of patterns.

The primary advantage of simulating parallel loading of the scan chains is it greatly reduces
simulation time compared to serial loading. You can directly (in parallel) load the simulation
model with the necessary test pattern values because you have access, in the simulator, to
internal nodes in the design. Parallel loading makes it practical for you to perform timing
simulations for the entire pattern set in a reasonable time using popular simulators like
ModelSim that utilize the Verilog format.

Parallel Scan Chain Loading

You accomplish parallel loading through the scan input and scan output pins of scan sub-chains
(a chain of one or more scan cells, modeled as a single library model) because these pins are
unique to both the timing simulator model and the Tessent Shell internal model. For example,
you can parallel load the scan chain by using Verilog force statements to change the value of the
scan input pin of each sub-chain.
After the parallel load, you apply the shift procedure a few times (depending on the number of
scan cells in the longest subchain, but usually only once) to load the scan-in value into the sub-
chains. Simulating the shift procedure only a few times can dramatically improve timing
simulation performance. You can then observe the scan-out value at the scan output pin of each
sub-chain.

Parallel loading ensures that all memory elements in the scan sub-chains achieve the same states
as when serially loaded. Also, this technique is independent of the scan design style or type of
scan cells the design uses. Moreover, when writing patterns using parallel loading, you do not
have to specify the mapping of the memory elements in a sub-chain between the timing
simulator and Tessent Shell. This method does not constrain library model development for
scan cells.

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Note
When your design contains at least one stable-high scan cell, the shift procedure period
must exceed the shift clock off time. If the shift procedure period is less than or equal to
the shift clock off time, you may encounter timing violations during simulation. The test
pattern formatter checks for this condition and issues an appropriate error message when
it encounters a violation.

For example, the test pattern timing checker would issue an error message when reading in the
following shift procedure and its corresponding timeplate:

timeplate gen_tp1 =
force_pi 0;
measure_po 100;
pulse CLK 200 100;
period 300; // Period same as shift clock off time
end;

procedure shift =
scan_group grp1;
timeplate gen_tp1;
cycle =
force_sci;
measure_sco;
pulse CLK; // Force shift clock on and off
end;
end;

The error message would state:

// Error: There is at least one stable high scan cell in the design. The
shift procedure period must be greater than the shift clock off time to
avoid simulation timing violations.

The following modified timeplate would pass timing rules checks:

timeplate gen_tp1 =
force_pi 0;
measure_po 100;
pulse CLK 200 100;
period 400; // Period greater than shift clock off time
end;

Sampling to Reduce Serial Loading Simulation Time

When you use the write_patterns command, you can save a sample of the full pattern set by
using the -Sample switch. This reduces the number of patterns in the pattern file(s), reducing
simulation time accordingly. In addition, the -Sample switch allows you to control how many
patterns of each type are included in the sample. By varying the number of sample patterns, you
can fine-tune the trade-off between file size and simulation time for serial patterns.

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Note
Using the -Start and -End switches limits file size as well, but the portion of internal
patterns saved does not provide a very reliable indication of pattern characteristics when
simulated. Sampled patterns more closely approximate the results you would obtain from
the entire pattern set.

After performing initial verification with parallel loading, you can use a sampled pattern set for
simulating series loading until you are satisfied test coverage is reasonably close to desired
specification. Then, perform a series loading simulation with the unsampled pattern set only
once, as your last verification step.

Note
The set_pattern_filtering command serves a similar purpose to the -Sample switch of the
write_patterns command. The set_pattern_filtering command creates a temporary set of
sampled patterns within the tool.

Test Pattern Data Support for IDDQ

For best results, you should measure current after each non-scan cycle if doing so catches
additional IDDQ faults. However, you can only measure current at specific places in the test
pattern sequence, typically at the end of the test cycle boundary. To identify when IDDQ current
measurement can occur, the pattern file adds the following command at the appropriate places:
measure IDDQ ALL;

Several test pattern data formats support IDDQ testing. There are special IDDQ measurement
constructs in TDL 91 (Texas Instruments), MITDL (Mitsubishi), TSTL2 (Toshiba), and FTDL-
E (Fujitsu). The tools add these constructs to the test data files. All other formats (WGL and
Verilog) represent these statements as comments.

Saving Patterns in Basic Test Data Formats

The write_patterns command saves the patterns in the basic test data formats including text,
binary, Verilog, and WGL (ASCII and binary). You can use these formats for timing
simulation.

Text Format
This is the default format that the tool generates when you run the write_patterns command. The
tool can read back in this format in addition to WGL, STIL, and binary format.
This format contains test pattern data in a text-based parallel format, along with pattern
boundary specifications. The main pattern block calls the appropriate test procedures, while the
header contains test coverage statistics and the necessary environment variable settings. This

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format also contains each of the scan test procedures, as well as information about each scan
memory element in the design.

To create a basic text format file, enter the following at the application command line:

ANALYSIS> write_patterns filename -ascii

The formatter writes the complete test data to the file named filename.

For more information on the write_patterns command and its options, refer to the write_patterns
description in the Tessent Shell Reference Manual.

Note
This pattern format does not contain explicit timing information. For more information
about this test pattern format, refer to “Test Pattern File Formats” on page 379.

Comparing the Text Format with Other Test Data Formats

The text format describes the contents of the test set in a human readable form. In many cases,
you may find it useful to compare the contents of a simulation or test data format with that of the
text format for debugging purposes. This section provides detailed information necessary for
this task.
Often, the first cycle in a test set must perform certain tasks. The first test cycle in all test data
formats turns off the clocks at all clock pins, drives Z on all bidirectional pins, drives an X on all
other input pins, and disables measurement at any primary output pins.

The test pattern set can contain two main parts: the chain test block, to detect faults in the scan
chain, and the scan test or cycle test block, to detect other system faults.

The Chain Test Block

The chain test applies the test_setup procedure, followed by the load_unload procedure for
loading scan chains, and the load_unload procedure again for unloading scan chains. Each
load_unload procedure in turn calls the shift procedure. This operation typically loads a
repeating pattern of “0011” into the chains. However, if scan chains with less than four cells
exist, then the operation loads and unloads a repeating “01” pattern followed by a repeating
“10” pattern. Also, when multiple scan chains in a group share a common scan input pin, the
chain test process separately loads and unloads each of the scan chains with the repeating
pattern to test them in sequence.
The test procedure file applies each event in a test procedure at the specified time. Each test
procedure corresponds to one or more test cycles. Each test procedure can have a test cycle with
a different timing definition. By default, all events use a timescale of 1 ns.

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Note
If you specify a capture clock with the set_capture_clock command, the test pattern
formatter does not produce the chain test block. For example, the formatter does not
produce a chain test block for IEEE 1149.1 devices in which you specify a capture clock
during tool setup.

The Scan Test Block

The scan test block in the pattern set starts with an application of the test_setup procedure. The
scan test block contains several test patterns, each of which typically applies the load_unload
procedure, forces the primary inputs, measures the primary outputs, and pulses a capture clock.
The load_unload procedure translates to one or more test cycles. The force, measure, and clock
pulse events in the pattern translate to the ATPG-generated capture cycle.
Each event has a sequence number within the test cycle. The sequence number’s default time
scale is 1 ns.

Unloading of the scan chains for the current pattern occurs concurrently with the loading of scan
chains for the next pattern. Therefore the last pattern in the test set contains an extra application
of the load_unload sequence.

More complex scan styles (for example, like LSSD) use master_observe and skewed_load
procedures in the pattern. For designs with sequential controllers, like boundary scan designs,
each test procedure may have several test cycles in it to operate the sequential scan controller.
Some pattern types (for example, RAM sequential and clock sequential types) are more
complex than the basic patterns. RAM sequential patterns involve multiple loads of the scan
chains and multiple applications of the RAM write clock. Clock sequential patterns involve
multiple capture cycles after loading the scan chains. Another special type of pattern is the
clock_po pattern. In these patterns, clocks may be held active throughout the test cycle and
without applying capture clocks.

If the test data format supports only a single timing definition, the tool cannot save both
clock_po and non-clock_po patterns in one pattern set. This is so because the tester cannot
reproduce one clock waveform that meets the requirements of both types of patterns. Each
pattern type (combinational, clock_po, ram_sequential, and clock_sequential) can have a
separate timing definition.

General Considerations
During a test procedure, you may leave many pins unspecified. Unspecified primary input pins
retain their previous state.

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Note
If you run ATPG after setting pin constraints, you should also ensure that you set these
pins to their constrained states at the end of the test_setup procedure. The
add_input_constraints command constrains pins for the non-scan cycles, not the test
procedures. If you do not properly constrain the pins within the test_setup procedure, the
tool does it for you, internally adding the extra force events after the test_setup
procedure. This increases the period of the test_setup procedure by one time unit. This
increased period can conflict with the test cycle period, potentially forcing you to re-run
ATPG with the modified test procedure file.

All test data formats contain comment lines that indicate the beginning of each test block and
each test pattern. You can use these comments to correlate the test data in the text format with
other test data formats.

These comment lines also contain the cycle count and the loop count, which help correlate tester
pattern data with the original test pattern data. The cycle count represents the number of test
cycles, with the shift sequence counted as one cycle. The loop count represents the number of
all test cycles, including the shift cycles. The cycle count is useful if the tester has a separate
memory buffer for scan patterns, otherwise the loop count is more relevant.

Note
The cycle count and loop count contain information for all test cycles—including the test
cycles corresponding to test procedures. You can use this information to correlate tester
failures to a pattern for fault diagnosis.

Binary
This format contains test pattern data in a binary parallel format, which is the only format (other
than text format) that the tool can read. A file generated in this format contains the same
information as text format, but uses a condensed form. You should use this format for archival
purposes or when storing intermediate results for very large designs.
To create a binary format file, enter the following command:

ANALYSIS> write_patterns filename -binary

The tool writes the complete test data to the file named filename.

For more information about the write_patterns command and its options, refer to the
write_patterns description in the Tessent Shell Reference Manual.

Verilog
This format contains test pattern data and timing information in a text-based format readable by
both the Verilog and Verifault simulators. This format also supports both serial and parallel

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loading of scan cells. The Verilog format supports all Tessent Shell timing definitions because
Verilog stimulus is a sequence of timed events.
To generate a basic Verilog format test pattern file, use the following arguments with the
write_patterns command:

write_patterns filename [-Parallel | -Serial] -Verilog

The Verilog pattern file contains procedures to apply the test patterns, compare expected output
with simulated output, and print out a report containing information about failing comparisons.
The tools write all patterns and comparison functions into one main file (filename), while
writing the primary output names in another file (filename.po.name). If you choose parallel
loading, they also write the names of the scan output pins of each scan sub-chain of each scan
chain in separate files (for example, filename.chain1.name). This allows the tools to report
output pins that have discrepancies between the expected and simulated outputs. For more
information about Verilog test benches, refer to “The Verilog Test Bench” on page 303.

For more information about the write_patterns command and its options, refer to the
write_patterns description in the Tessent Shell Reference Manual.

Wave Generation Language (ASCII)

The Wave Generation Language (WGL) format contains test pattern data and timing
information in a structured text-based format. You can translate this format into a variety of
simulation and tester environments, but you must first read it into the Waveform database and
use the appropriate translator. This format supports both serial and parallel loading of scan cells.
Some test data flows verify patterns by translating WGL to stimulus and response files for use
by the chip foundry’s golden simulator. Sometimes this translation process uses its own parallel
loading scheme, called memory-to-memory mapping, for scan simulation. In this scheme, each
scan memory element in the ATPG model must have the same name as the corresponding
memory element in the simulation model. Due to the limitations of this parallel loading scheme,
you should ensure the following:

1) there is only one scan cell for each DFT library model (also called a scan subchain), 2) the
hierarchical scan cell names in the netlist and DFT library match those of the golden simulator
(because the scan cell names in the ATPG model appear in the scan section of the parallel WGL
output), and 3) the scan-in and scan-out pin names of all scan cells are the same.

To generate a basic WGL format test pattern file, use the following arguments with the
write_patterns command:

write_patterns filename [-Parallel | -Serial] -Wgl

For more information about the write_patterns command and its options, refer to the
write_patterns description in the Tessent Shell Reference Manual.

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Standard Test Interface Language (STIL)

To generate a STIL format test pattern file, use the following arguments with the write_patterns
command:
write_patterns filename [-Parallel | -Serial] -STIL

For more information about the write_patterns command and its options, refer to the
write_patterns description in the Tessent Shell Reference Manual.

Saving in ASIC Vendor Data Formats

The ASIC vendor test data formats include Texas Instruments TDL 91, Fujitsu FTDL-E,
Mitsubishi MITDL, and Toshiba TSTL2. The ASIC vendor’s chip testers use these formats.
All the ASIC vendor data formats are text-based and load data into scan cells in a parallel
manner. Also, ASIC vendors usually impose several restrictions on pattern timing. Most ASIC
vendor pattern formats support only a single timing definition. Refer to your ASIC vendor for
test pattern formatting and other requirements.

The following subsections briefly describe the ASIC vendor pattern formats.

TI TDL 91
This format contains test pattern data in a text-based format.
The tool supports features of TDL 91 version 3.0 and of TDL 91 version 6.0. The version 3.0
format supports multiple scan chains, but allows only a single timing definition for all test
cycles. Thus, all test cycles must use the timing of the main capture cycle. TI’s ASIC division
imposes the additional restriction that comparison should always be done at the end of a tester
cycle.

To generate a basic TI TDL 91 format test pattern file, use the following arguments with the
write_patterns command:

write_patterns filename -TItdl

The formatter writes the complete test data to the file filename. It also writes the chain test to
another file (filename.chain) for separate use during the TI ASIC flow.

For more information about the write_patterns command and its options, refer to the
write_patterns description in the Tessent Shell Reference Manual.

Fujitsu FTDL-E
This format contains test pattern data in a text-based format. The FTDL-E format splits test data
into patterns that measure 1 or 0 values, and patterns that measure Z values. The test patterns
divide into test blocks that each contain 64K tester cycles.

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To generate a basic FTDL-E format test pattern file, use the following arguments with the
write_patterns command:

write_patterns filename -Fjtdl

The formatter writes the complete test data to the file named filename.fjtdl.func. If the test
pattern set contains IDDQ measurements, the formatter creates a separate DC parametric test
block in a file named filename.ftjtl.dc.

For more information about the write_patterns command and its options, refer to the
write_patterns description in the Tessent Shell Reference Manual.

Mitsubishi TDL
This format contains test pattern data in a text-based format. To generate a basic Mitsubishi Test
Description Language (TDL) format test pattern file, use the following arguments with the
write_patterns command:
write_patterns filename -MItdl

The formatter represents all scan data in a parallel format. It writes the test data into two files:
the program file (filename.td0), which contains all pin definitions, timing definitions, and scan
chain definitions; and the test data file (filename.td1), which contains the actual test vector data
in a parallel format.

For more information about the write_patterns command and its options, refer to the
write_patterns description in the Tessent Shell Reference Manual.

Toshiba TSTL2
This format contains only test pattern data in a text-based format. The test pattern data files
contain timing information. This format supports multiple scan chains, but allows only a single
timing definition for all test cycles. TSTL2 represents all scan data in a parallel format.
To generate a basic Toshiba TSTL2 format test pattern file, use the following arguments with
the write_patterns command:

write_patterns filename -TSTl2

The formatter writes the complete test data to the file named filename. For more information
about the write_patterns command and its options, refer to the write_patterns description in the
Tessent Shell Reference Manual.

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Test Pattern File Formats

This chapter provides information about the ASCII and BIST pattern file formats.

ASCII File Format

The ASCII file that describes the scan test patterns is divided into five sections, which are
named header_data, setup_data, functional_chain_test, scan_test, and scan_cell. Each section
(except the header_data section) begins with a section_name statement and ends with an end
statement. Also in this file, any line starting with a double slash (//) is a comment line. The
format of the data contained in each section is described as follows.

Header_Data
The header_data section contains the general information, or comments, associated with the test
patterns. This is an optional section that requires a double slash (//) at the beginning of each line
in this section. The data printed may be in the following format:
// model_build_version - the version of the model build program that was used to create
the scan model.
// design_name - the design name of the circuit to be tested.
// date - the date in which the scan model creation was performed.
// statistics - the test coverage, the number of faults for each fault class, and the total
number of test patterns.
// settings - the description of the environment in which the ATPG is performed.
// messages - any warning messages about bus contention, pins held, equivalent pins,
clock rules, and so on are noted.

Setup_Data
The setup_data section contains the definition of the scan structure and general test procedures
that will be referenced in the description of the test patterns.
Note
Additional formats are added to the setup_data section for BIST patterns. For information
on these formats, see “BIST Pattern File Format” on page 387”.

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The data printed will be in the following format:

SETUP =
<setup information>
END;

The setup information will include the following:

declare input bus “PI” = <ordered list of primary inputs>;

This defines the list of primary inputs that are contained in the circuit. Each primary input will
be enclosed in double quotes and be separated by commas. For bidirectional pins, they will be
placed in both the input and output bus.

declare output bus “PO” = <ordered list of primary outputs>;

This defines the list of primary outputs that are contained in the circuit. Each primary output
will be enclosed in double quotes and be separated by commas.

CLOCK “clock_name1” =
OFF_STATE = <off_state_value>;
PULSE_WIDTH = <pulse_width_value>;
END;
CLOCK “clock_name2” =
OFF_STATE = <off_state_value>;
PULSE_WIDTH = <pulse_width_value>;
END;

This defines the list of clocks that are contained in the circuit. The clock data will include the
clock name enclosed in double quotes, the off-state value, and the pulse width value. For edge-
triggered scan cells, the off-state is the value that places the initial state of the capturing
transition at the clock input of the scan cell.

WRITE_CONTROL “primary_input_name” =
OFF_STATE = <off_state_value>;
PULSE_WIDTH = <pulse_width_value>;
END;

This defines the list of write control lines that are contained in the circuit. The write control line
will include the primary input name enclosed in double quotes, the off-state value, and the pulse
width value. If there are multiple write control lines, they must be pulsed at the same time.

PROCEDURE TEST_SETUP “test_setup” =

FORCE “primary_input_name1” <value> <time>;
FORCE “primary_input_name2” <value> <time>;
....
....
END;

This is an optional procedure that can be used to set nonscan memory elements to a constant
state for both ATPG and the load/unload process. It is applied once at the beginning of the test
pattern set. This procedure may only include force commands.

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SCAN_GROUP “scan_group_name1” =
<scan_group_information>
END;
SCAN_GROUP “scan_group_name2” =
<scan_group_information>
END;
....
....

This defines each scan chain group that is contained in the circuit. A scan chain group is a set of
scan chains that are loaded and unloaded in parallel. The scan group name will be enclosed in
double quotes and each scan group will have its own independent scan group section. Within a
scan group section, there is information associated with that scan group, such as scan chain
definitions and procedures.

SCAN_CHAIN “scan_chain_name1” =
SCAN_IN = “scan_in_pin”;
SCAN_OUT = “scan_out_pin”;
LENGTH = <length_of_scan_chain>;
END;
SCAN_CHAIN “scan_chain_name2” =
SCAN_IN = “scan_in_pin”;
SCAN_OUT = “scan_out_pin”;
LENGTH = <length_of_scan_chain>;
END;
....
....

The scan chain definition defines the data associated with a scan chain in the circuit. If there are
multiple scan chains within one scan group, each scan chain will have its own independent scan
chain definition. The scan chain name will be enclosed in double quotes. The scan-in pin will be
the name of the primary input scan-in pin enclosed in double quotes. The scan-out pin will be
the name of the primary output scan-out pin enclosed in double quotes. The length of the scan
chain will be the number of scan cells in the scan chain.

PROCEDURE <procedure_type> “scan_group_procedure_name” =

<list of events>
END;

The type of procedures may include shift procedure, load and unload procedure, shadow-control
procedure, master-observe procedure, shadow-observe procedure, and skew-load procedure.
The list of events may be any combination of the following commands:

FORCE “primary_input_pin” <value> <time>;

This command is used to force a value (0,1, X, or Z) on a selected primary input pin at a given
time. The time values must not be lower than previous time values for that procedure. The time
for each procedure begins again at time 0. The primary input pin will be enclosed in double
quotes.

APPLY “scan_group_procedure_name” <#times> <time>;

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This command indicates the selected procedure name is to be applied the selected number of
times beginning at the selected time. The scan group procedure name will be enclosed in double
quotes. This command may only be used inside the load and unload procedures.

FORCE_SCI “scan_chain_name” <time>;

This command indicates the time in the shift procedure that values are to be placed on the scan
chain inputs. The scan chain name will be enclosed in double quotes.

MEASURE_SCO “scan_chain_name” <time>;

This command indicates the time in the shift procedure that values are to be measured on the
scan chain outputs. The scan chain name will be enclosed in double quotes.

Functional_Chain_Test
The functional_chain_test section contains a definition of a functional scan chain test for all
scan chains in the circuit to be tested. For each scan chain group, the scan chain test will include
a load of alternating double zeros and double ones (00110011...) followed by an unload of those
values for all scan chains of the group. The format is as follows:
CHAIN_TEST =
APPLY “test_setup” <value> <time>;
PATTERN = <number>;
APPLY “scan_group_load_name” <time> =
CHAIN “scan_chain_name1” = “values....”;
CHAIN “scan_chain_name2” = “values....”;
....
....
END;
APPLY “scan_group_unload_name” <time> =
CHAIN “scan_chain_name1” = “values....”;
CHAIN “scan_chain_name2” = “values....”;
....
....
END;
END;

The optional “test_setup” line is applied at the beginning of the functional chain test pattern if
there is a test_setup procedure in the Setup_Data section. The number for the pattern is a zero-
based pattern number where a functional scan chain test for all scan chains in the circuit is to be
tested. The scan group load and unload name and the scan chain name will be enclosed in
double quotes. The values to load and unload the scan chain will be enclosed in double quotes.

During the loading of the scan chains, each value of the corresponding scan chain will be placed
at its scan chain input pin. The shift procedure will shift the value through the scan chain and
continue shifting the next value until all values for all the scan chains have been loaded. Since
the number of shifts is determined by the length of the longest scan chain, X’s (don’t care) are
placed at the beginning of the shorter scan chains. This will ensure that all the values of the scan
chains will be loaded properly.

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During the unloading of the scan chains, each value of the corresponding scan chain will be
measured at its scan chain output pin. The shift procedure will shift the value out of the scan
chain and continue shifting the next value until all values for all the scan chains have been
unloaded. Again, since the number of shifts is determined by the length of the longest scan
chain, X’s (don’t measure) are placed at the end of the shorter scan chains. This will ensure that
all the values of the scan chains will be unloaded properly.

Here is an example of a functional scan chain test:

CHAIN_TEST =
APPLY “test_setup” 1 0;
PATTERN = 0;
APPLY “g1_load” 0 =
CHAIN “c2” = “XXXXXXXXX0011001100110011001100”;
CHAIN “c1” = “XXXXXXXXXXXXX001100110011001100”;
CHAIN “c0” = “0011001100110011001100110011001”;
END;
APPLY “g1_unload” 1 =
CHAIN “c2” = “0011001100110011001100XXXXXXXXX”;
CHAIN “c1” = “001100110011001100XXXXXXXXXXXXX”;
CHAIN “c0” = “0011001100110011001100110011001”
END;
END;

Scan_Test
The scan_test section contains the definition of the scan test patterns that were created by
Tessent Shell. A scan pattern will normally include the following:
SCAN_TEST =
PATTERN = <number>;
FORCE “PI” “primary_input_values” <time>;
APPLY “scan_group_load_name” <time> =
CHAIN “scan_chain_name1” = “values....”;
CHAIN “scan_chain_name2” = “values....”;
....
....
END;
FORCE “PI” “primary_input_values” <time>;
MEASURE “PO” “primary_output_values” <time>;
PULSE “capture_clock_name1” <time>;
PULSE “capture_clock_name2” <time>;
APPLY “scan_group_unload_name” <time> =
CHAIN “scan_chain_name1” = “values....”;
CHAIN “scan_chain_name2” = “values....”;
....
....
END;
....
....
....
END;
The number of the pattern represents the pattern number in which the scan chain is loaded,
values are placed and measured, any capture clock is pulsed, and the scan chain is unloaded.

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The pattern number is zero-based and must start with zero. An additional force statement will be
applied at the beginning of each test pattern, if transition faults are used. The scan group load
and unload names and the scan chain names will be enclosed by double quotes. All the time
values for a pattern must not be lower than the previous time values in that pattern. The values
to load and unload the scan chain will be enclosed in double quotes. Refer to the
“Functional_Chain_Test” section on how the loading and unloading of the scan chain operates.

The primary input values will be in the order of a one-to-one correspondence with the primary
inputs defined in the setup section. The primary output values will also be in the order of a one-
to-one correspondence with the primary outputs defined in the setup section.

If there is a test_setup procedure in the Setup_Data section, the first event, which is applying the
test_setup procedure, must occur before the first pattern is applied:

APPLY “test_setup” <value> <time>;

If there are any write control lines, they will be pulsed after the values have been applied at the
primary inputs:

PULSE “write_control_input_name” <time>;

If there are capture clocks, then they will be pulsed at the same selected time, after the values
have been measured at the primary outputs. Any scan clock may be used to capture the data into
the scan cells that become observed.

Scan patterns will reference the appropriate test procedures to define how to control and observe
the scan cells. If the contents of a master is to be placed into the output of its scan cell where it
may be observed by applying the unload operation, the master_observe procedure must be
applied before the unloading of the scan chains:

APPLY “scan_group_master_observe_name” <value> <time>;

If the contents of a shadow is to be placed into the output of its scan cell where it may be
observed by applying the unload operation, the shadow_observe procedure must be applied
before the unloading of the scan chains:

APPLY “scan_group_shadow_observe_name” <value> <time>;

If the master and slave of a scan cell are to be at different values for detection, the skew_load
procedure must be applied after the scan chains are loaded:

APPLY “scan_group_skew_load_name” <value> <time>;

Each scan pattern will have the property that it is independent of all other scan patterns. The
normal scan pattern will contain the following events:

1. Load values into the scan chains.

2. Force values on all non-clock primary inputs.

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3. Measure all primary outputs not connected to scan clocks.

4. Exercise a capture clock. (optional)
5. Apply observe procedure (if necessary)
6. Unload values from scan chains.
When unloading the last pattern, the tool loads the last pattern a second time to completely shift
the contents of the last capture cycle so that the output matches the calculated value. For more
information, see “Handling of Last Patterns” in the Tessent TestKompress User’s Manual.

Although the load and unload operations are given separately, it is highly recommended that the
load be performed simultaneously with the unload of the preceding pattern when applying the
patterns at the tester.

For observation of primary outputs connected to clocks, there will be an additional kind of scan
pattern that contains the following events:

1. Load values into the scan chains.

2. Force values on all primary inputs including clocks.
3. Measure all primary outputs that are connected to scan clocks.

Scan_Cell
The scan_cell section contains the definition of the scan cells used in the circuit. The scan cell
data will be in the following format:
SCAN_CELLS =
SCAN_GROUP “group_name1” =
SCAN-CHAIN “chain_name1” =
SCAN_CELL = <cellid> <type> <sciinv> <scoinv>
<relsciinv> <relscoinv> <instance_name>
<model_name> <input_pin> <output_pin>;
....
END;
SCAN_CHAIN “chain_name2” =
SCAN_CELL = <cellid> <type> <sciinv> <scoinv>
<relsciinv> <relscoinv> <instance_name>
<model_name> <input_pin> <output_pin>;
....
END;
....
END;
....
END;

The fields for the scan cell memory elements are the following:

• cellid - A number that identifies the position of the scan cell in the scan chain. The
number 0 indicates the scan cell closest to the scan-out pin.

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• type - The type of scan memory element. The type may be MASTER, SLAVE,
SHADOW, OBS_SHADOW, COPY, or EXTRA.
• sciinv - Inversion of the library input pin of the scan cell relative to the scan chain input
pin. The value may be T (inversion) or F (no inversion).
• scoinv - Inversion of the library output pin of the scan cell relative to the scan chain
output pin. The value may be T (inversion) or F (no inversion).
• relsciinv - Inversion of the memory element relative to the library input pin of the scan
cell. The value may be T (inversion) or F (no inversion).
• relscoinv - Inversion of the memory element relative to the library output pin of the scan
cell. The value may be T (inversion) or F (no inversion).
• instance_name - The top level boundary instance name of the memory element in the
scan cell.
• model_name - The internal instance pathname of the memory element in the scan cell
(if used - blank otherwise).
• input_pin - The library input pin of the scan cell (if it exists, blank otherwise).
• output_pin - The library output pin of the scan cell (if it exists, blank otherwise).

Example Circuit
Figure 11-1 illustrates the scan cell elements in a typical scan circuit.

Figure 11-1. Example Scan Circuit

Boundary (set up
in model definition)
scan_in scan_out

sciinv Scan cell scoinv

In Out

T if odd # of inverters T if odd # of inverters

F if even # of inverters F if even # of inverters

relsciinv relscoinv

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BIST Pattern File Format

BIST Pattern File Format

All of the test pattern file formats described in the previous section directly apply to BIST
patterns also. However, an additional set of BIST-specific formats are added to the ASCII
pattern file format. The BIST-specific formats are described in this section.

Setup_Data
The setup_data section for BIST contains a subsection that defines the BIST pattern-specific
configuration. This subsection includes a file version identifier and definitions for signature
registers (prpg_register and misr_register) in the BIST pattern.
A BIST-specific statement is used to identify that the following configuration is BIST pattern-
specific. The BIST-specific statement has the following format:

declare bist pattern specific configuration =

A version tag is used to indicate the BIST pattern version. The version tag has the following
format:

BIST_ASCII_PATTERN_FILE_SUBVERSION

Here is an example of a BIST pattern-specific statement combined with a version tag statement:

declare bist pattern specific configuration =

BIST_ASCII_PATTERN_FILE_SUBVERSION = 1;
Note
The “bist pattern specific configuration” statement disappears if a LFSM value is not
included.

A pattern_internal_view switch that indicates whether the BIST pattern internal view is written
to the pattern file; its possible values are “ON” or “OFF”. If the switch is set to on, as shown in
the following example, BIST patterns are included in the pattern file. For information about
pattern format types, refer to the LBISTArchitect Reference Manual.

pattern_internal_view = “on”

The prpg_register and misr_register identifiers define signature registers such as PRPG and
MISR in the BIST pattern. The signature register statements use the following format:

signature_register <name of signature register> =

length = <length of the register>;
type = <type of the register>;
init_value = <initial state of the register>;
end;

Here are some examples of signature register statements:

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prpg_register “decomp1” =
length = 22;
type = PRPG;
init_value = “0000000000000000000000”;
end;

prpg_register “decomp2” =
length = 12;
type = PRPG;
init_value = “000000000000”;
end;

prpg_register “prpg1” =
length = 16;
type = PRPG;
init_value = “0000000000000000”;
end;

prpg_register “prpg2” =
length = 16;
type = PRPG;
init_value = “0000000000000000”;
end;

misr_register “misr1” =
length = 32;
type = MISR;
init_value = “11111111111111111111111111111111”;
end;

misr_register “misr2” =
length = 24;
type = MISR;
init_value = “111111111111111111111111”;
end;

Scan_Test
Each BIST pattern includes one lfsm_snapshot statement. Within the pattern statement, the
keyword pre_load, pre_unload, or post_unload is followed by a register name to indicate when
the snapshot for the register is to be taken. The keywords are applied as follows:
• pre_load — Used for PRPG type registers.
• pre_unload and post_unload — Used for MISR type registers.
The snapshot statement is composed so that the simulation of each BIST pattern is independent
from all others:

• Given the pre_load value of PRPG registers, the loading data for each BIST pattern can
be computed.
• Given the pre_unload value of MISR registers, the MISR registers after unload can be
computed and thus verified.

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BIST Pattern File Format

Here is an example of a pattern containing a lfsm_snapshot statement:

pattern = 0;
lfsm_snapshot =
pre_load “decomp1” = “0000000000000000000000”;
pre_load “decomp2” = “000000000000”;
pre_load “prpg1” = “1111111111111111”;
pre_load “prpg2” = “1111111111111111”;
pre_unload “misr1” = “11111111111111111111111111111111”;
pre_unload “misr2” = “111111111111111111111111”;
post_unload “misr1” = “11111111110001111110111111101111”;
post_unload “misr2” = “111111110000000011111111”;
end;

apply “grp1_load” 0 =
...
end;

force “PI” “....” 1;

measure “PO” “....” 2;

apply “grp1_unload” 3 =
...
end;

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 Chapter 12
Power-Aware DRC and ATPG

This chapter describes the power-aware DRC and ATPG flow for use with the ATPG tool, and
contains the following sections:
• Power-Aware Overview
o Assumptions and Limitations
o Multiple Power Mode Test Flow
o Power-Aware ATPG for Traditional Fault Models
• CPF and UPF Parser
• Power-Aware ATPG Procedure
• Power-Aware Flow Examples

Power-Aware Overview
The electronics industry has adopted low-power features in major aspects of the design
continuum. In response, EDA venders and major semiconductor companies have defined the
commonly-used power data standard formats to describe the power requirements: UPF and
CPF. Tessent Shell supports the following versions of the UPF and CPF formats:
• IEEE 1801 standard / UPF 2.0
• Common Power Format (CPF) 1.0 and 1.1
You load this power data directly into the tool to collect the power information. Once loaded,
the tools perform the necessary DRC’s to ensure that the DFT logic is inserted properly with
respect to the design’s power domains and, if the design passes the rule checks, perform ATPG
with the given power mode configuration. For information about the power-aware DRC (V)
rules, refer to the Tessent Shell Reference Manual.

The tool’s low-power functionality provides you with a method to do the following:

• Provide DRC’s to trace the active power mode and ensure that the scan operation works
under the current power configuration.
• Provide capability to generate test for the traditional fault models while aware of the
power mode configuration.

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Power-Aware Overview

Assumptions and Limitations

The power-aware functionality comes with the following assumptions and limitations:
• Circuit hierarchy is preserved in the CPF or UPF file and is the same as the netlist. Note
that for modular EDT design, the module hierarchy that includes EDT logic needs to be
preserved, otherwise the tool may not be able to associate an EDT block with a power
domain. Consequently, some power DRC rules related to EDT logic may not be
performed.
• The power-aware DRC rules and reporting are based on the loaded power data (UPF or
CPF). The DRC rules do not include testing that crosses different power modes.
Additionally, the DRC rules do not cover the testing of the shutoff power domain such
as the retention cells testing.

Multiple Power Mode Test Flow

For a design with multiple test modes, the scan chain configuration may be different for each
power mode. You should perform the DRC and ATPG separately for each power mode using
the following steps:
1. Configure the power mode to be tested using the test_setup procedure.
2. Load the CPF file. The tool automatically identifies the power mode that the system is
currently configured to and report it to user.
In addition, new V rules are defined and are checked when you switch to a to non-SETUP
mode. (See “Power-Aware Rules (V Rules) in the Tessent Shell Reference Manual.) This ensure
that the loaded UPF/CPF file contains consistent power information and the inserted DFT logic
(for example, scan insertion) has considered the design power domains properly.

Note
For the purpose of DRC, you must load to load a UPF/CPF file in the SETUP mode.
When loading a UPF/CPF file in a non-SETUP mode, the tool does not perform V DRC
until you switch the tool from the SETUP mode to a non-SETUP mode.

Pattern Generation
ATPG generates patterns only for the current power mode. If there are different power modes
which enable the identical power domains (just in different voltage configuration), then the
pattern set can be reused by loading the pattern file and performing fault grading. Reuse of the
pattern set is at your discretion.
Despite the pattern reuse, you should still write one test_setup procedure for every power mode
to be tested, and perform DRC check for each power mode to ensure the scan chains can operate
properly in the power mode. In addition, a new pattern set should be stored to reflect the
updated test_setup for the corresponding power mode. Finally, when writing out patterns after

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Power-Aware Overview

ATPG, the tool should also save the current power mode information (as a comment) to the
pattern file for the user’s reference.

Power-Aware ATPG for Traditional Fault Models

During the traditional fault model test stage, the circuit is configured into a static power mode
by the test procedure at the beginning of the capture cycle and stays at the same power mode for
the entire capture cycle. The ATPG engine explicitly prevents the power control logic from
changing the active power mode.
A low-power DRC is performed before ATPG to check if the active power mode can be
disturbed and the analysis result is used by ATPG to determine if any additional ATPG effort is
needed to keep the static power mode. This is similar to E10 rule for the bus contention check
which is used by ATPG to enable the extra justification to prevent the bus contention. If the
circuit contains the power mode that powers on all power domains (called ALL_ON state), you
can use this state for the traditional fault models. An ALL_ON state allows the circuit to be
tested for logic faults in one test set run. In addition, this state also allows the tool to perform
DRC for entire circuit in this run.

Power Partitioning
If the circuit needs to partition with partial power domains powered in a given time, then you
must perform multiple ATPG runs; specifically, each run with its procedure file and the scan
configuration. You must ensure that every power domain is covered by at least one run.
Additionally, the chip- level test coverage can be computed manually from each separate run.
In the case of multiple power partition flow, the always on power domain, the faults may be
targeted multiple times. To reduce the creation of patterns for same faults in the always-on
domains, you can use the following command:

add_faults –power_domains always_on -delete

This explicitly removes the faults in always_on domain if the faults have been targeted by other
run.

The other way to avoid creation of duplicated patterns is to load the previous generated fault list
(using read_faults –merge) so that the detected faults by previous runs will not be re-targeted
again.

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CPF and UPF Parser

CPF and UPF Parser

The following table lists the power data commands relevant to the power-aware DRC and
ATPG process. The other CPF and UPF commands are parsed by the tool but discarded.

Table 12-1. Power Data Commands Directly Related to ATPG and DRC
Power Features IEEE 1801 / UPF 2.0 CPF 1.0 / CPF 1.1
Power Domains create_power_domain create_power_domain
add_domain_elements update_power_domain
create_composite_domain
merge_power_domain
Power Modes add_power_state1 assert_illegal_domain_configurations
(Power States) add_pst_state create_assertion_control2
create_pst create_power_mode
update_power_mode
State Transitions describe_state_transition3 create_mode_transition
Power Network add_port_state
(to derive power-on create_power_switch
domains and active map_power_switch4
state) set_power_switch
create_supply_port
create_supply_net
connect_supply_net
set_domain_supply_net
Retention Cells map_retention_cell create_state_retention_rule
set_retention update_state_retention_rule
set_retention_control define_state_retention_cell6
set_retention_elements5
Isolation Cells map_isolation_cell create_isolation_rule
set_isolation update_isolation_rule
set_isolation_control define_isolation_cell8
use_interface_cell7
Level Shifters map_level_shifter_cell create_level_shifter_rule
set_level_shifter update_level_shifter_rule
use_interface_cell9 define_level_shifter_cell10

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CPF and UPF Parser

Table 12-1. Power Data Commands Directly Related to ATPG and DRC
Power Features IEEE 1801 / UPF 2.0 CPF 1.0 / CPF 1.1
Design Scope set_design_top Include
set_scope get_parameter
upf_version set_design
load_upf set_instance
load_upf_protected set_cpf_version
Tcl support set_time_unit
set_power_mode_control_group
set_array_naming_style
set_macro_model
end_macro_model
Tcl support
1. add_power_state, describe_state_transition and map_power_switch are not used currently for DRC and
ATPG purpose.
2. The CPF commands for defining power cells are not used by ATPG to identify the power cells that exist in
the design. These commands may be needed during test synthesis to automatically insert power cells.
3. add_power_state, describe_state_transition and map_power_switch are not used currently for DRC and
ATPG purpose.
4. add_power_state, describe_state_transition and map_power_switch are not used currently for DRC and
ATPG purposes.
5. The UPF commands set_retention_elements and use_interface_cell are not used to identify the location of
power cells. These commands may be needed during test synthesis to automatically insert power cells.
6. The CPF commands for defining power cells are used for DRC but are not used by ATPG to identify the
power cells that exist in the design. These commands may be needed during test synthesis to automatically
insert power cells.
7. The UPF commands set_retention_elements and use_interface_cell are not used to identify the location of
power cells. These commands may be needed during test synthesis to automatically insert power cells.
8. The CPF commands for defining power cells are used for DRC but are not used by ATPG to identify the
power cells that exist in the design. These commands may be needed during test synthesis to automatically
insert power cells.
9. The UPF commands set_retention_elements and use_interface_cell are not used to identify the location of
power cells. These commands may be needed during test synthesis to automatically insert power cells.
10. The CPF commands for defining power cells are used for DRC but are not used by ATPG to identify the
power cells that exist in the design. These commands may be needed during test synthesis to automatically
insert power cells.

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Power-Aware ATPG Procedure

Power-Aware ATPG Procedure

When using the power-aware ATPG flow, you should separate the test for the logic faults from
the test for the low-power features. When testing the logic faults, the traditional fault models
(for example, the stuck-at fault model and the transition fault model) are applied and the power
switch logic need not to be presented.
The power data in UPF or CPF format should be applied to the tool so the low-power rules (see
“Power-Aware Rules (V Rules)”) can be checked and the circuit can be simulated according to
the active power mode.

Procedure
In general, you perform the following steps with the power-aware ATPG flow, either directly
from the command line or scripted in a dofile:

1. Invoke the Tessent Shell, set the context the “patterns -scan,” and read in the low-power
design. For example:
SETUP> read_verilog low-power_design.v

2. Add scan chains and other configuration information as appropriate.

3. Load the power data using the read_cpf or read_upf command. For example:
SETUP> read_upf test.upf

4. Set the mode to analysis. For example:

SETUP> set_system_mode analysis

At this point, the tool performs the DRC checks. See “Power-Aware Rules (V Rules) in
the Tessent Shell Reference Manual.
5. Add faults using the applicable power-aware switch to the add_faults command. For
example:
SETUP> add_faults -on_domains

6. Optionally write the faults to a fault list for multiple power-aware test methodologies—
see “Multiple Power Mode Test Flow.” For example:
SETUP> write_faults on_domain_fault_list.txt

7. Create the patterns using the create_patterns command.

The power-aware ATPG flow supports reporting mechanisms tailored to the power information.
See the following commands for more information:

• report_faults
• report_power_data

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Power-Aware Flow Examples

This section contains examples of the power-aware flow.

Example 1
Table 12-2 shows one example design with four power domains and four power modes.

Table 12-2. Example Design With Four Power domains and Power Modes
Power Domains
Power Modes CPU MEM1 CTL Radio test_setup file
active Active Active ON Tx,Rx test_setup_active
standby Idle Sleep ON Rx test_setup_standby
idle Sleep Sleep ON Rx test_setup_idle
sleep Sleep Sleep OFF Off N/A
chain1
scan chains chain2 chain3 chain4 none

The design contains four scan chains arrayed as follows:

• Chain1 and Chain 2 — Located in the CPU power domain.

• Chain3 — Located in MEM1 power domain.
• Chain4 — Located in the CTL power domain.
To test the design requires multiple test_setup procedures, one for each power mode except for
the one turning off all power domains. To reduce the maintenance overhead of test procedure
files, test_setup procedures are written in the following separate files:

• test_setup_active
• test_setup_standby
• test_setup_idle
The main test procedure file, which contains the rest of procedures, uses an include statement to
include the test_setup for the power mode to be tested. For example, using the following include
statement:

# include test_setup_active”

will test the “active” power mode.

The recommended flow is to test the power mode with all power domains active first, if such
power mode exists (power mode “active” in Table 12-2). This allows the tool to view all scan

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Power-Aware Flow Examples

chains and check the “Power-Aware Rules (V Rules)” rules crossing all power domains. Note
that when testing different power modes, the scan chains need to be defined accordingly to
prevent DRC violations. For example when testing “standby” mode where “MEM1” power
domain is off, chain3 should not be defined, otherwise it is a V8 DRC violation.

See ““Power-Aware Rules (V Rules)” in the Tessent Shell Reference Manual for a complete
discussion.

Example 2
In this example, the design contains the following:

• Three power domains {D1, D2, D3}

• Two power mode
o S1: (D1=ON, D2=ON, D3=OFF)
o S2: (D1=OFF, D2=ON, D3=ON)
Additionally, assume that the design holds PS1 property (specifically, during the shift cycles
and capture cycles), and the design can be kept in the same power mode.

To test the design, you must perform the following multiple tool runs:

• run 1 — ATPG for power mode S1

a. Configure the design to power mode S1 by the end of test_setup.
b. Add scan chains only in power domain D1 and D2.
c. Add faults only in the power ON domains (D1 and D2) using
the add_faults –on_domains command.
d. Create patterns and write patterns.
e. Write faults to a file named flist_S1.txt using the write_faults command.
f. Write isolation faults to a file named flist_S1_iso.txt using write_faults –isolation to
save for later use.
g. Write level-shifter faults to a file named flist_S1_ls.txt using
write_faults –level_shifter to save for later use.
• run 2 — ATPG for power mode S2
a. Configure the design to power mode S2 by the end of test_setup.
b. Add scan chains only in power domain D2 and D3.
c. Add faults only in the power ON domains (D2 and D3) using add_faults -all
command.

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Power-Aware Flow Examples

d. Load the previously-saved S1 fault list using the following command:

ANALYSIS> read_faults flist_S1.txt –merge –Power_check on

This updates the fault status of faults in D2 but discard faults in D3 as they are not
power on faults. See the read_faults command for more information.
e. Create patterns and write patterns.
f. Write faults to a file named flist_S2.txt using the write_faults command.
g. Write isolation faults to a file named flist_S2_iso.txt using write_faults –isolation to
save for later use.
h. Write level-shifter faults to a file named flist_S2_ls.txt using
write_faults –level_shifter to save for later use.
After run 2, you can calculate the entire fault coverage by loading multiple fault lists into the
tool. You do this by using the read_faults –POwer_check OFF command and switch as shown
in the following steps:

• run 3 — Overall test coverage report

a. Load the previously-saved fault list for D1:
ANALYSIS> read_faults flist_S1.txt –merge –Power_check off

b. Load the previously-saved fault list for D2:

ANALYSIS> read_faults flist_S2.txt –merge –Power_check off

c. Issue the report_statistics command to report the overall test coverage.

d. Write the faults to a file named faults flist_all.txt using the write_faults command.
The final tool run reports isolation fault test coverage ATPG and level-shifter fault test coverage
and saves these in separate fault lists.

1. Report the isolation fault test coverage:

ANALYSIS> report_statistics -isolation

2. Report the level-shifter fault test coverage:

ANALYSIS> report_statistics –level_shifter

3. Write all isolation faults to a file:

ANALYSIS> write_faults flist_all_iso.txt –isolation

4. Write all level-shifter faults to a file:

ANALYSIS> write_faults flist_all_ls.txt –level_shfiter

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 Chapter 13
Testing Low-Power Designs

This chapter describes the low-power scan insertion and ATPG flow for use with Tessent Scan
and the ATPG tool, and contains the following sections:
Low-Power Testing Overview. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 402
Assumptions and Limitations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 402
Low-Power CPF/UPF Parameters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 402
Test Insertion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 404
Low-Power Test Flow. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 404
Scan Insertion with Tessent Scan . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 405
Power-Aware Design Rule Checks . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 409
Low-Power DRCs. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 409
Low-Power DRC Troubleshooting . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 409
Power State-Aware ATPG . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 411
Power Domain Testing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 411
Low-Power Cell Testing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 413

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Low-Power Testing Overview

Low-Power Testing Overview

Tessent Scan supports the following operations:
• Insertion of dedicated wrapper cells in the presence of isolation cells.
• Assigning dedicated wrapper cells to power domains based on the logic it is driving and
the priority of the power domains.
The low-power design flow includes the following steps:

1. Specify low-power data specifications in the CPF/UPF file. See “Low-Power CPF/UPF
Parameters”.
2. Insert scan cells and EDT logic in the design. See “Test Insertion”.
3. Validate low-power data, scan, and EDT logic. See “Power-Aware Design Rule
Checks”.
4. Generate power domain-aware test patterns. See “Power State-Aware ATPG”.
5. Test low-power design components.

Assumptions and Limitations

The power-aware functionality comes with the following assumptions and limitations:
• Tessent Scan does not write or update CPF/UPF files. Tessent Scan assumes that
everything that belongs to a power domain is explicitly listed in the UPF/CPF file.
Because of this assumption, you must make the following changes to the UPF/CPF file
after the scan insertion step and before ATPG:
o Add any inserted input wrapper cells to the correct power domain in the CPF/UPF
file.
o Explicitly define all isolation cells and their control signals, and level shifters in the
CPF/UPF file. Tessent Scan checks for their presence but does not add them. For
more information, see “Scan Insertion with Tessent Scan.”

Low-Power CPF/UPF Parameters

Tessent Scan uses CPF/UPF files to identify the power domains in the design and to constrain
scan insertion within the power domain boundaries. Tessent Scan supports power structure and
configuration data from the following formats:
• CPF 1.0 and 1.1
• UPF 2.0 (IEEE1801)

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Low-Power Testing Overview

If you are a member of Si2, you can access complete information about CPF standards at the
following URL:

https://www.si2.org/openeda.si2.org/project/showfiles.php?group_id=51

You can purchase complete information on UPF standards at the following URL:

http://www.techstreet.com/standards/ieee/1801_2009?product_id=1744966

In the CPF/UPF file, you must specify power domains and information related to the power
domains, power states (modes) of the design, and isolation cells and control signals. You can
use the commands listed in Table 13-1 to specify this information.

Table 13-1. Power Data Commands Directly Related to ATPG

Power Features IEEE 1801 / UPF 2.0 CPF 1.0 / CPF 1.1
Power Domains create_power_domain create_power_domain
add_domain_elements update_power_domain
create_composite_domain
merge_power_domain
Power Modes add_power_state assert_illegal_domain_configurations
(Power States) add_pst_state create_assertion_control
create_pst create_power_mode
update_power_mode
Isolation Cells set_isolation_command create_isolation_rule
set_isolation_control define_isolation_cell
map_isolation_cell update_isolation_cell
Level Shifters map_level_shifter_cell create_level_shifter_rule
update_level_shifter_rule

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Test Insertion

Test Insertion
Test insertion includes scan substitution and stitching. You can insert scan chains on a power
domain basis. Alternately, you can group test logic based on power states.
Figure 13-1 show an example of chains inserted on a power domain basis. Note, no scan chains
span the power domains because the tool does not insert scan across power domains.

Figure 13-1. Scan Chain Insertion by Power Domain

Low-Power Test Flow

This section outlines the steps for first loading low-power data and then using the loaded power
domain information to guide the process of scan partitioning and scan insertion.
1. Load the design data and libraries.
2. Read the CPF or UPF file:
read_cpf filename
read_upf filename

3. Set up appropriate design constraints.

4. Enter analysis mode:
SETUP> set_system_mode analysis

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5. If wrapper cells are enabled, scan chain partitioning will be affected. Refer to
“Managing Wrapper Cells with Power Domains” for more information.
6. Report the scan partitions created based on the power domain information in the
CPF/UPF file:
> report_scan_partitions -all
-------------------------------------------------------------------------
ScanPartitionName TotalNumCells/ScannableCells Members
-------------------------------------------------------------------------
PD_interleaver 16693/16693 i0 [instance]
PD_mem_ctrl 12/12 mc0 [instance]
default_scan_partition 9/9 <all_remaining_cells>
-------------------------------------------------------------------------

7. Specify scan chains based on maximum scan chain length or scan chain number using
the set_power_domain command:
set_power_domain {{Power_domain_name… | -All [-EDT]}
[-Number_of_chains integer | -Max_length integer]}

For example:
> set_power_domain PD_interleaver -number 65
> set_power_domain PD_mem_ctrl -max_length 6

Any scan flip-flop not specified by the set_power_domain command is added to the
default top-level power domain.
8. Insert the test structures into the design netlist using the insert_test_logic command.
9. Report the design statistics using the report_statistics command.
10. Write out the netlist and ATPG setup.

Scan Insertion with Tessent Scan

You can use Tessent Scan to do both scan cell substitution and scan stitching. By loading the
low-power CPF or UPF file for the design before beginning the scan insertion and stitching
process, the power domain information can be used to guide the scan partitioning process.
Tessent Scan does not insert scan across power domains.
You can explicitly specify the number and length of scan chains in each power domain of the
design using the set_power_domain command. If you do not use set_power_domain, Tessent
Scan adopts the default settings based on the following command/options settings:

• insert_test_logic: -MAx_length and -NUmber arguments

• add_scan_partition: -NUmber and -MAx_length arguments
• set_wrapper_chains: -INPUT_NUMber, -INPUT_MAX_length, -OUTPUT_NUMber,
and -OUTPUT_MAX_length arguments

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Note
Because Tessent Scan does not function with a concept of power mode or power state,
chain balancing does not occur during low-power scan insertion.

During scan insertion, the tool does not generate a design that is clean in terms of power
domains. Tessent Scan creates a design that is functionally correct in the context of power
domains but it does not insert isolation cells or level shifters when routing global signals across
power domains. You will need to add these objects themselves using a synthesis or physical
implementation tool and then manually add them to the CPF/UPF file.

For more information on isolation cells and level shifters, see “Low-Power Cell Testing”.

Managing Scan Enable Signals for Multiple Power

Domains
The low-power test flow allows you to insert multiple scan enable signals in your design. This
means you can insert a scan enable signal for each power domain in the design; you can also
insert multiple scan enable signals for a single power domain. The behavior of multiple scan
enable signals is the same regardless of whether your design has no power domains defined or
has multiple power domains defined.
You assign one scan enable signal to a power domain by using the set_scan_enable command
and specifying the power domain with the partition_name argument as shown in the example
here:

set_scan_enable scan_en_pin_path pd1 pd2

For more information on assigning a scan enable signal to a power domain, refer to the
set_scan_enable description in the Tessent Shell Reference Manual.

To assign more than one scan enable signal to a single power domain, you use the
set_scan_enable_sharing command and specify the -scan_partition switch which assigns a
unique scan enable signal to each of the specified power domains. By default, the types of the
added scan enable signals are different for the core chains, input wrapper chains, and output
wrapper chains

set_scan_enable_sharing -prefix base_name -scan_partition

Managing Subchains and Wrapper Chains with Power

Domains
By default, the set_scan_enable command assigns scan enable signals to core scan chains. You
can use the -wrapper_chain argument to specify the type of scan chain the scan enable signals
are to be assigned to.

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Test Insertion

For more information, refer to the set_scan_enable description in the Tessent Shell Reference
Manual.

Managing Wrapper Cells with Power Domains

Newly-inserted dedicated wrapper cells are stitched into wrapper chains based on the power
domain affiliation of the wrapper cells (dedicated and shared) within the wrapper chains. The
tool identifies the power domain a dedicated wrapper cell is affiliated with during wrapper cell
identification; the power domain is chosen based on which power domain the PI or PO being
registered is connected to. If a PI is connected to more than one power domain, the dedicated
wrapper cell is associated with the first encountered power domain.
Figure 13-2 shows the stitching of the input wrapper cells for each power domain. The
newly added dedicated wrapper cells are stitched with the shared wrapper cells in the same
power domain.

Figure 13-2. Input Wrapper Cells and Power Domains

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Figure 13-2 shows the stitching of output wrapper cells for each power domain. The newly
added dedicated wrapper cells are stitched with the shared wrapper cells in the same power
domain.

Figure 13-3. Output Wrapper Cells and Power Domains

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Power-Aware Design Rule Checks

Power-Aware Design Rule Checks

Tessent Scan performs power-aware design rule checking to validate power data and
power-aware reporting to facilitate troubleshooting design rule violations.

Low-Power DRCs
In addition to checking design logic and the cell library, power-aware design rules
(DRCs V1-V21) check the design for violations and correctness of power data provided through
CPF/UPF files.
Note
DFTVisualizer does not have any direct support for power domain-related DRC failures.
However, you can use standard design tracing features in DFTVisualizer to pinpoint
issues.

• DRCs V1 to V7 — Checked after reading a power data file.

• DRCs V8 to V21 — Checked when you switch from setup mode to a non-setup mode.
For information on specific V DRCs, see “Power-Aware Rules (V Rules)” in the Tessent Shell
Reference Manual.

Low-Power DRC Troubleshooting

You can use the report_scan_partitions command to generate a scan cell report that includes
power domain information as shown in the following example. Tessent Scan treats power
domains as scan partitions during scan chain stitching.
The report_scan_partitions command extracts all updated information from the CPF/UPF files
and reports each power domain (scan partition) and the pathnames of all of the sequential
instances in that power domain; newly added dedicated wrapper cells and lockup cells are
indicated by the string “(new cell)” following their name.

DFT> report_scan_partitions -all -expand

-------------------------------------------
ScanPartitionName Members
-------------------------------------------
PD2 I6
I10/lckup3 (new cell)
I10/lckup2 (new cell)
I10/lckup1 (new cell)
I10/outreg2 (new cell)
I10/outreg1 (new cell)
I10/sf_0001_1
I10/sf_0002_1
I10/sf_0003_1
PD3 I2/I2/lckup3 (new cell)

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I2/I2/lckup2 (new cell)

I2/I2/lckup1 (new cell)
I2/I2/outreg1 (new cell)
I2/I2/sf_0001_1
I2/I2/sf_0002_1
I2/I2/sf_0003_1
I2/I2/I1_2
DEF lckup2 (new cell)
lckup1 (new cell)
inreg2 (new cell)
inreg1 (new cell)
outreg1 (new cell)
-------------------------------------------

Reporting Power Domains for Scan Cells

You can use the report_scan_cells command to generate a scan cell report that includes power
domain information as shown here:
ANALYSIS> report_scan_cells -power_domain

For more information, refer to the report_scan_cells description in the Tessent Shell Reference
Manual.

Reporting Power Domains for Gates

You can use the set_gate_report command to add power domain information to the results of the
report_gates report as shown in the example here:
ANALYSIS> set_gate_report -power on
ANALYSIS> report_gates 154421
// /mc0/present_state_reg_3_/inst1122_4/mlc_dff (154421) DFF
// "S" I(ON) 153682-
// "R" I(ON) 153681-
// "C0" I(ON) 26078-
// "D0" I(ON) 43013-
// "OUT" O(ON) 1115-
// MASTER cell_id=0 chain=chain66 group=grp1 invert_data=FFFF
// in power domain PD_mem_ctrl

For more information, refer to the set_gate_report description in the Tessent Shell Reference
Manual.

Tracing Power Domain Violations

You can use DFTVisualizer design tracing capabilities to pinpoint failures and issues reported
during design rule checking.
For more information on pinpointing DRC violations in DFTVisualizer, see “Analysis a DRC
Violation” in the Tessent Shell Reference Manual.

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Power State-Aware ATPG

Power State-Aware ATPG

The recommended method for ATPG is to have all power domains on if possible. With this
method, you can manage power considerations due to switching activity during test by
sequencing the test of design blocks and using low-power ATPG. This is the simplest approach
and maximizes fault coverage. If you cannot use this method due to static power limitations,
you should select a minimum set of power states to achieve the required fault coverage for all
the blocks in the design as well as inter-block connectivity.

Power Domain Testing

You can run ATPG in multiple power modes and power states by setting the appropriate power
mode/state conditions as specified in the CPF or UPF. To demonstrate, assume a design has the
following two power modes:
• PM1 — All domains on
• PM2 — PD_mem_ctrl powered down when mc_pwr is “0”
By setting a pin constraint on mc_pwr as shown here, ATPG recognizes which power mode is
currently set and runs appropriately:

SETUP> add_input_constraints mc_pwr -c1

SETUP> set_system_mode analysis
…
// ---------------------------------------------------------------------
// Begin circuit learning analyses.
// ---------------------------------------------------------------------
…
SETUP> create_patterns
No faults in fault list. Adding all faults...
…
// Enabling power-aware ATPG: initial power mode in capture cycle = PM2
…

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If the power mode can change during the capture cycle, the tool will issue V12 violations. When
V12 handling is a warning (default), the tool automatically adds ATPG constraints to fix the
power mode during the capture cycles:

…
// 1 ATPG constraint is added to fix power mode during capture.
// The active power mode at the beginning of capture cycle is PM2.
// Power-aware ATPG is enabled.
…
When power-aware ATPG is enabled, the tool performs ATPG and fault simulation according
to the power mode: the regular gates in a power-off domain will be X and the corresponding
faults are AU. The fault grouping will classify the fault as AU due to power off if faults in a
power domain are added:

ANALYSIS> create_patterns
ANALYSIS> report_statistics
---------------------- --------------
FU (full) 354
-------------------- --------------
DS (det_simulation) 52 (14.69%)
DI (det_implication) 40 (11.30%)
UU (unused) 26 ( 7.34%)
RE (redundant) 134 (37.85%)
AU (atpg_untestable) 102 (28.81%)
--------------------------------------
Untested Faults
--------------------
AU (atpg_untestable)
POFF (power_off) 102 (28.81%)
--------------------------------------
Coverage
--------------------
test_coverage 47.42%
fault_coverage 25.99%
atpg_effectiveness 100.00%
--------------------------------------

Instead of adding all faults, you can choose to add, delete, report, and write faults on power-on
domains or any user-specified power domains:

add_/delete_/report_/write_faults
[[-ON_domains] | [-OFf_domains] |
[ -POWer_domains {domain_name …}]]
[ -ISolation_cells] [ -LEvel_shifters ] [ -REtention_cells ]

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Power State-Aware ATPG

Low-Power Cell Testing

The following sections describe issues related to test coverage.

Level Shifters
In most cases, you can handle level shifters as standard buffers and do not need any special
handling to achieve full test coverage. However, if one or more power domains in the design
can operate at more than one supply voltage, you should run ATPG for all permutations of the
supply voltage for both input and output sides of any given level shifter to ensure full coverage.

Isolation Cells
You can test isolation cells in one of two modes:
• Normal transmission mode — In this mode, when an isolation cell is disabled and
stuck-at “0” or “1”, the fault can be detected on the input and output of the cell. A
stuck-at fault (isolation on) for the isolation enable pin is also detected because this
failure forces the cell into isolation and prevents the transmission of data.
• Isolation mode — In this mode, input side stuck-at faults are not detectable if the
isolation cell is a clamp-style cell (output held to “0” or “1” when isolation on). If the
isolation cell is a latch-style cell, input side stuck-at faults are detectable by latching the
value of the driving domain before enabling isolation.

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Using MTFI Files

This chapter describes MTFI (Mentor Tessent Fault Information) features, which are available
for use in the ATPG tool and Tessent LogicBIST. MTFI is a common and extendable file
format for storing fault status information.
This chapter describes MTFI syntax, as well as the major MTFI features:

• “MTFI File Format” on page 416

• “Support of Fault Classes and Sub-Classes” on page 419
• “Support of Stuck and Transition Fault Information” on page 421
• “Support of N-Detect Values” on page 421
• “Support of Different Fault Types in the Same File” on page 423
• “Support for Hierarchical Fault Accounting” on page 423
• “Commands that Support MTFI” on page 425

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MTFI File Format

MTFI File Format

MTFI is the default (but optional) file format for reading and writing fault information in the
“dft -edt” and “patterns -scan” contexts.
MTFI is an ASCII file format for storing fault information about an instance that the ATPG tool
can read and write.
Format
The following simple MTFI example shows the minimum required keywords in bold:
FaultInformation {
version : 1;
FaultType (Stuck) {
FaultList {
Format : Identifier, Class, Location;
Instance ("/i1") {
0, DS, "F1";
1, DS, "F2";
}
}
}
}

Parameters
The basic syntax elements of MTFI are as follows:
• Comments
The “//” identifies the start of the comment until the end of line.
• FaultInformation { … }
Keyword that specifies the top-level block and file type.
• version : integer
Keyword that specifies the syntax version of the MTFI file.
• FaultType (type) { … }
Keyword that specifies the fault type of the data: Stuck, Iddq, Toggle, Transition,
Path_delay, Bridge, and Udfm. The keywords are identical to those available for the
set_fault_type command described in the Tessent Shell Reference Manual.
• FaultList { … }
Keyword that defines a data block that stores per-fault data.
• UnlistedFaultsData { … }
Keyword that defines a data block that stores the coverage information for specific
graybox instances to allow hierarchical fault accounting. For more information, refer to
“Support for Hierarchical Fault Accounting” on page 423.

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MTFI File Format

• FaultCollapsing { true | false }

Keyword that specifies the fault collapsing status in the fault list. A value of “true”
means that the faults in the list are collapsed; a value of “false” means that the faults in
the list are uncollapsed. You can use this statement only inside the FaultList data block.
• DetectionLimit : integer
Keyword that specifies the detection drop limit for a DS fault.
• Format
Keyword that specifies the sequence of values stored in a specific data block (FaultList
or UnlistedFaultsData). The following keywords are available for use in the Format
statement:
• Class
Required keyword that specifies the fault category, and optionally the sub category
(for example, DS, UC, AU.BB). For more information, refer to “Support of Fault
Classes and Sub-Classes” on page 419.
• Location
Keyword that specifies the pin pathname. You can use this keyword only in the
FaultList data block. In the case of a stuck-at fault, the keyword is required.
• Identifier
Required keyword that specifies the fault identifier. In the case of stuck-at or
transition faults, the value can be 0 or 1. No other fault types are supported.
• Detections
Optional keyword that specifies the number of detections for the fault types stuck-at
and transition. For more information, refer to “Support of N-Detect Values” on
page 421.
• CollapsedFaultsCount
Required keyword that specifies the number of collapsed faults. You can use this
keyword only in the UnlistedFaultsData data block.
• UncollapsedFaultsCount
Required keyword that specifies the number of uncollapsed faults. You can use this
keyword only in the UnlistedFaultsData data block.
• Instance ( “pathname“ ) { … }
Required keyword that specifies the instance pathname to all of the pins in the data
block. You must enclose the pathname in parentheses and double quotes. The pathname
can be an empty string.

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Example
The following is an example of a typical MTFI file:

//
// Tessent FastScan v9.6
//
// Design = test.v
// Created = Tue Dec 20 20:08:46 2011
//
// Statistics:
// Test Coverage = 50.00%
// Total Faults = 6
// UC (uncontrolled) = 2
// DS (det_simulation) = 1
// DI (det_implication) = 1
// AU (atpg_untestable) = 2
//
FaultInformation {
version : 1;
FaultType ( Stuck ){
FaultList {
FaultCollapsing : false;
Format : Identifier, Class, Location;
Instance ( “” ) {
1, DS, “/i1/IN0”;
1, AU, “/i1/IN1”;
1, EQ, “/i2/Y”;
0, UC, “/i5/Z”;
1, DI, “/i5/Z”;
0, AU, “/i4/IN1”;
1, UC, “/i4/IN1”;
}
}
}
}

MTFI Features
The following text explains the major features of the MTFI format:
• “Support of Fault Classes and Sub-Classes” on page 419
• “Support of Stuck and Transition Fault Information” on page 421
• “Support of N-Detect Values” on page 421
• “Support of Different Fault Types in the Same File” on page 423
• “Support for Hierarchical Fault Accounting” on page 423

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Support of Fault Classes and Sub-Classes

The following example shows how MTFI supports fault classes and sub-classes using the
Format statement’s Class keyword:
FaultInformation {
version : 1;
FaultType ( Stuck ){
FaultList {
FaultCollapsing : false;
Format : Identifier, Class, Location;
Instance ( “” ) {
1, DS, “/i1/IN0”;
1, AU, “/i1/IN1”;
1, EQ, “/i2/Y”;
0, UC, “/i5/Z”;
1, DI, “/i5/Z”;
0, AU, “/i4/IN1”;
1, UC, “/i4/IN1”;
}
}
}
}

In the following example, one AU fault is identified as being AU due to black box, and one DI
fault is specified as being due to LBIST. So for these two faults, the MTFI file reports both class
and sub-class values.

FaultInformation {
version : 1;
FaultType ( Stuck ) {
FaultList {
FaultCollapsing : false;
Format : Identifier, Class, Location;
Instance ( “” ) {
1, DS, “/i1/IN0”;
1, AU, “/i1/IN1”;
1, EQ, “/i2/Y”;
0, UC, “/i5/Z”;
1, DI, “/i5/Z”;
1, DI.LBIST, “/i5/Z”;
0, AU.BB, “/i4/IN1”;
1, UC, “/i4/IN1”;
}
}
}
}

In the preceding example, the sub-class information is part of the class value and separated by
the dot. Any of the AU, UC, UO and DI fault classes can be further divided into different sub-
classes. For all the available AU fault sub-classes, refer to the set_relevant_coverage command
description in the Tessent Shell Reference Manual. The tool supports the following UC and UO

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fault sub-classes: ATPG_Abort (AAB), Unsuccessful (UNS), EDT Abort (EAB). The tool
supports the pre-defined fault sub-class EDT for DI faults.

User-Defined Fault Sub-Classes

Besides the predefined fault sub-classes described previously, the tool also supports a user-
defined sub-class for AU and DI faults. The user-defined sub-class can be any string based on
the following character set:
• A-Z
• a-z
• 0-9
• -
• _
It is your responsibility to ensure that the name of the user-defined sub-class differs from any of
the predefined sub-classes. Otherwise, the faults may be accounted for incorrectly.

The statistics report displays the breakdown of DI faults when there are some DI faults in
specific subcategories. The following example shows the statistics report with seven DI faults
declared in different subcategories; in this example, the sub-class analysis for DI.EDT is
disabled:

> report_statistics
Statistics Report
Stuck-at Faults
-------------------------------------------------------------------------
Fault Classes #faults
(total)
---------------------------------------------------------- --------------
FU (full) 61
-------------------------------------------------------- -------------
DS (det_simulation) 4 ( 6.56%)
DI (det_implication) 7 (11.48%)
UU (unused) 13 (21.31%)
BL (blocked) 1 ( 1.64%)
AU (atpg_untestable) 36 (59.02%)
-------------------------------------------------------------------------
DI Faults
--------------------------------------------------------
DI (det_imp)
DUMMY 5 ( 8.20%)
DUMMY_alias 1 ( 1.64%)
dummy 1 ( 1.64%)
Untested Faults
--------------------------------------------------------
AU (atpg_untestable)
TC* (tied_cells) 27 (44.26%)
my_TC 4 ( 6.56%)

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mya_TC 5 ( 8.20%)
*Use "report_statistics -detailed_analysis" for details.
-------------------------------------------------------------------------
Coverage
--------------------------------------------------------
test_coverage 23.40%
fault_coverage 18.03%
atpg_effectiveness 100.00%
-------------------------------------------------------------------------
#test_patterns 0
#simulated_patterns 0
CPU_time (secs) 18.0
-------------------------------------------------------------------------

Support of Stuck and Transition Fault Information

MTFI is able to represent stuck fault information in a fashion similar to the classic format, as
shown in the following example:
FaultInformation {
version : 1;
FaultType ( Stuck ) {
FaultList {
FaultCollapsing : false;
Format : Identifier, Class, Location;
Instance ( “” ) {
1, UC, “/in1”;
0, DS, “/in1”;
0, DS, “/i1/IN0”;
1, DS, “/i1/OUT”;
1, DS, “/i1/IN0”;
1, AU, “/i1/IN1”;
1, EQ, “/i2/Y”;
0, UC, “/i5/Z”;
1, DS, “/out”;
1, DI, “/i5/Z”;
}
}
}
}

Support of N-Detect Values

The information about how often a fault has been detected can be handled for a single fault in
the FaultList block, as well as for a group of faults in the UnlistedFaultsData block.
In the FaultList block, you can add the value to the list using the Format statement’s Detections
keyword. The value must be a positive integer that specifies the number of detections of this
fault. All classes other than DS must have a value of 0.

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In the following example, the detection drop limit is 4. For any DS faults that are detected four
times or more, the tool reports the detection value as “4”.

FaultInformation {
version : 1;
FaultType ( Stuck ) {
FaultList {
FaultCollapsing : false;
DetectionLimit : 4;
Format : Identifier, Class, Detections, Location;
Instance ( “” ) {
1, DS, 3, “/i1/OUT”;
1, DS, 4, “/i1/IN0”;
1, AU.BB, 0, “/i1/IN1”;
1, EQ, 0, “/i2/Y”;
0, UC, 0, “/i5/Z”;
1, DS, 2, “/out”;
1, DI, 0, “/i5/Z”;
}
}
}
}

In general, the same rules are valid for the UnlistedFaultsData block, as shown in the following
example:

FaultInformation {
version : 1;
FaultType ( Stuck ) {
UnlistedFaultsData {
DetectionLimit : 4;
Format : Class,Detections,CollapsedFaultCount,
UncollapsedFaultCount;
Instance ( “/CoreD/i1” ) {
UC, 0, 1252, 2079;
UC.EAB, 0, 452, 543;
DS, 1, 69873, 87232;
DS, 2, 12873, 21432;
DS, 3, 9752, 11974;
DS, 4, 4487, 6293;
AU.BB, 0, 8708, 10046;
AU, 0, 2374, 3782;
}
}
}
}

In the case of the UnlistedFaultsData, the preceding example shows the fault count after the
number of detections. Typically, there is one line per detection until reaching the current
detection limit. So in the preceding example, the number of faults have reached the detection
limit of 4.

Also note that in the previous example, the line

UC, 0, 1252, 2079

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shows the collapsed and uncollapsed fault count for the UC faults that are in the unclassified
sub-class (that is, those that do not fall into any of the predefined or user-defined sub-classes).
Similarly, the line
AU, 0, 2374, 3782
shows the fault counts for the unclassified AU faults.

While loading the MTFI file, the n-detection number stored in the file is capped at the detection
limit currently set in the tool. This applies to both the detection number in FaultList data block
and that in UnlistedFaultsData block. Depending on the switch, the n-detection data in the
external MTFI file can be appended to the internal detection number of the corresponding fault,
or replace the detection number of the corresponding fault in the internal fault list.

Support of Different Fault Types in the Same File

MTFI allows you to manually specify multiple fault types in a single file.
When reading an MTFI file using the read_faults command, the current fault type (set using
set_fault_type) must match the specified fault type within the MTFI file. In order to share fault
information between the fault types Stuck and Transition, MTFI allows you to manually specify
both fault types in the same file, as shown in the following example:

FaultInformation {
version : 1;
FaultType ( Stuck, Transition ) {
FaultList {
FaultCollapsing : false;
Format : Identifier, Class, Location;
Instance ( "" ) {
1, AU.PC, "/i1/OUT";
1, AU.PC, "/i1/IN0";
1, AU.BB, "/i1/IN1";
1, AU.BB, "/i2/Y";
0, AU.TC, "/i5/Z";
1, AU.TC, "/out";
1, AU.TC, "/i5/Z";
}
}
}
}

MTFI does not support combinations other than Stuck and Transition. And note that none of the
tools that support MTFI can output MTFI files that contain multiple fault types.

Support for Hierarchical Fault Accounting

MTFI allows you to efficiently handle fault information for hierarchical designs, where
individual small cores are tested and their fault statistics saved in individual MTFI files. You
can then instantiate those small cores in a larger core using “graybox” versions of the small
cores plus the fault information from their MTFI files. Using this method, the test patterns for

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the larger core need not deal with the internal details of the small cores, only with their graybox
versions (which contain only the subset of the logic needed for testing at the next higher level).
Consider the example in Figure 14-1. Core_A and Core_B are stand-alone designs with their
own individual test patterns and fault lists. After running ATPG on Core_A and Core_B, you
store fault information in MTFI files using the following commands:

ANALYSIS> write_faults Core_A.mtfi

ANALYSIS> write_faults Core_B.mtfi

Note that the logic in Core_A and Core_B is fully observable and controllable, so there are no
unlisted faults in the Core_A.mtfi or Core_B.mtfi files.

If there are multiple fault lists for a given core, each reflecting the coverage achieved by a
different test mode, and the intent is to merge these results into a single coverage for the core,
you must perform the merge during a core-level ATPG run. This is because when you use the
“read_faults -graybox” command at the next level of hierarchy, the command supports only a
single fault list per core. Note that you can only merge fault lists of a single fault type. You
cannot, for example, merge fault lists for stuck and transition at the core level.

Taking the example of Core_A previously described, you would first merge the fault lists of any
other previously-run test modes before writing out the final fault list for the core. The following
sequence of commands is an example of this:

ANALYSIS> read_faults Core_A_mode1.mtfi -merge

ANALYSIS> read_faults Core_A_mode2.mtfi -merge
ANALYSIS> write_faults Core_A.mtfi

Next you create Core_C by instantiating graybox versions of Core_A and Core_B, and then you
use the fault information you wrote previously using the following commands:

ANALYSIS> read_faults Core_A.mtfi -Instance Core_C/Core_A -Graybox

ANALYSIS> read_faults Core_B.mtfi -Instance Core_C/Core_B -Graybox

The -Graybox switch directs the tool to map any faults it can to existing nets in Core_C and to
essentially discard the remaining faults by classifying them as unlisted. The tool retains the fault
statistics for the unlisted faults.

After you’ve run ATPG on Core_C, you can write out the fault information to an MTFI file:

ANALYSIS> write_faults Core_C.mtfi

The command writes out data for both the listed and unlisted faults that were created when you
issued the read_faults command. You could then instantiate a graybox version of Core_C in a
larger core, Core_D, and load the fault information from the Core_C.mtfi file. That way,
Core_A, Core_B, and Core_C will have already been fully tested and would not be tested again,
but the fault statistics from all three cores would be available for viewing and analysis with the
listed faults in Core_D.

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 Using MTFI Files
Commands that Support MTFI

Figure 14-1. Using MTFI with Hierarchical Designs

Core_A.mtfi
FaultList {
Core_A } Core_C

Core_A.v Core_C.mtfi
Core_A FaultList {
Core_A (graybox)
graybox.v
}
UnlistedFaultsData {
instance Core_A {
Core_B.mtfi
Core_B FaultList {
Core_B.v Core_B }
} instance Core_B {
(graybox)

}
Core_B }
graybox.v

Core_C.v

Key to faults in the MTFI files:

= Faults within the core that are classified as listed
(faults whose sites are present in the netlist).
= Faults within the core that are classified as unlisted
(faults whose sites are not present in the netlist).

For more information about the read_faults and write_faults commands, refer to their
descriptions in the Tessent Shell Reference Manual.

Commands that Support MTFI

The following table summarizes the commands that support MTFI.

Table 14-1. MTFI Command Summary

Command Description
read_faults Updates the current fault list with the faults listed in a
specified file.
report_faults Displays data from the current fault list.
write_faults Writes data from the current fault list to a specified file.

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Commands that Support MTFI

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 Chapter 15
Graybox Overview

Graybox functionality streamlines the process of scan insertion and ATPG processing in a
hierarchical design by allowing you to perform scan and ATPG operations on a sub-module,
and then allowing you to use a simplified, graybox representation of that sub-module when
performing scan and ATPG operations at the next higher level of hierarchy. Because the
graybox representation of a sub-module contains only a minimal amount of interconnect
circuitry, the use of grayboxes in a large, hierarchical design can dramatically reduce the
amount of memory and tool runtime required to perform scan insertion, optimize timing,
analyze faults, and create test patterns.
Note
Currently, only Mux-DFF scan architecture is supported with the graybox functionality.

Table 15-1 summarizes the commands that support graybox functionality, which is available in
the ATPG tool.

Table 15-1. Graybox Command Summary

Command Description
analyze_graybox Identifies the instances and nets to be included in the graybox
netlist.
set_attribute_value Assigns an attribute and value to specified design objects. For
a list of graybox attributes, refer to the analyze_graybox
command description.
report_graybox_statistics Reports the statistics gathered by graybox analysis.
write_design Writes the current design in Verilog netlist format to the
specified file. Optionally writes a graybox netlist.

What is a Graybox?
A graybox is a simplified representation of a sub-module that contains only the minimum
amount of interconnect circuitry (primary inputs/outputs, wrapper chains, and the glue logic
outside of the wrapper chains) required to process the grayboxed sub-module at the next higher
level of hierarchy.
To understand a graybox representation of a sub-module, first consider the full netlist
representation shown in Figure 15-1. This figure shows the input and output wrapper chains, the
core scan chains, and the combinational logic that exists both inside and outside the wrapper
chains.

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Graybox Overview
What is a Graybox?

After performing scan insertion, fault accounting, and pattern creation for this sub-module, you
create a graybox representation of the sub-module, as shown in Figure 15-2.

Figure 15-1. Full Hierarchical Block Netlist

Input Wrapper Core Output Wrapper
Scan-In Scan-In Scan-In

FF FF Comb FF
Logic

FF
FF FF

Primary Comb
FF Logic
Inputs Comb FF FF
Logic
FF Primary
Outputs
Comb FF
FF Logic
FF Comb
Logic

FF
Comb
Logic FF FF

FF FF FF

Input Wrapper Core Output Wrapper

Scan-Out Scan-Out Scan-Out

Figure 15-2 is a graybox representation of the sub-module shown in Figure 15-1. Note that the
graybox contains only the primary inputs/outputs, wrapper chains, and combinational logic that
exists outside of the wrapper chains (that is, any combinational logic between a primary input or
output and the nearest connected flip-flop).

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 Graybox Overview
Graybox Process Overview

Figure 15-2. Graybox Version of Block Netlist

Input Wrapper Output Wrapper
Scan-In Scan-In

FF FF

FF
FF

Primary FF
Inputs Comb FF
Logic
FF Primary
Outputs
FF

FF Comb
Logic

FF
Comb
Logic FF

FF FF

Input Wrapper Output Wrapper

Scan-Out Scan-Out

Graybox Process Overview

The following is a description of the overall process of generating a graybox netlist. Graybox
functionality is available when the tool is in the analysis system mode and the design is in
external mode. External mode means that the input wrapper chains are used in regular test
modes (both capture and shift), whereas the output wrapper chains are used only in a non-
capture mode (shift, hold or rotate). Wrapper chains inserted by Tessent Scan are configured in
external mode by constraining the output wrapper chain scan_enable signal to active during
both shift and capture phases.
The dofile used for graybox netlist generation does the following:

1. Defines clock pins used in external mode (using the add_clocks command).
2. Constrains test control pins that place the circuit in external mode (using the
add_input_constraints command).
3. Defines wrapper chains (using the add_scan_chains command).

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Graybox Overview
Graybox Process Overview

4. Places the circuit in external mode using a test procedure file. This test procedure file
should do the following:
• Define a test-setup procedure for the external mode to force primary inputs that
enable signal paths to wrapper cells.
• Define a shift and load-unload procedure to force wrapper chain scan enable signals
and toggle the shift clocks for the external mode.
Other types of scan and clock procedures (such as master-observe or shadow-observe)
and non-scan procedures (such as capture) might also be required to ensure that the
circuit operates correctly in external mode.

Note
Test-setup procedures are not allowed if they pulse the clocks to initialize non-scan cells
to constant values in order to sensitize the control signals of external mode. In other
words, the only allowable control signals that place the circuit in external mode are the
primary inputs to the block (core).

5. Identifies graybox logic using the analyze_graybox command. The command also
displays a summary to indicate the combinational and sequential logic gates identified
by the analysis. The tool marks the identified graybox instances by setting their
“in_graybox” attribute. You can also include additional instances in the graybox netlist
(or exclude specific instances from the graybox netlist) by turning on/off this attribute
using the set_attribute_value command.
The graybox analysis performs the identification by tracing backward from all primary
output pins and wrapper chains. However, the scan-out pins of the core chains are
excluded from the backward tracing. Since core chains are not defined with the
add_scan_chains command, you accomplish this by setting the ignore_for_graybox
attribute of the scan-out pins using the set_attribute_value command.
6. The “write_design -graybox” command writes out all the instances marked with the
in_graybox attribute. The tool uniquifies all modules that are included in the graybox
netlist (except the top module). The interface (port declarations) of a uniquified module
is preserved. The uniquification is required because the partial inclusion of the logic
inside a module into the graybox netlist could cause conflicts between the different
instances of the module, as these instances could be interacting differently with the
wrapper chains.

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Graybox Process Overview

Example dofile for Creating a Graybox Netlist

The following dofile example shows how to create a graybox netlist:
# Define clock pins used for external mode
add_clocks 0 NX2
add_clocks 0 NX1

# Set up for external mode

# Hold output wrapper chain scan enable active
add_input_constraints sen_out -C1

# Define wrapper chains

add_scan_groups grp1 external_mode.testproc
add_scan_chains wrapper_chain1 grp1 scan_in1 scan_out1
add_scan_chains wrapper_chain2 grp1 scan_in2 scan_out2

# Ignore core chains scan_out pins for graybox analysis to exclude the
# logic intended for internal test mode
set_attribute_value scan_out3 –name ignore_for_graybox –value true
set_attribute_value scan_out4 –name ignore_for_graybox –value true

set_system_mode analysis

# Identify graybox logic

analyze_graybox -collect_reporting_data
report_graybox_statistics -top 10

# NOTE: At this point, you can use the set_attribute_value command with
# the in_graybox attribute to include/exclude specific instances into/from
# graybox netlist.

# Write graybox netlist

write_design –graybox –output_file graybox.v -replace

Generating Graybox Netlist for EDT Logic Inserted

Blocks
A graybox netlist can also be generated for a block whose scan chains are driven by EDT logic.
Below is an example dofile to generate a graybox netlist for a block whose wrapper chain scan
I/O ports are accessed directly through wrapper-extest ports, bypassing the EDT logic.
These ports are inserted to the block by Tessent Scan when you insert the wrapper chains using
the “set_wrapper_chains -wrapper_extest_ports” command. The wrapper-extest ports are
activated by constraining the global signal wrapper_extest to a logic 1. The EDT logic and
subsequently the core logic are excluded from graybox netlist by marking the EDT channel
outputs with the ignore_for_graybox attribute before performing the graybox analysis.

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Graybox Overview
Graybox Process Overview

# Define clocks
add_clocks 0 clk

# Set up for external mode

# Enable access to wrapper chain extest ports and
# hold output wrapper chain scan enable active
add_input_constraints wrapper_extest -C1
add_input_constraints scan_en_out -C1

# Define wrapper chains

add_scan_groups grp1 cpu_block1_extest.testproc
add_scan_chains chain1 grp1 scan_in1 scan_out2
add_scan_chains chain2 grp1 scan_in3 scan_out4
add_scan_chains chain3 grp1 scan_in5 scan_out6
add_scan_chains chain4 grp1 scan_in7 scan_out8

# Ignore all EDT channel outputs to exclude EDT logic and core chains
# from the graybox netlist.
set_attribute_value edt_channels_out1 -name ignore_for_graybox -value true
set_attribute_value edt_channels_out2 -name ignore_for_graybox -value true
set_attribute_value edt_channels_out3 -name ignore_for_graybox -value true
set_attribute_value edt_channels_out4 -name ignore_for_graybox -value true

# DRC
set_system_mode analysis

# Identify graybox logic

analyze_graybox

# Write graybox netlist

write_design -graybox -output_file graybox.v

In the following example, the block does not have dedicated wrapper-extest ports and therefore
scan I/O ports are accessed only through EDT logic during extest. The setup of the EDT logic is
skipped to simplify the DRC process as there will be no pattern generation in the same tool run.
The EDT logic is inserted such a way that an exclusive subset of the channel I/O pins are used
only for wrapper chains. This allows excluding the core logic from graybox netlist by marking
the core chain EDT channel outputs with ignore_for_graybox attribute. Graybox analysis can
identify the EDT logic that is sensitized for extest if the depth of sequential pipeline stages on
channel outputs allocated for wrapper chains is less than 2. However, in this example, the EDT
block is specified as a preserve instance to include it in the graybox netlist entirely.

# Define clocks
add_clocks 0 clk

# No EDT setup
set_edt_options off

# Set up for external mode

add_pin_constraints seno C1

# Define wrapper chains

add_scan_groups grp1 setup.testproc
add_scan_chains -internal chain1 grp1 \
/edt_block_i/edt_scan_in[0]/edt_block_i/edt_scan_out[0]

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 Graybox Overview
Graybox Process Overview

add_scan_chains -internal chain2 grp1 \

/edt_block_i/edt_scan_in[1] /edt_block_i/edt_scan_out[1]

# Ignore EDT channel outputs that access core logic in graybox analysis
set_attribute_value edt_channels_out3 -name ignore_for_graybox -value true
set_attribute_value edt_channels_out4 -name ignore_for_graybox -value true

# DRC
set_system_mode analysis

# Identify graybox logic

analyze_graybox -preserve_instances edt_block_i

# Write graybox netlist

write_design -graybox -output_file graybox.v

The EDT logic can also be included in the graybox netlist without using the -preserve_instances
switch explicitly. EDT Finder can identify the parts of the EDT logic that are sensitized for
extest and automatically adds them as preserve instances for graybox analysis. This also allows
any sequential pipeline stages on channel I/O pins to be included in the graybox netlist. The
following example dofile shows a typical setup to utilize the EDT finder in graybox analysis.
The EDT channel pins allocated for core chains are constrained/masked to ignore for EDT
Finder. However, this is usually effective when separate EDT blocks are inserted for wrapper
and core chains.

# Define clocks
add_clocks 0 clk

# Set up for external mode

add_pin_constraints seno C1

# Set up EDT Finder

# Ignore EDT channels that are used for core logic
add_clocks 0 edt_clock
add_pin_constraints edt_clock C0
set_edt_finder on
add_input_constraints edt_channels_in2 -C0
add_output_masks edt_channels_out2

# Define EDT block and all chains

add_edt_blocks edt_block
set_edt_options -channels 2 -longest_chain_range 4 28
set_edt_pins input_channel 1 edt_channels_in1 -pipeline_stages 4
set_edt_pins input_channel 2 edt_channels_in2 -pipeline_stages 4
set_edt_pins output_channel 1 edt_channels_out1 -pipeline_stages 3
set_edt_pins output_channel 2 edt_channels_out2 -pipeline_stages 3
add_scan_groups grp1 setup.testproc
add_scan_chains -internal chain1 grp1 /edt_block_i/edt_scan_in[0] \
/edt_block_i/edt_scan_out[0]
add_scan_chains -internal chain2 grp1 /edt_block_i/edt_scan_in[1] \
/edt_block_i/edt_scan_out[1]
add_scan_chains -internal chain3 grp1 /edt_block_i/edt_scan_in[3] \
/edt_block_i/edt_scan_out[3]
add_scan_chains -internal chain4 grp1 /edt_block_i/edt_scan_in[4] \
/edt_block_i/edt_scan_out[4]

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Graybox Process Overview

# Ignore EDT channel outputs that access core logic in graybox analysis
set_attribute_value edt_channels_out3 -name ignore_for_graybox -value true
set_attribute_value edt_channels_out4 -name ignore_for_graybox -value true

# DRC
set_system_mode analysis

# Identify graybox logic

analyze_graybox

# Write graybox netlist

write_design -graybox -output_file graybox.v

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 Appendix A
Clock Gaters

The use of clock gaters to reduce power consumption is becoming a widely adopted design
practice. Although effective for reducing power, clock gaters create new challenges for ATPG
tools because of the additional complication introduced in clock paths. Among the challenges,
the most frequently encountered in the tool is the C1 DRC violation. The C1 rule is fairly strict,
requiring clock ports of scan cells to be at their off states when all clock primary inputs (PIs) are
at their off states. Not all designs abide by this rule when there are clock gaters in clock paths.

Basic Clock Gater Cell

There is one common type of clock gater cell.
Figure A-1 shows the structure of the cell.

Figure A-1. Basic Clock Gater Cell

scan_en
en D Q

gclk
clk

The cell (inside the dashed box) has three inputs: scan_en, en, and clk. The output of the cell is
the gated clock, gclk. The scan_en input is for test mode and is usually driven by a scan enable
signal to allow shifting. The en input is for functional mode and is usually driven by internal
control logic. Sometimes test_en is used to drive the scan_en input to avoid any trouble caused
by the clock gater. However, this ends up keeping the clock gater always on and results in loss
of coverage in the en fanin cone. The latch in the clock gater eliminates potential glitches.
Depending on the embedding, there could be inversions of the clk signal both in front of clk and
after gclk, which in the figure are shown as inverters on the dashed lines representing the clk
input and gclk output.

Two Types of Embedding

The key factor affecting whether the tool produces DRC violations is whether the clk signal gets
inverted in front of the clk input to the cell.

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Clock Gaters
Two Types of Embedding

To better understand the impact of such an inversion, it is helpful to understand two common
ways designers embed the basic clock gater cell (see Figure A-1) in a design. Figure A-2 shows
the two possibilities, referred to as Type-A and Type-B.

Figure A-2. Two Types of Embedding for the Basic Clock Gater
Type-A
D Q

gclk1

PI /clock
Type-B
D Q

gclk2

In the figure, assume the PI /clock drives two clock gaters and that the off state of /clock is 0.
The behavior of the two types of embeddings is as follows:

• Type-A — When PI /clock is at its off state, the latch in the clock gater is transparent,
and the AND gate in the clock gater gets a controlling value at its input. As a result, the
output gclk1 is controlled to a deterministic off state.
• Type-B — When /clock is at its off state, the latch in the clock gater is not transparent,
and the AND gate in the clock gater gets a non-controlling value at its input. As a result,
the output gclk2 is not controlled to a deterministic off state. Its off state value will
depend on the output of the latch.

Note
The preceding classification has nothing to do with whether there is inversion after gclk.
It only depends on whether the off state of /clock can both make the latch transparent and
control the AND gate.

Ideal Case (Type-A)

The tool fully supports the Type-A embedding for the following reasons:
• Since the latch is transparent, when scan_en is asserted at the beginning of load_unload,
the clock gater is immediately turned on. Therefore, subsequent shifting is reliable and
there will be no scan chain tracing DRC violations.
• Since gclk is controlled to a deterministic off state, there are no C1 DRC violations for
downstream scan cells driven by it.

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 Clock Gaters
Two Types of Embedding

• The tool can pick up full fault coverage in the fanin cone logic of the en and scan_en
inputs of the clock gater. This coverage is often too significant to be sacrificed.
Type-A embedding is the preferred design practice and is strongly recommended.

Tip: Type-A embedding does not necessarily mean the downstream scan cells have to be
either all leading edge (LE) flip-flops or all trailing edge (TE) flip-flops. A designer can
have both of them by inserting inversions after gclk.

Potential DRC Violator (Type-B)

Type-B embeddings have the following undesirable characteristics:
• When PI /clock is at its off state, the latch in the clock gater is not transparent. This
means a change on its D input at the beginning of a cycle will not immediately be
reflected at its Q output. In other words, the action of turning the clock gater on or off
lags its instruction.
• The clock off state of gclk is not deterministic. It depends on what value the latch in the
clock gater captures in the previous cycle. Therefore, the tool will issue a C1 DRC
violation.
The kinds of problems that can arise due to these undesirable characteristics are described in the
following sections.

Avoiding Type-B Scan Chain Tracing Failures (T3

Violations)
If the scan flip-flops downstream from the Type-B clock gater are LE triggered, the result will
be scan chain tracing failures.
This is because the first shift edge in the load_unload is missed, as illustrated in Figure A-3 and
Figure A-4. Figure A-3 shows an example circuit where a Type-B clock gater drives both TE
and LE scan flip-flops.

Note
Assuming the off state of the PI /clock is 0, an inverter is needed after gclk to make the
downstream flip-flop LE. The inverted gclk is indicated as gclk_b.

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Clock Gaters
Two Types of Embedding

Figure A-3. Type-B Clock Gater Causes Tracing Failure

TE scan flip-flop
scan_en
en D Q

LE scan flip-flop
gclk

PI /clock
gclk_b

Figure A-4. Sample EDT Test Procedure Waveforms

load_unload capture load_unload

/clock

scan_en

Q_bar

gclk_b

First LE shift edge is missed First TE shift edge survives

Figure A-4 shows the timing diagram of several signals during capture and the beginning of
load_unload. Suppose the latch in the Type-B clock gater captures a 1 in the last capture cycle
(shown as Q_bar going low in the capture window). You can see that the first leading edge is
suppressed because the latch holds its state after capture and into load_unload. Although
scan_en is high at the beginning of load_unload, the output of the latch will not change until the
first shift clock arrives. This lag between the “instruction” and the “action” causes the
downstream LE scan flip-flops to miss their first shift edge. However, TE-triggered scan flip-
flops are still able to trigger on the first trailing edge.

Tip: To work around this problem, you can use test_en instead of scan_en to feed the
clock gater. But be aware that this workaround loses fault coverage in the fanin cone of
the clock gater’s en input.

Preventing Scan Cell Data Captures When Clocks are Off

T3 violations rarely occur in designs with Type-B clock gaters that drive LE scan flip-flops,
typically because the downstream scan flip-flops in those designs are all made TE by synthesis

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 Clock Gaters
Cascaded Clock Gaters

tools. The tools handle Type-B clock gaters that drive TE scan flip-flops just fine, and change
any downstream latches, LE nonscan flip-flops, RAMs, and POs to TIEX. This prevents C1
DRC violations, coverage loss in the en cone of the clock gater, and potential simulation
mismatches when you verify patterns in a timing based simulator.

Accounting for Type-B Clock Gaters Driving LE Flops

When you have a Type-B clock gater driving LE flops in the chain, you must do the following
in sequence:
1. Initialize the clock gaters in one cycle of test_setup by driving scan_enable high and
pulsing the clocks.
2. Issue the set_stability_check command using the following arguments:
set_stability_check on -sim_static_atpg_constraints on

3. Add ATPG constraints of “1” on the functional enable of the Type-B clock gaters using
the add_atpg_constraints command using the following arguments:
add_atpg_constraints 1 functional_enable_pin -static

Cascaded Clock Gaters

One level of clock gating is sometimes not enough to implement a complex power controlling
scheme.
Figure A-5 shows a two-level clock gating scheme using the clock gater cell. The dashed lines
in the figure indicate places where inversions may exist. PI /clock is first gated by a level-1
gater and then the output gclk1 is further gated by a level-2 gater, generating the final clock
gclk2.

Figure A-5. Two-level Clock Gating

Level-2
Level-1
scan_en
scan_en en D Q
en D Q

gclk2
gclk1
PI /clock

scan cell
scan cell

Understanding a Level-2 Clock Gater

In a two-level clock gating scheme, if the level-1clock gater is of Type-A, then the clock off
value of gclk1, being deterministic, has no problem propagating to the level-2 clock gater.

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Clock Gaters
Cascaded Clock Gaters

The latter can be understood based on the definition in “Basic Clock Gater Cell” on page 435.

However, when the level-1 gater is of Type-B, the tool cannot apply the basic definition to the
level-2 gater directly, since the clock off value of gclk1 is undetermined. In this case, the type
for the level-2 clock gater is defined by assuming the level-1 clock gater has been turned on.
Therefore, even if the level-1 clock gater is of Type-B, the tool can still classify the level-2
clock gater.

Example Combinations of Cascaded Clock Gaters

Following are the four possible combinations of two-level cascading clock gaters and the
support the tool provides for them:
• Type-A Level-1 + Type-A Level-2 — This is the preferred combination, is fully
supported by the tool, and is strongly recommended.
• Type-A Level-1 + Type-B Level-2 — This can be reduced to a single level type-B case.
All earlier discussion of type-B clock gaters applies. See “Potential DRC Violator
(Type-B)” on page 437 for more information.
• Type-B Level-1 + Type-A Level-2 — Avoid this combination. Scan chain tracing will
be broken even if the downstream scan flip-flops of the level-2 gater are TE. The
problem occurs when both level-1 and level-2 clock gaters capture a 1 in the last cycle of
the capture window (meaning both clock gaters are turned off). Then the downstream
scan flip-flops will miss both the first LE shift edge and the first TE shift edge in the
subsequent load_unload. This is a design problem, not a tool limitation.
• Type-B Level-1 + Type-B Level-2 — Supported only if all the downstream scan flip-
flops are TE.

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 Clock Gaters
Summary

Summary
The following table summarizes the support that the tool provides for each clock gater
configuration.

Table A-1. Clock Gater Summary

Gater Configuration LE Scan Flip-flops TE Flip-flops Latches, RAM,
Downstream Downstream ROM, PO
Downstream
Type-A Supported, strongly Supported, strongly Supported.
recommended. recommended.
Type-B T3 DRC violations. A Supported. Changed to TIEX.
design issue. Not
supported.
Type-A Level-1 + Supported. Supported. Supported.
Type-A Level-2
Type-A Level-1 + Same as Type-B Same as Type-B Same as Type-B
Type-B Level-2 alone. alone. alone.
Type-B Level-1 + T3 DRC violations. T3 DRC violations. Changed to TIEX.
Type-A Level-2 Not supported. Not supported.
Type-B Level-1 + T3 DRC violations. Supported. Changed to TIEX.
Type-B Level-2 Not supported.

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Summary

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 Appendix B
Debugging State Stability

State stability refers to state elements that are stable, hold their values across shift cycles and
patterns, and their values can be used for DRC (for example, scan chain tracing). Debugging
state stability is useful when you set up a state (for example, in a TAP controller register, in the
test_setup procedure) that you use for sensitizing the shift path. Unless the state value is
preserved for all shift cycles and all patterns, then scan chain tracing can fail. If you set up the
correct value in the test_setup procedure but the value is changed due to the application of shift
or the capture cycle, the value cannot be depended on. For such cases, you can debug state
stability by identifying what caused a stable state element to unexpectedly change states.
Note
The information in this appendix uses the set_stability_check command set to On (the
default for this command), except as noted in Example 3 and Example 9.

Displaying the State Stability Data

You have several options for displaying state stability data.
Using the set_gate_report command with the Drc_pattern option, you can display simulation
data for different procedures using the report_gates command or DFTVisualizer. The following
examples show how to use the set_gate_report command:

set_gate_report drc_pattern test_setup

set_gate_report drc_pattern load_unload
set_gate_report drc_pattern shift

For “drc_pattern load_unload” and “drc_pattern shift”, the superimposed values for all
applications of the procedure are reported. This means a pin that is 1 for only the first
application of shift, but 0 or X for the second application of shift shows up as X.

When you set the Drc_pattern option of the set_gate_report command to State_stability as
shown in the following example:

set_gate_report drc_pattern state_stability

the state stability report also includes the load_unload and shift data for the first application of
these procedures.

When you debug state stability, you normally compare the state stability values (the values
during the first application of the procedures) with the superimposed (stable) values from
“drc_pattern load_unload” and “drc_pattern shift.”

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State Stability Data Format

State stability data is only available after DRC and only if there is a test_setup procedure. The
format of the state stability data displayed for a pin is shown in the following example:
(ts) (ld) (shift) (ld) (shift) (cell_con) (cap) (stbl)
// CLK I ( 0) ( 0) (010~0) (00) (010) ( 0 ) (0X0) ( 0)
The first row lists the column labels in parentheses. The second row displays the pin name and
five or more groups of data in parentheses. Each group has one or more bits of data
corresponding to the number of events in each group.

Following is a description of the data columns in the state stability report:

• ts — Last event in the test_setup procedure. This is always one bit.

• ld — Any event in the load_unload procedure that is either before and/or after the apply
shift statement. When load_unload is simulated, all primary input pins that are not
explicitly forced in the load_unload procedure or constrained with an
add_input_constraints command are set to X. Constrained pins are forced to their
constrained values at the end of test_setup prior to load_unload. This is because even if
your test_setup procedure gives those pins different values coming out of test_setup, in
the final patterns saved, an event is added to test_setup to force those pins to their
constrained values. (You can see that event in the test_setup procedure if you issue
write_procfile after completing DRC.) Consequently, for the first pattern, a pin has its
constrained value going into load_unload. For all subsequent patterns, load_unload
follows the capture cycle, during which the tool would have enforced the constrained
value. Therefore, for all patterns, a constrained pin is forced to its constrained value
going into load_unload and hence this value can be used for stability analysis. This is
also true for the state stability analysis where the first application of load_unload is
simulated.
The number of bits in this group depends on the number of events in the procedure prior
to the apply shift statement. If there are no events prior to the apply shift statement, this
group still exists (with one bit).
• shift — Apply shift statement (main shift or independent shift). If the load_unload
procedure has multiple apply shift statements, then multiple (shift) groups are displayed.
If this is also the main shift application (that is, the number of shifts applied is greater
than one), the group includes a tilde (~). The values before the tilde correspond to the
very first application of the shift procedure, with the number of bits corresponding to the
number of primary input events in the shift procedure. The group without a tilde (~) is
the independent shift.
The last bit (after the tilde) corresponds to the stable state after application of the last
shift in the main shift statement. By default, the precise number of shifts is not
simulated. The stable state shown corresponds to an infinite number of shift cycles; that

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is, in some cases, when for instance the depth of non-scan circuitry is deeper than the
scan chain length, sequential circuitry that should be 1 or 0 after the first load_unload
may show up as X. For more information, see the description of the “set_stability_check
all_shift” command in “Example 4 — Single Post Shift” on page 455.
After the third group, there could be additional groups if there are the following:
o Multiple apply shift statements
o Events in the load_unload procedure after the apply shift statement
• shdw_con — Shadow_control procedure. This column is not shown in the examples.
Look at the test procedure file to determine the meaning of this group.
• cell_con — Cell constraints. If there are cell constraints, an extra simulation event is
added to show the value. This is always one bit.
• cap — Capture procedure. This is the simulation of the first capture cycle. Notice that
this is not the simulation of pattern 0 or any specific pattern. Therefore, normally the
capture clock going to 0X0 (or 1X1 for active low clocks) is displayed, indicating that
the clock may or may not pulse for any given pattern. If you issue the
“set_capture_clock -Atpg” command to force the tool to use a specific capture clock
exactly once for every pattern, the capture cycle simulation is less pessimistic and “010”
or “101” is reported.
• stbl — Final stable values after several iterations of simulating load_unload and capture
procedures. If the value of a gate is always the same at the end of the test_setup and
capture procedures, then its stable value is the same; otherwise, the stable value is X.

Note
The skew_load, master_observe, or shadow_observe procedures are not simulated as part
of state stability analysis, so their simulation data is not displayed in a state stability
report. To view the simulation data for any of these procedures, use the
set_gate_report Drc_pattern command with the specific procedure_name of interest.

State Stability Examples

This section contains examples of state stability reporting that show different behaviors in state
stability analysis.
The examples are based on the design shown in Figure B-1. The design has the following three
clocks:

• clk1 — The only scan clock

• clk2 — Clocks a particular non-scan flip-flop
• reset — Resets four non-scan flip-flops connected as a register

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The design has five pins (A, B, C, D, and E) that are exercised in the procedures to specifically
show state stability analysis. The circuit has the following characteristics:

• Pin D has a C0 pin constraint. There are no other constrained pins.

• Non-scan flip-flop ff00 is clocked by clk1 and driven by pin A. The flip-flop is
initialized in test_setup, but loses state when it is pulsed after test_setup.
• Non-scan flip-flop ff10 is clocked by clk1 and driven by pin D. Notice that this one gets
initialized and “sticks” (is converted to TIE0) due to the pin constraint.
• Non-scan flip-flop ff20 is clocked by clk2 and driven by pin A.
• Non-scan flip-flop ff32 is the fourth flip-flop in a serial shift register. The first flip-flop
in the register is ff30 and driven by pin A. The flip-flops are all clocked by clk1 and
reset by the reset signal. ff32 is reset in test_setup, maintains state through the first
application of shift, but then loses stability.
• There is one scan chain with three scan cells: sff1, sff2, and sff3.
With the exception of Example 3, the only difference between the eleven examples is the test
procedure file. The timeplate and procedures from this file that were used for the first (basic)
example are shown following the design. The procedures used in subsequent examples differ
from the basic example as follows:

Example 1 — Basic Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 449

Example 2 — Multiple Cycles in Load_unload Prior to Shift . . . . . . . . . . . . . . . . . . . . . . 453
Example 3 — Drc_pattern Reporting for Pulse Generators . . . . . . . . . . . . . . . . . . . . . . . . 454
Example 4 — Single Post Shift . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 455
Example 5 — Single Post Shift with Cycles Between Main and Post Shift . . . . . . . . . . . 456
Example 6 — Cycles After Apply Shift Statement in Load_unload . . . . . . . . . . . . . . . . . 458
Example 7 — No Statements in Load_unload Prior to Apply Shift. . . . . . . . . . . . . . . . . . 459
Example 8 — Basic with Specified Capture Clock . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 459
Example 9 — Setting Stability Check to Off and All_shift . . . . . . . . . . . . . . . . . . . . . . . . 460
Example 10 — Pin Constraints, Test_setup, and State Stability . . . . . . . . . . . . . . . . . . . . 461
Example 11 — Single Pre Shift . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 463
Example 12 — Basic with Enhanced Stability Check for NCPs . . . . . . . . . . . . . . . . . . . . 464

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Figure B-1. Design Used in State Stability Examples

timeplate gen_tp1 =
force_pi 0;
measure_po 10;
pulse clk1 20 10;
pulse clk2 20 10;
pulse reset 20 10;
period 40;
end;

procedure capture =
timeplate gen_tp1;
cycle =
force_pi;
measure_po;
pulse_capture_clock;
end;
end;

procedure shift =
scan_group grp1;
timeplate gen_tp1;

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cycle =
force_sci;
measure_sco;
pulse clk1;
force C 0;
end;
end;

procedure test_setup =
scan_group grp1;
timeplate gen_tp1;
// First cycle, one PI event (force)
cycle =
force clk1 0;
force clk2 0;
force reset 0;
force A 0;
force B 0;
force C 0;
force D 0;
force E 0;
end;
// Second cycle, two PI events (pulse on, pulse off)
cycle =
pulse reset;
pulse clk2;
end;
// Third cycle, three PI events (force, pulse on, and pulse off)
cycle =
force A 1;
pulse clk1;
end;
end;

procedure load_unload =
scan_group grp1;
timeplate gen_tp1;
// First cycle, one PI event (force)
cycle =
force clk1 0;
force clk2 0;
force reset 0;
force scan_en 1;
force B 1;
force C 1;
end;
apply shift 3;
end;

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Example 1 — Basic Example

In the basic example, the tool is invoked on the design and the following commands are entered:

add_clocks 0 clk1 clk2 reset

add_scan_groups grp1 scan1.testproc
add_scan_chains c0 grp1 scan_in1 scan_out1
add_input_constraints D -c0
set_gate_report drc_pattern state_stability
set_system_mode analysis
report_gates A B C D E
...
// (ts)(ld)(shift)(cap)(stbl)
// A O ( 1)( 1)(111~1)(XXX)( X )
// B O ( 0)( 1)(111~1)(XXX)( X )
// C O ( 0)( 1)(000~0)(XXX)( X )
// D O ( 0)( 0)(000~0)(000)( 0 )
// E O ( 0)( 0)(000~0)(XXX)( X )

The results of this operation are:

• Pins A through E are explicitly forced in test_setup.

• Pins A, B, C, and E are forced in load_unload.
• Pin D is constrained (but not forced in load_unload) and is the only one of the pins with
an explicit value during capture.
A typical initialization problem is illustrated in ff00. Figure B-2 shows the relevant circuitry
and statements in the test_setup procedure that seemingly initialize this flip-flop.

Figure B-2. Typical Initialization Problem

procedure test_setup =
... A
// Third cycle...
cycle =
force A 1; PI ff00
pulse clk1;
end; D Q
...
clk1 CLK QB
PI dff

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In this case, you might expect the output to always be 1 after initialization because the third
cycle of the test_setup procedure forced a 1 on input A and pulsed clk1. For comparison
purposes, following is the reported state_stability data together with the data reported for
load_unload and shift. Notice especially the Q output:

state_stability load_unload shift

// /ff00 dff
// (ts)(ld)(shift)(cap)(stbl) (ld)(shift)
// CLK I ( 0)( 0)(010~0)(0X0)( 0 ) ( 0)( 010) (010)
// D I ( 1)( X)(XXX~X)(XXX)( X ) ( X)( XXX) (XXX)
// Q O ( 1)( 1)(1XX~X)(XXX)( X ) ( 1)( XXX) (XXX)
// QB O ( 0)( 0)(0XX~X)(XXX)( X ) ( 0)( XXX) (XXX)

You can see from the state stability display that, after test_setup, the output of Q is set to 1. In
the first application of load_unload it is still 1, but it goes to X during the first shift. Compare
this to what is shown for “drc_pattern load_unload” and “drc_pattern shift”.

A stable initialization value can be better achieved by doing for ff00 something similar to what
occurs for ff10, where the D input is connected to the constrained D pin:

state_stability load_unload shift

// /ff00 dff
// (ts)(ld)(shift)(cap)(stbl) (ld)(shift)
// CLK I ( 0)( 0)(010~0)(0X0)( 0 ) ( 0)( 010) (010)
// D I ( 1)( X)(000~0)(000)( X ) ( X)( 000) (000)
// Q O ( 1)( 1)(000~0)(000)( X ) ( 0)( 000) (000)
// QB O ( 0)( 0)(111~1)(111)( X ) ( 1)( 111) (111)

Another interesting observation can be made for ff32. This flip-flop is at the end of a 4-bit shift
register where all the flip-flops are reset during test_setup as shown in Figure B-3.

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Figure B-3. Three-bit Shift Register (Excerpted from Figure D-1)

reset
PI ff30 ff31 ff31b ff32
R R R R
Q Q Q Q
A D D D D
QB QB QB QB
PI CLK CLK CLK CLK
dffr dffr dffr dffr
clk1
PI
procedure test_setup =
...
// Second cycle...
cycle =
pulse reset;
...
end;
...

Notice how Q in this case is stable for the first application of load_unload and shift, but the
stable state after the last shift (after ~) is X. This is due to an optimization the tool does by
default for the state_stability check. (Compare this output to example Example 9 — Setting
Stability Check to Off and All_shift.)

state_stability load_unload shift

// /ff32 dffr
// (ts)(ld)(shift)(cap)(stbl) (ld)(shift)
// R I ( 0)( 0)(000~0)(0X0)( 0 ) ( 0)( 000) (000)
// CLK I ( 0)( 0)(010~0)(0X0)( 0 ) ( 0)( 010) (010)
// D I ( 0)( 0)(000~0)(XXX)( X ) ( 0)( 000) (XXX)
// Q O ( 0)( 0)(000~1)(XXX)( X ) ( 0)( 000) (XXX)
// QB O ( 1)( 1)(111~1)(XXX)( X ) ( 1)( 111) (XXX)

Non-scan flip-flop ff20 is clocked by clk2, which is not a shift clock. This flip-flop is also
initialized during test_setup as shown in Figure B-4.

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Figure B-4. Initialization with a Non-Shift Clock

procedure test_setup =
...
// First cycle... A
cycle =
...; PI ff20
force A 0;
... D Q
end;
// Second cycle... clk2 CLK QB
cycle =
... PI dff
pulse clk2;
end;
...

The Q output is disturbed during capture, not during shift, because this element is not exercised
during shift:

state_stability load_unload shift

// /ff20 dff
// (ts)(ld)(shift)(cap)(stbl) (ld)(shift)
// CLK I ( 0)( 0)(000~0)(0X0)( 0 ) ( 0)( 000) (000)
// D I ( 1)( X)(XXX~X)(XXX)( X ) ( X)( XXX) (XXX)
// Q O ( 0)( 0)(000~0)(0XX)( X ) ( 0)( 000) (XXX)
// QB O ( 1)( 1)(111~1)(1XX)( X ) ( 1)( 111) (XXX)

Notice that the load_unload and shift data for ff32 and ff20 is almost identical (except for the
clock data), but that the state_stability data enables you to see that they become unstable in very
different ways.

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Example 2 — Multiple Cycles in Load_unload Prior to Shift

The main difference between this example and the basic example is that in the load_unload
procedure there are multiple cycles prior to the apply shift statement (statements that are new in
this example are highlighted in bold):

procedure load_unload =
scan_group grp1 ;
timeplate gen_tp1 ;
// First cycle, one PI event (force)
cycle =
force clk1 0 ;
force clk2 0 ;
force reset 0 ;
force scan_en 1 ;
force B 1;
force C 1;
force E 1;
end ;
// Second cycle, three PI events (force, pulse on, pulse off)
cycle =
force E 0;
pulse clk2;
end ;
// Third cycle, two PI events (pulse on, pulse off)
cycle =
pulse clk2;
end ;
apply shift 3;
end;

As a result, multiple events are displayed in the second group of state stability data. Notice there
are now three cycles. The following gate report excerpts show six bits of data (in bold),
corresponding to the total number of events. The first bit is from the first cycle (one event), the
next three bits are from the second cycle (three events), and the last two bits are from the third
cycle, which has two events.

// /E primary_input
// (ts)( ld )(shift)(cap)(stbl)
// E O ( 0)(100000)(000~0)(XXX)( X)
// /A primary_input
// (ts)( ld )(shift)(cap)(stbl)
// A O ( 1)(XXXXXX)(XXX~X)(XXX)( X)
// /clk2 primary_input
// (ts)( ld )(shift)(cap)(stbl)
// clk2 O ( 0)(001010)(000~0)(0X0)( 0)

// /ff20 dff
// (ts)( ld )(shift)(cap)(stbl)
// CLK I ( 0)(001010)(000~0)(0X0)( 0)
// D I ( 1)(XXXXXX)(XXX~X)(XXX)( X)
// Q O ( 0)(00XXXX)(XXX~X)(XXX)( X)
// QB O ( 1)(11XXXX)(XXX~X)(XXX)( X)
Notice how A goes to X for the load_unload simulation. This is because it is not explicitly
forced in the load_unload procedure (or constrained with an add_input_constraints command).

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Example 3 — Drc_pattern Reporting for Pulse Generators

If a circuit contains a pulse generator (PG) with user-defined timing, the tool performs
additional simulation steps for the PG’s output changes. When a rising edge event occurs at its
input, a PG outputs a 1 after a certain delay, which the tool simulates as an additional event.
After another delay, the PG’s output signal returns to 0, which is also simulated as a separate
event. Both output events are added to reports within a pair of brackets ([ ]) following the input
event.

Suppose the non-scan flip-flop ff20 of the preceding example is clocked by a PG as shown in
Figure B-5. Excerpts below the figure show how the PG events would appear in gate reports.

Figure B-5. Clocking ff20 with a Pulse Generator

A
PI ff20

pg1 D Q
clk2 clk out CLK QB
pulse_gen
PI dff
// /clk2 primary_input
// (ts)( ld )(shift)( cap )(stbl)
// clk2 O ( 0)(001[11]01[11]0)(000~0)(0X[X]0)( 0) /pg1/clk

// /pg1 pulse_gen
// (ts)( ld )(shift)( cap )(stbl)
// clk I ( 0)(001[11]01[11]0)(000~0)(0X[X]0)( 0) /clk2
// out O ( 0)(000[10]00[10]0)(000~0)(00[X]0)( 0) /ff20/CLK

// /ff20 dff
// (ts)( ld )(shift)( cap )(stbl)
// CLK I ( 0)(000[10]00[10]0)(000~0)(00[X]0)( 0) /pg1/out
// D I ( 1)(XXX[XX]XX[XX]X)(XXX~X)(XX[X]X)( X) /A
// Q O ( 0)(000[XX]XX[XX]X)(XXX~X)(XX[X]X)( X)
// QB O ( 1)(111[XX]XX[XX]X)(XXX~X)(XX[X]X)( X)

The rising edge events on clk2 initiate pg1’s two output pulses (highlighted in bold). Notice the
pulses are not shown simultaneous with the input changes that caused them. This is an
exception to the typical display of output changes simultaneous with such input changes, as
shown for ff20. Notice also how the active clock edge at ff20’s CLK input is one event later
than clk2’s active edge and is seen to be a PG signal due to the brackets.

For an introduction to pulse generators, see “Pulse Generators” on page 109. For detailed
information about the tool’s pulse generator primitive, see “Pulse Generators with User Defined
Timing” in the Tessent Cell Library Manual.

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Example 4 — Single Post Shift

This example has multiple cycles in the load_unload procedure. What is new is the single post
shift (highlighted in bold).

procedure load_unload =
scan_group grp1 ;
timeplate gen_tp1 ;
// First cycle, one PI event (force)
cycle =
force clk1 0 ;
force clk2 0 ;
force reset 0 ;
force scan_en 1 ;
force B 1;
force C 1;
end ;
// Second cycle, three PI events (force, pulse on, pulse off)
cycle =
force E 0;
pulse clk2;
end;
apply shift 2;
apply shift 1;
end;

In this case, the state stability data has an additional group (shown in bold) between the main
shift and the capture cycle. This corresponds to the first application of the post shift:

// /ff32 dffr
// (ts)( ld )(shift)(shift)(cap)(stbl)
// R I ( 0)(0000)(000~0)( 000 )(0X0)( 0) /reset
// CLK I ( 0)(0000)(010~0)( 010 )(0X0)( 0) /clk1
// D I ( 0)(0000)(000~X)( XXX )(XXX)( X) /ff31b/Q
// Q O ( 0)(0000)(000~X)( XXX )(XXX)( X)
// QB O ( 1)(1111)(111~X)( XXX )(XXX)( X)

You can see that ff32 is really stable during the first application of shift. If you use the
set_stability_check All_shift command in this case, the output is slightly different:

set_stability_check all_shift
set_system_mode setup
set_system_mode analysis
report_gates ff32
// /ff32 dffr
// (ts)( ld )(shift)(shift)(cap)(stbl)
// R I ( 0)(0000)(000~0)( 000 )(0X0)( 0) /reset
// CLK I ( 0)(0000)(010~0)( 010 )(0X0)( 0) /clk1
// D I ( 0)(0000)(000~1)( 1XX )(XXX)( X) /ff31b/Q
// Q O ( 0)(0000)(000~0)( 011 )(1XX)( X)
// QB O ( 1)(1111)(111~1)( 100 )(0XX)( X)

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Notice how ff32 is now 0 throughout the main shift application, but is set to 1 during the post
shift. This is due to how A is set to 1 in test_setup and this pulse is clocked through.

Example 5 — Single Post Shift with Cycles Between Main and Post Shift
In this example, a post shift exists but there is an additional cycle (shown in bold font) between
the main shift and the post shift. This causes yet another group of data to be displayed when you
report the state stability data.

procedure load_unload =
scan_group grp1 ;
timeplate gen_tp1 ;
// First cycle, one PI event (force)
cycle =
force clk1 0 ;
force clk2 0 ;
force reset 0 ;
force scan_en 1 ;
force B 1;
force C 1;
end ;
// Second cycle, three PI events (force, pulse on, pulse off)
cycle =
force A 1;
pulse clk2;
end ;
apply shift 3;
// Third cycle, three PI events (force, pulse on, pulse off)
cycle =
force C 1;
force A 0;
pulse clk2;
end ;
apply shift 1;
end;

procedure shift =
scan_group grp1 ;
timeplate gen_tp1 ;
cycle =
force_sci ;
measure_sco ;
pulse clk1 ;
force C 0;
end;
end;

The fourth data group (highlighted in bold) represents the cycle between the two applications of
shift (for the first application of the load_unload procedure). The fifth data group represents the
application of the post shift, and the last (sixth) group represents capture. Notice how pins A and
C vary state due to the values forced on them in the load_unload and shift procedures.

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// /A primary_input
// (ts)( ld )(shift)( ld)(shift)(cap)(stbl)
// A O ( 1)(X111)(111~1)(000)( 000 )(XXX)( X)
// /B primary_input
// (ts)( ld )(shift)( ld)(shift)(cap)(stbl)
// B O ( 0)(1111)(111~1)(111)( 111 )(XXX)( X)
// /C primary_input
// (ts)( ld )(shift)( ld)(shift)(cap)(stbl)
// C O ( 0)(1111)(000~0)(111)( 000 )(XXX)( X)

Notice also that clk2 in this case is pulsed during load_unload only, not in test_setup. Here is the
modified test_setup procedure (with the clk2 pulse removed from the second cycle):

procedure test_setup =
scan_group grp1 ;
timeplate gen_tp1 ;
// First cycle, one event (force)
cycle =
force clk1 0 ;
force clk2 0 ;
force reset 0;
force A 0;
force B 0;
force C 0;
force D 0;
force E 0;
end ;
// Second cycle, two events (pulse on, pulse off)
cycle =
pulse reset ;
end;
// Third cycle, three events (force, pulse on, pulse off)
cycle =
force A 1;
pulse clk1 ;
end;
end;

This results in the following behavior for ff20, which is clocked by clk2:

// /ff20 dff
// (ts)( ld )(shift)( ld)(shift)(cap)(stbl)
// CLK I ( 0)(0010)(000~0)(010)( 000 )(0X0)( 0) /clk2
// D I ( 1)(X111)(111~1)(000)( 000 )(XXX)( X) /A
// Q O ( X)(XX11)(111~1)(100)( 000 )(0XX)( X)
// QB O ( X)(XX00)(000~0)(011)( 111 )(1XX)( X)

Notice how the state of this flip-flop is disturbed during capture.

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Example 6 — Cycles After Apply Shift Statement in Load_unload

This example reuses the load_unload procedure of the basic example (with only one apply shift
statement), but adds three new cycles (shown in bold font) after the application of shift.

procedure load_unload =
scan_group grp1 ;
timeplate gen_tp1 ;
// First cycle, one PI event (force)
cycle =
force clk1 0 ;
force clk2 0 ;
force reset 0 ;
force scan_en 1 ;
force B 1;
force C 1;
end ;
apply shift 3;
// Second cycle, one PI event (force)
cycle =
force C 1;
end ;
// Third cycle, three PI events (force, pulse on, pulse off)
cycle =
force C 0;
pulse clk2;
end ;
// Fourth cycle, one PI event (force)
cycle =
force C 1;
end ;
end;
procedure shift =
scan_group grp1 ;
timeplate gen_tp1 ;
cycle =
force_sci ;
measure_sco ;
pulse clk1 ;
force C 0;
end;
end;

In this case, the second data group in the state stability report represents the one event in the
cycle before the apply shift statement. The fourth data group represents the events in the three
cycles after the apply shift statement (but still for the first application of load_unload). The first
bit is the one event in the second cycle, the next three bits represent the third cycle, and the last
bit represents the fourth cycle:

// /C primary_input
// (ts)(ld)(shift)( ld )(cap)(stbl)
// C O ( 0)( 1)(000~0)(10001)(XXX)( X)
// /clk2 primary_input
// (ts)(ld)(shift)( ld )(cap)(stbl)
// clk2 O ( 0)( 0)(000~0)(00100)(0X0) ( 0) /ff20/CLK

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Example 7 — No Statements in Load_unload Prior to Apply Shift

In this example, there are no statements in the load_unload procedure prior to the apply shift
statement and scan_en is forced in the shift procedure instead of in a separate cycle in
load_unload. Other procedures are as in the basic example. Here are the modified load_unload
and shift procedures:

procedure load_unload =
scan_group grp1 ;
timeplate gen_tp1 ;
apply shift 3;
end;

procedure shift =
scan_group grp1 ;
timeplate gen_tp1 ;
cycle =
force_scan_en 1 ;
force sci ;
measure_sco ;
pulse clk1 ;
force C 0;
end;
end;

In this case, the report still shows an event for load_unload in the second data group. This makes
it easier to see differences between the end of test_setup and the entry into the first load_unload.
Such differences can occur because during load_unload the tool sets to X any primary input pins
that are not constrained with an add_input_constraints command or explicitly forced in the
load_unload procedure. This is the case for pin A in this example:

// /A primary_input
// (ts)(ld)(shift)(cap)(stbl)
// A O ( 1)( X)(XXX~X)(XXX)( X) /ff30/D /ff20/D /ff00/D
// /ff20 dff
// (ts)(ld)(shift)(cap)(stbl)
// CLK I ( 0)( 0)(000~0)(0X0)( 0) /clk2
// D I ( 1)( X)(XXX~X)(XXX)( X) /A
// Q O ( 0)( 0)(000~0)(0XX)( X)
// QB O ( 1)( 1)(111~1)(1XX)( X)

Example 8 — Basic with Specified Capture Clock

If you specify a capture clock using the set_capture_clock command with the -Atpg switch, the
format looks slightly different: the specified clock shows up as “010” (or “101”) during capture

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instead of “0X0” (or “1X1”). If the set_capture_clock clk1 -Atpg command is used, the state
stability display for a scan cell clocked by clk1 in the example design looks like the following:
// /sff1 sff
// (ts)(ld)(shift)(cap)(stbl)
// SE I ( X)( 1)(111~1)(XXX)( X) /scan_en
// D I ( X)( X)(XXX~X)(XXX)( X) /gate1/Y
// SI I ( X)( X)(XXX~X)(XXX)( X) /sff2/QB
// CLK I ( 0)( 0)(010~0)(010)( 0) /clk1
// Q O ( X)( X)(XXX~X)(XXX)( X) /gate2/A0
// QB O ( X)( X)(XXX~X)(XXX)( X) /scan_out1

Example 9 — Setting Stability Check to Off and All_shift

The preceding examples, except as noted in example 3, used the default On setting of the
set_stability_check command. This command has two other settings: All_shift and Off.

For All_shift, the tool simulates the exact number of shifts. This means for situations where
particular events take place during shift, the tool will simulate these exactly rather than
simulating the stable state after an “infinite” number of shifts. The following state_stability
displays show the difference between using On and using All_shift for ff32 and the procedures
of the basic example (values of interest are highlighted in bold):

set_stability_check On:

// /ff32 dffr
// (ts)(ld)(shift)(cap)(stbl)
// R I ( 0)( 0)(000~0)(0X0)( 0) /reset
// CLK I ( 0)( 0)(010~0)(0X0)( 0) /clk1
// D I ( 0)( 0)(000~X)(XXX)( X) /ff31b/Q
// Q O ( 0)( 0)(000~X)(XXX)( X)
// QB O ( 1)( 1)(111~X)(XXX)( X)

set_stability_check All_shift:

// /ff32 dffr
// (ts)(ld)(shift)(cap)(stbl)
// R I ( 0)( 0)(000~0)(0X0)( 0) /reset
// CLK I ( 0)( 0)(010~0)(0X0)( 0) /clk1
// D I ( 0)( 0)(000~X)(XXX)( X) /ff31b/Q
// Q O ( 0)( 0)(000~1)(1XX)( X)
// QB O ( 1)( 1)(111~0)(0XX)( X)

In the All_shift case, notice how the stable state after shift differs from the On case. After
exactly three applications of the shift procedure, the state is 1, but after “infinite” applications of
the shift procedure, it is X.

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 Debugging State Stability

When stability checking is set to off, the tool reports only dashes:

// /ff32 dffr
// R I (-) /reset
// CLK I (-) /clk1
// D I (-) /ff31/Q
// Q O (-)
// QB O (-)

Example 10 — Pin Constraints, Test_setup, and State Stability

In this example, the test_setup procedure is modified so that pin D, which is constrained to C0
by an add_input_constraints command, is not forced in the first cycle, but is forced to 1 in the
second cycle. It is never forced to its constrained value 0.

It is a good practice to always force the constrained pins to their constrained state at the end of
the test_setup procedure. If you do not do that, the tool will add an additional cycle to the
test_setup procedure when you write out the patterns. This recommended practice is not
followed in this example:

procedure test_setup =
scan_group grp1
timeplate gen_tp1 ;
// First cycle, three PI events (force, pulse on, pulse off)
cycle =
force clk1 0 ;
force clk2 0 ;
force reset 0 ;
force A 0 ;
force B 0 ;
force C 0 ;
force E 0 ;
pulse clk1 ;
end ;
// Second cycle, three PI events (force, pulse on, pulse off)
cycle =
force D 1 ;
pulse clk1 ;
end ;
end;

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Notice what happens during test_setup:

set_gate_report drc_pattern test_setup

report_gates D ff10
// /D primary_input
// D O /ff10/D
//
// Cycle: 0 1
// ------ --- ---
// 23 467
// Time: 000 000
// ------ --- ---
// D XXX 111
//
// /ff10 dff
// CLK I /clk1
// D I /D
// Q O
// QB O
//
// Cycle: 0 1
// ------ --- ---
// 23 467
// Time: 000 000
// ------ --- ---
// CLK 010 010
// D XXX 111
// Q XXX X11
// QB XXX X00

Then, for state stability, notice how D changes from 1 to 0 between test_setup and the first
application of load_unload. This is because of the pin constraint to 0 on this pin:

set_gate_report drc_pattern state_stability

report_gates D ff10
// /D primary_input
// (ts)(ld)(shift)(cap)(stbl)
// D O ( 1)( 0)(000~0)(000)( X) /ff10/D
// /ff10 dff
// (ts)(ld)(shift)(cap)(stbl)
// CLK I ( 0)( 0)(010~0)(0X0)( 0) /clk1
// D I ( 1)( 0)(000~0)(000)( X) /D
// Q O ( 1)( 1)(100~X)(XXX)( X)
// QB O ( 0)( 0)(011~X)(XXX)( X)

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Example 11 — Single Pre Shift

In the load_unload procedure for this example, a single pre shift is followed by one cycle and
then the main shift. Also, pins C and E vary in the different cycles of the load_unload and shift
operations. The differences from the basic versions of these procedures are highlighted in bold.

procedure load_unload =
scan_group grp1 ;
timeplate gen_tp1 ;
// First cycle, one event (force)
cycle =
force clk1 0 ;
force clk2 0 ;
force reset 0 ;
force scan_en 1 ;
force B 1;
force C 1;
force E 1;
end ;
apply shift 1; // Pre shift
// Second cycle, three PI events (force, pulse on, pulse off)
cycle =
force C 0;
force E 0;
pulse clk2;
end;
apply shift 2; // Main shift
end;

procedure shift =
scan_group grp1 ;
timeplate gen_tp1 ;
cycle =
force_sci ;
measure_sco ;
pulse clk1 ;
force C 0;
end;
end;

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The third group in the state_stability display (highlighted in bold) is the pre-shift. The fourth
group is the cycle between the shift applications, and the fifth group is the main shift.

set_gate_report drc_pattern state_stability

report_gates C E clk1 clk2
// /C primary_input
// (ts)(ld)(shift)( ld)(shift)(cap)(stbl)
// C O ( 0)( 1)( 000 )(000)(000~0)(XXX)( X)
// /E primary_input
// (ts)(ld)(shift)( ld)(shift)(cap)(stbl)
// E O ( 0)( 1)( 111 )(000)(000~0)(XXX)( X)
// /clk1 primary_input
// (ts)(ld)(shift)( ld)(shift)(cap)(stbl)
// clk1 O ( 0)( 0)( 010 )(000)(010~0)(0X0)( 0)/ff32/CLK
// /ff31b/CLK…
// /clk2 primary_input
// (ts)(ld)(shift)( ld)(shift)(cap)
// clk2 O ( 0)( 0)( 000 )(010)(000~0)(0X0)/ff20/CLK

Example 12 — Basic with Enhanced Stability Check for NCPs

If you enable the enhanced stability check for NCPs using “set_stability_check
-sim_capture_procedures”, the report_gates output format looks slightly different. Notice that
for the “(cap)” column of the stability report below, the tool presents two simulation values
separated with a tilde character (~). The first value is the value at the beginning of capture, and
the second value is the value at the end of capture. The tool determines these values by taking
the common value among all NCP simulation values. Since different NCPs can be of different
length in sequence, the tool uses the tilde separator to skip the events in between in order to
highlight only the first and last value.

There is only one exception to the display of the tilde separator, which is when the simulation is
a constant 1 or 0 among all NCP cycles. Then instead of showing the tilde separator, the middle
value is the same constant value (1 or 0). The first and last value is usually of the most interest
when debugging stability analysis.

In the following alteration to the basic example on page 449, the tool is already invoked on the
design, and you enter the following commands:

add_clocks 0 clk1 clk2 reset

add_scan_groups grp1 scan1.testproc
add_scan_chains c0 grp1 scan_in1 scan_out1
add_input_constraints D -c0
set_gate_report drc_pattern state_stability
set_stability_check -sim_capture_procedures on # enables more detailed analysis
set_system_mode analysis

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 Debugging State Stability

report_gates A B C D E
...
// (ts)(ld)(shift)(cap)(stbl)
// A O ( 1)( 1)(111~1)(X~X)( X )
// B O ( 0)( 1)(111~1)(X~X)( X )
// C O ( 0)( 1)(000~0)(X~X)( X )
// D O ( 0)( 0)(000~0)(0~0)( 0 )
// E O ( 0)( 0)(000~0)(X~X)( X )

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 Appendix C
Running Tessent Shell as a Batch Job

You can use Tessent Shell in either an interactive or non-interactive manner. You conduct a tool
session interactively by entering the commands manually, or the session can be completely
scripted and driven using a dofile. This non-interactive mode of operation allows the entire
session to be conducted without user interaction. This method of using Tessent Shell can be
further expanded to allow the session to be scheduled and run as a true batch or cron job. This
appendix focuses on the features of Tessent Shell that support its use in a batch environment.

Commands and Variables for the dofile

The following shell script invokes Tessent Shell and then runs a dofile that sets the context to
“patterns -scan,” reads a design, and reads a cell library. The dofile also specifies an option to
exit from the dofile upon encountering an error:
set_dofile_abort exit

The exit option sets the exit code to a non-zero value if an error occurs during execution of the
dofile. This allows a shell script that launches a Tessent Shell session to control process flow
based on the success or failure of a tool operation. Note the line check for the exit status
following the line that invokes Tessent Shell.

#!/bin/csh -b
##
## Add the pathname of the <Tessent_Tree_Path>/bin directory to the PATH
## environment variable so you can invoke the tool without typing the full
## pathname
##
setenv PATH <Tessent_Tree_Path>/bin:${PATH}
##
setenv DESIGN `pwd`
##
##
tessent -shell -dofile ${DESIGN}/tshell.do \
-license_wait 30 -log ${DESIGN}/`date +log_file_%m_%d_%y_%H:%M:%S`
setenv proc_status $status
if ("$proc_status" == 0 ) then
echo "Session was successful"
echo " The exit code is: " $proc_status
else echo "Session failed"
echo " The exit code is: " $proc_status
endif
echo $proc_status " is the exit code value."

You can use environment variables in a Tessent Shell dofile. For example, the shell script sets
the DESIGN environment variable to the current working directory. When a batch job is

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Command Line Options

created, the process may not inherit the same environment that existed in the shell environment.
To assure that the process has access to the files referenced in the dofile, the DESIGN
environment variable is used. A segment of a Tessent Shell dofile displaying the use of an
environment variable follows:

# The shell script that launches this dofile sets the DESIGN environment
# variable to the current working directory.
add_scan_groups g1 ${DESIGN}/procfile
#
add_scan_chains c1 g1 scan_in CO
...
#
write_faults ${DESIGN}/fault_list -all -replace

You can also use a startup file to alias common commands. To set up the predefined alias
commands, use the file .tessent_startup (located by default in your home directory). For
example:

alias save_my_pat write_patterns $1/pats.v -$2 -replace

The following dofile segment displays the use of the alias defined in the .tessent_startup file:

# The following alias is defined in the .tessent_startup file

#
save_my_pat $DESIGN verilog

Another important consideration is to exit in a graceful manner from the dofile. This is required
to assure that Tessent Shell exits instead of waiting for additional command line input.

# The following command terminates the Tessent Shell dofile.

#
exit -force

Command Line Options

Several Tessent Shell command line options are useful when running Tessent Shell as a batch
job. One of these options is the -LICense_wait option, which sets a limit for retrying license
acquisition after you set a context. The default is no time limit for a license.
If Tessent Shell is unable to obtain a license after the specified number of retries, the tool exits.
An example of the Tessent Shell invocation line with this option follows:

% tessent -shell-dofile tshell.do -license_wait 30 \

-log ${DESIGN}/`date +log_file_%m_%d_%y_%H:%M:%S`

Another item of interest is the logfile name created using the Linux “date” command for each
Tessent Shell run. The logfile is based on the month, day, year, hour, minute, and second that
the batch job was launched. An example of the logfile name that would be created follows:

log_file_05_30_12_08:42:37

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Command Line Options

Scheduling a Batch Job for Execution Later

You can schedule a batch job using the Linux “at” or “cron” command. For more information
about using either of these commands, use the Linux “man” command.

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Command Line Options

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 Appendix D
Getting Help

There are several ways to get help when setting up and using Tessent software tools. Depending
on your need, help is available from documentation, online command help, and Mentor
Graphics Support.

Documentation
The Tessent software tree includes a complete set of documentation and help files in PDF
format. Although you can view this documentation with any PDF reader, if you are viewing
documentation on a Linux file server, you must use only Adobe® Reader® versions 8 or 9, and
you must set one of these versions as the default using the MGC_PDF_READER variable in
your mgc_doc_options.ini file.
For more information, refer to “Specifying Documentation System Defaults” in the Managing
Mentor Graphics Tessent Software manual.

You can download a free copy of the latest Adobe Reader from this location:

http://get.adobe.com/reader

You can access the documentation in the following ways:

• Shell Command — On Linux platforms, enter mgcdocs at the shell prompt or invoke a
Tessent tool with the -Manual invocation switch. This option is available only with
Tessent Shell and the following classic point tools: Tessent FastScan, Tessent
TestKompress, Tessent Diagnosis, and DFTAdvisor.
• File System — Access the Tessent bookcase directly from your file system, without
invoking a Tessent tool. From your product installation, invoke Adobe Reader on the
following file:
$MGC_DFT/doc/pdfdocs/_bk_tessent.pdf

• Application Online Help — You can get contextual online help within most Tessent
tools by using the “help -manual” tool command:
> help dofile -manual

This command opens the appropriate reference manual at the “dofile” command
description.

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Getting Help
Mentor Graphics Support

Mentor Graphics Support

Mentor Graphics software support includes software enhancements, access to comprehensive
online services with SupportNet, and the optional On-Site Mentoring service.
For details, refer to this page:

http://supportnet.mentor.com/about

If you have questions about a software release, you can log in to SupportNet and search
thousands of technical solutions, view documentation, or open a Service Request online:

http://supportnet.mentor.com

If your site is under current support and you do not have a SupportNet login, you can register for
SupportNet by filling out a short form here:

http://supportnet.mentor.com/user/register.cfm

All customer support contact information is available here:

http://supportnet.mentor.com/contacts/supportcenters/index.cfm

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A B C D E F G H I J K L M N O P Q R S T U V W X Y Z

Index

add_clocks, 187
Index

—A— add_false_paths, 234

Abort limit, 207 add_faults, 190, 194, 195
Aborted faults, 207 add_input_constraints, 177, 179
changing the limits, 207 add_lists, 191, 192
reporting, 207 add_nofaults, 189
Ambiguity add_output_masks, 179
edge, 243 add_primary_inputs, 178
path, 243 add_scan_chains, 188
ASCII WGL format, 376 add_scan_groups, 187
ASIC Vector Interfaces, 19 add_slow_pad, 182
ATPG add_tied_signals, 181
applications, 31 analyze_atpg_constraints, 200
ATPG method, 171 to 176 analyze_bus, 183
basic operations, 176 analyze_fault, 239
constraints, 198 analyze_graybox, 427
default run, 205 analyze_restrictions, 200
defined, 30 create_flat_model, 75
features, 31 delete_atpg_constraints, 200
for IDDQ, 211 delete_atpg_functions, 200
for path delay, 234 delete_cell_constraints, 188
for transition fault, 220 to 225 delete_clocks, 187
full scan, 31 delete_false_paths, 234
function, 198 delete_fault_sites, 239
inputs and outputs, 170 delete_faults, 194
MacroTest, using, 282 delete_input_constraints, 177, 182
non-scan cell handling, 105 to 108 delete_multicycle_paths, 234
pattern types, 172 to 176 delete_nofaults, 189
process, 197 delete_primary_inputs, 178
setting up faults, 190, 194 delete_primary_outputs, 178
test cycles, 172 delete_scan_chains, 188
timing model, 172 delete_scan_groups, 187
tool flow, 168 delete_tied_signals, 181
ATPG Commands read_fault_sites, 239
create_patterns, 269 read_faults, 195
set_atpg_timing, 271, 274 read_patterns, 190
ATPG commands report_aborted_faults, 207
add_ambiguous_paths, 239 report_atpg_constraints, 200
add_capture_handling, 186 report_atpg_functions, 200
add_cell_constraints, 188 report_bus_data, 183

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report_cell_constraints, 188 write_patterns, 208

report_clocks, 187 ATPG tool, definition of, 17
report_false_paths, 234 At-speed test, 33, 219 to 258
report_fault_sites, 239
report_faults, 191, 195, 206 —B—
report_gates, 183 Batch mode
report_input_constraints, 177, 182 see also Appendix C
report_nofaults, 189 Bidirectional pin
report_nonscan_cells, 108 as primary input, 178
report_paths, 239 as primary output, 178
report_primary_inputs, 178 Boundary scan, defined, 21
report_primary_outputs, 178 Bridge fault testing
report_scan_chains, 188 creating the test set, 232
report_scan_groups, 187 static bridge fault model, 39
report_statistics, 191 Bus
report_tied_signals, 181 dominant, 79
reset_state, 192 float, 103
run, 205 Bus contention, 103
set_abort_limit, 207, 224 checking during ATPG, 183
set_bus_handling, 183 fault effects, 183
set_capture_clock, 191 —C—
set_checkpointing_options, 204 Capture handling, 185
set_clock_restriction, 188 Capture point, 41
set_contention_check, 183 Capture procedure, see Named capture
set_drc_handling, 186 procedure
set_driver_restriction, 183 Chain test, 373
set_fault_mode, 196 Checkpointing example, 204
set_fault_type, 190, 194, 224 Checkpoints, setting, 203
set_learn_report, 182 Clock
set_list_file, 191, 192 capture, 187, 277
set_net_dominance, 183 list, 187
set_net_resolution, 183 off-state, 187
set_pattern_buffer, 182 scan, 187
set_pattern_type, 108, 174, 175, 224 Clock PO patterns, 173
set_possible_credit, 182, 196 Clock procedure, 172
set_random_atpg, 207 Clock sequential patterns, 174
set_random_clocks, 191 Clocked sequential test generation, 107
set_random_patterns, 191 Clocks, merging chains with different, 145
set_split_capture_cycle, 186 Combinational loop, 94, 95, 96, 97, 98, 99
set_system_mode, 177, 189 cutting, 95
set_tied_signals, 181 Constant value loops, 95
set_z_handling, 184 Constraints
simulate_patterns, 191 ATPG, 198
write_fault_sites, 239 pin, 181
write_faults, 195 scan cell, 188

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Contention, bus, 84 Differential scan input pins, 372

Control points Distributed ATPG
manual identification, 134 troubleshooting SSH, 325
Controllability, 27 Dominant bus, 79
Copy, scan cell element, 70
Coupling loops, 98 —E—
CPF commands, 394 Edge ambiguity, 243
create_patterns, 269 Existing scan cells
Creating a delay test set, 220 handling, 128
Creating patterns External pattern generation, 31
default run, 205 Extra, scan cell element, 71
Cycle count, 375 —F—
Cycle test, 373 Fault
—D— aborted, 207
Data capture simulation, 185 classes, 55 to 65
Debugging simulation mismatches, 305 collapsing, 36
Defect, 32 detection, 54
Delay test coverage, 266 no fault setting, 189
Design Compiler, handling pre-inserted scan representative, 36, 65
cells, 128 simulation, 190
Design flattening, 75 to 80 undetected, 207
Design flow, delay test set, 220 Fault grading
Design rules checking in multiple fault model flow, 263
blocked values, 88 pattern generation and, 263
bus keeper analysis, 87 Fault models
bus mutual-exclusivity, 84 bridge, 39
clock rules, 86 path delay, 41
constrained values, 88 pseudo stuck-at, 38
data rules, 86 stuck-at, 36, 39
extra rules, 87 toggle, 37
forbidden values, 88 transition, 40
general rules, 83 Fault sampling, 203
introduction, 83 Faults
procedure rules, 83 small delay, 265
RAM rules, 86 Feedback loops, 94
scan chain tracing, 85 Fixed-order file, 142
scannability rules, 88 Flattening, design, 75 to 80
shadow latch identification, 85 Frame, simulation, 201
transparent latch identification, 86 Full scan, 24, 122
Design-for-Test, defined, 17 Functional test, 33
Deterministic test generation, 31 —G—
DFF modeling, 201 Gate duplication, 96
DFTVisualizer Good simulation, 192
see "DFTVisualizer" in the Tessent Shell
User’s Manual

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—H— leading & trailing edge observation,

Head register, attaching, 143 301
Hierarchical instance, definition, 75 synchronous memories, 299
Hold_pi, 274 macro boundary
defining, 289
—I— with instance name, 289
IDDQ testing, 211 with trailing edge inputs, 292
creating the test set, 211 without instance name, 290
defined, 33 reporting & specifying observation
methodologies, 33 sites, 291
pseudo stuck-at fault model, 38 overview, 282
test pattern formats, 372 qualifying macros, 286
vector types, 34 recommendations for using, 295
Incomplete designs, 117 test values, 293
Instance, definition, 75 when to use, 287
Internal scan, 21, 22 Manufacturing defect, 32
—L— Mapping scan cells, 123
Latch modeling, 201 Mask_po, 274
Latches Masking primary outputs, 181, 182
handling as non-scan cells, 104 Master, scan cell element, 69
scannability checking of, 94 Memories, testing, 282
Launch point, 41 Merging scan chains, 145
Layout-sensitive scan insertion, 145 Module, definition, 75
Learning analysis, 80 to 82 MTFI, 415
dominance relationships, 82 Multiple load patterns, 174
equivalence relationships, 80 —N—
forbidden relationships, 82 Named capture procedure, 245
implied relationships, 81 internal and external mode of, 247 to 253
logic behavior, 81 on-chip clocks and, 246
Line holds, 56, 57 No fault setting, 189
Loop count, 375 Non-scan cell handling, 104 to 108
Loop cutting, 95 ATPG, 105
by constant value, 95 clocked sequential, 107
by gate duplication, 96 sequential transparent, 106
for coupling loops, 98 tie-0, 105
single multiple fanout, 96 tie-1, 105
Loop handling, 94 tie-X, 105
—M— transparent, 105
Macros, 34 Non-scan sequential instances
MacroTest, 282 reporting, 136
basic flow, 283, 284 —O—
capabilities, summary, 283 Observability, 27
examples, 297, 299, 301 Observe points
basic 1-Cycle Patterns, 297 manual identification, 134

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Off-state, 73, 127, 187 power-aware ATPG procedure, 396

One-shot circuit, 274 power-aware overview, 391
Output mask, 274 Pre-inserted scan cells
handling, 128
—P— Primary inputs
Parallel scan chain loading, 370 bidirectional pins as, 178
Partition scan, 25, 122 constraining, 181
Path ambiguity, 243 constraints, 181
Path definition file, 240 Primary outputs
Path delay testing, 41, 234 bidirectional pins as, 178
basic procedure, 244 masking, 182
limitations, 245 Primitives, simulation, 77
multiple fault models and Pulse generators
flow, 263 gate reporting for, 454
path ambiguity, 243 introduction, 109
path definition checking, 243
path definition file, 240 —R—
patterns, 235 RAM
robust detection, 236 ATPG support, 112
transition detection, 236 common read and clock lines, 114
Path sensitization, 54 common write and clock lines, 114
Pattern formats pass-through mode, 112
binary, 375 RAM sequential mode, 113
text, 372 RAM sequential mode, read/write clock
Verilog, 375 requirement, 113
WGL (ASCII), 376 read-only mode, 112
Pattern generation related commands, 115 to 116
deterministic, 31 rules checking, 116 to 117
external source, 31 sequential patterns, 175
multiple fault models and testing, 111 to 117
flow, 263 Random pattern generation, 30
random, 30 Registers
Pattern generation phase head, attaching, 143
test procedure waveforms, example, 438 tail, attaching, 143
Pattern types report_graybox_statistics, 427
basic scan, 172 ROM
clock PO, 173 ATPG support, 112
clock sequential, 174 related commands, 115 to 116
multiple load, 174 rules checking, 116
RAM sequential, 175 testing, 111 to 117
sequential transparent, 176
Performing ATPG, 197 to 208 —S—
Pin constraints, 181 s, 65
plusargs, Verilog, 303 Sampling, fault, 203
Possible-detect credit, 59, 196 Saving patterns
Possible-detected faults, 59 Serial versus parallel, 370

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Scan Sequential loop, 94, 100

basic operation, 23 Sequential transparent latch handling, 106
clock, 23 Sequential transparent patterns, 176
Scan and/or Wrapper test structures, 123 Serial scan chain loading, 371
Scan cell set_atpg_timing, 271, 274
concepts, 67 Shadow, 69
constraints, 188 Simulating captured data, 185
existing, 128 Simulation data formats, 372
mapping, 123 Simulation formats, 369
pre-inserted, 128 Simulation frame, 201
Scan cell elements Simulation mismatches, debugging, 305
copy, 70 Simulation primitives, 77 to 80
extra, 71 AND, 77
master, 69 BUF, 77
shadow, 69 BUS, 79
slave, 69 DFF, 78
Scan chains INV, 77
assigning scan pins, 142 LA, 78
definition, 71 MUX, 78
fixed-order file, 142 NAND, 77
head and tail registers, attaching, 143 NMOS, 79
merging, 145 NOR, 77
parallel loading, 370 OR, 77
serial loading, 371 OUT, 80
serial versus parallel loading, 370 PBUS, 79
specifying, 187 PI, 77
Scan clocks, 73, 127 PO, 77
specifying, 127, 187 RAM, 80
Scan design ROM, 80
simple example, 23 STFF, 78
Scan design, defined, 21 STLA, 78
Scan groups, 72, 187 SW, 79
Scan insertion SWBUS, 79
layout-sensitive, 145 TIE gates, 78
process, 119 TLA, 78
Scan output mapping, 123 TSD, 78
Scan patterns, 172 TSH, 78
Scan pins WIRE, 79
assigning, 142 XNOR, 78
Scan related events, 366 XOR, 78
Scan sub-chain, 370 ZHOLD, 79
Scan test, 373 ZVAL, 77
Scannability checks, 93 Single multiple fanout loops, 96
SDF file, 272 Sink gates, 185
Sequential element modeling, 201 Slack, 265

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Slave, 69 report_scan_cells, 147

Small delay fault model, 265 report_scan_chains, 147
Source gates, 185 report_scan_groups, 147
SSH protocol report_scan_models, 136
troubleshooting, 325 report_scan_pins, 142
Static-inactive path, 274 report_sequential_instances, 136, 138
Structural loop, 94 report_shift_registers, 140
combinational, 94 report_statistics, 138
sequential, 94 report_test_logic, 127
Structured DFT, 18 set_scan_pins, 142
System-class set_system_mode, 131
non-scan instances, 134 write_primary_inputs, 128
scan instances, 134 Tessent Scan, available in Tessent Shell, 17
Tessent Shell, providing ATPG and Scan
—T— functionality, 17
Tail register, attaching, 143 Tessent TestKompress, available in Tessent
Tessent Core Description File (TCD), 154, 339 Shell, 17
top-level, 343 Test bench, Verilog, 303
Tessent FastScan, available in Tessent Shell, Test clock, 109, 124
17 Test logic, 93, 109, 124
Tessent Scan Test Pattern File Format, 379
block-by-block scan insertion, 148 to 151 Functional Chain Test, 382
features, 29 Header_Data, 379
inputs and outputs, 121 Scan Cell, 385
invocation, 123 Scan Test, 383
process flow, 119 Test patterns, 30
supported test structures, 122 chain test block, 373
Tessent Scan commands scan test block, 374
add_clocks, 127 Test points
add_nonscan_instances, 135 definition of, 123
add_scan_chains, 129 understanding, 27
add_scan_groups, 128 Test procedure file
add_scan_pins, 142, 144 in Tessent Scan, 122
delete_cell_models, 127 Test structures
delete_clocks, 127 full scan, 24 to 25, 122
delete_nonscan_instances, 136 partition scan, 25 to 27, 122
delete_nonscan_models, 136 Scan and/or Wrapper, 123
delete_scan_instances, 136 supported by Tessent Scan, 122
delete_scan_models, 136 test points, 27 to 28, 123
delete_scan_pins, 142 Test types
report_cell_models, 127 at-speed, 34
report_clocks, 127 functional, 33
report_control_signals, 137 IDDQ, 33
report_dft_check, 137, 147 Test vectors, 30
report_nonscan_models, 136 Testability, 17
report_primary_inputs, 128

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Testing
memories, 282
TI TDL 91 format, 377 to ??
Tie-0 gate, 105
TIE0, scannable, 93
Tie-1 gate, 105
TIE1, scannable, 93
Tie-X gate, 105
Timing-aware ATPG
detection paths, 274
example, 269
limitations, 267
reducing run time, 271
setting up, 268
troubleshooting, 271
Toshiba TSTL2 format, 378
Transition ATPG, 266
Transition fault testing, 220 to 225
Transition testing
basic procedures, 225
patterns, 222
Transparent latch handling, 105
Transparent slave handling, 107
Troubleshooting
sdf file, 272
timing-aware ATPG, 271
—U—
UDFM, 42
Undetected faults, 207
UPF commands, 394
User-class
non-scan instances, 134
scan instances, 136
test points, 134
User-Defined Fault Model, 42
—V—
Verilog, 375
Verilog plusargs, 303
Verilog test bench, 303

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 Third-Party Information
For information about third-party software included with this release of Tessent products, refer to the Third-Party Software for
Tessent Products.
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reference in this section. Nothing in this section shall restrict Mentor Graphics’ right to bring an action (including for example a motion
for injunctive relief) against Customer in the jurisdiction where Customer’s place of business is located. The United Nations
Convention on Contracts for the International Sale of Goods does not apply to this Agreement.

18. SEVERABILITY. If any provision of this Agreement is held by a court of competent jurisdiction to be void, invalid, unenforceable or
illegal, such provision shall be severed from this Agreement and the remaining provisions will remain in full force and effect.

19. MISCELLANEOUS. This Agreement contains the parties’ entire understanding relating to its subject matter and supersedes all prior
or contemporaneous agreements. Some Software may contain code distributed under a third party license agreement that may provide
additional rights to Customer. Please see the applicable Software documentation for details. This Agreement may only be modified in
writing, signed by an authorized representative of each party. Waiver of terms or excuse of breach must be in writing and shall not
constitute subsequent consent, waiver or excuse.

Rev. 140201, Part No. 258976