MP4560

2A, 2MHz, 55V

Step-Down Converter
The Future of Analog IC Technology

DESCRIPTION FEATURES
The MP4560 is a high frequency step-down  Wide 3.8V to 55V Operating Input Range
switching regulator with integrated internal high-  250mΩ Internal Power MOSFET
side high voltage power MOSFET. It provides 2A  Up to 2MHz Programmable Switching
output with current mode control for fast loop Frequency
response and easy compensation.  140μA Quiescent Current
 Ceramic Capacitor Stable
The wide 3.8V to 55V input range accommodates  Internal Soft-Start
a variety of step-down applications, including
 Up to 95% Efficiency
those in automotive input environment. A 12µA
 Output Adjustable from 0.8V to 52V
shutdown mode supply current allows use in
battery-powered applications.  Available in 3x3 10-Pin QFN and SOIC8
with Exposed Pad Packages
High power conversion efficiency over a wide
load range is achieved by scaling down the APPLICATIONS
switching frequency at light load condition to  High Voltage Power Conversion
reduce the switching and gate driving losses.  Automotive Systems
The frequency foldback helps prevent inductor  Industrial Power Systems
current runaway during startup and thermal  Distributed Power Systems
shutdown provides reliable, fault tolerant  Battery Powered Systems
operation. All MPS parts are lead-free and adhere to the RoHS directive. For MPS green
status, please visit MPS website under Products, Quality Assurance page.
By switching at 2MHz, the MP4560 is able to “MPS” and “The Future of Analog IC Technology” are registered trademarks of
prevent EMI (Electromagnetic Interference) noise Monolithic Power Systems, Inc.
problems, such as those found in AM radio and
ADSL applications.
The MP4560 is available in small 3mm x 3mm
10-pin QFN and SOIC8 with exposed pad
packages.

TYPICAL APPLICATION
Efficiency @VOUT=3.3V
fs=500kHz
100
Vin=12V
90
80
70
EFFICIENCY (%)

60 Vin=55V
50
40
30
20
10
0
0 0.5 1 1.5 2
OUTPUT CURRENT (A)

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 MP4560 – 2A, 2MHz, 55V STEP-DOWN CONVERTER

ORDERING INFORMATION
Part Number Package Top Marking
MP4560DQ* 3x3 QFN10 T8
MP4560DN** SOIC8E MP4560DN

* For Tape & Reel, add suffix –Z (e.g. MP4560DQ–Z).

For RoHS compliant packaging, add suffix –LF (e.g. MP4560DQ–LF–Z)
** For Tape & Reel, add suffix –Z (e.g. MP4560DN–Z).
For RoHS compliant packaging, add suffix –LF (e.g. MP4560DN–LF–Z)

PACKAGE REFERENCE

TOP VIEW
TOP VIEW
SW 1 10 BST
SW 1 8 BST
SW 2 9 VIN
EN 2 7 VIN
EN 3 8 VIN
COMP 3 6 FREQ
COMP 4 7 FREQ
FB 4 5 GND
FB 5 6 GND

EXPOSED PAD
EXPOSED PAD

3x3 QFN10 SOIC8E

ABSOLUTE MAXIMUM RATINGS (1) Thermal Resistance

(4)
θJA θJC
Supply Voltage (VIN).....................–0.3V to +60V 3x3 QFN10..............................50 ...... 12 ... C/W
Switch Voltage (VSW)............ –0.5V to VIN + 0.5V SOIC8 (Exposed Pad) ............50 ...... 10 ... C/W
BST to SW .....................................–0.3V to +5V
All Other Pins .................................–0.3V to +5V Notes:
(2) 1) Exceeding these ratings may damage the device.
Continuous Power Dissipation(TA = +25°C) 2) The maximum allowable power dissipation is a function of the
3x3 QFN10……………………………………2.5W maximum junction temperature TJ (MAX), the junction-to-
ambient thermal resistance θJA, and the ambient temperature
SOIC8 (Exposed Pad)………………………2.5W TA. The maximum allowable continuous power dissipation at
Junction Temperature ...............................150C any ambient temperature is calculated by PD (MAX) = (TJ
(MAX)-TA)/θJA. Exceeding the maximum allowable power
Lead Temperature ....................................260C dissipation will cause excessive die temperature, and the
Storage Temperature.............. –65°C to +150C regulator will go into thermal shutdown. Internal thermal
shutdown circuitry protects the device from permanent
(3) damage.
Recommended Operating Conditions 3) The device is not guaranteed to function outside of its
Supply Voltage VIN ...........................3.8V to 55V operating conditions.
Output Voltage VOUT .........................0.8V to 52V 4) Measured on JESD51-7, 4-layer PCB.
Operating Junction Temp. (TJ). -40°C to +125°C

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 MP4560 – 2A, 2MHz, 55V STEP-DOWN CONVERTER

ELECTRICAL CHARACTERISTICS
VIN = 12V, VEN = 2.5V, VCOMP = 1.4V, TA= +25C, unless otherwise noted.
Specifications over temperature are guaranteed by design and characterization.
Parameter Symbol Condition Min Typ Max Units
4.5V < VIN < 55V 0.780 0.800 0.820
Feedback Voltage VFB V
–40°C to +85C 0.772 0.829
VBST – VSW = 5V 175 250 330
Upper Switch On Resistance (5) RDS(ON) mΩ
–40°C to +85C 160 400
Upper Switch Leakage VEN = 0V, VSW = 0V 1 μA
2.6 3.2 4.5
Current Limit A
–40°C to +85C 2.2 4.7
COMP to Current Sense
GCS 5.7 A/V
Transconductance
Error Amp Voltage Gain 400 V/V
Error Amp Transconductance ICOMP = ±3µA 120 µA/V
Error Amp Min Source current VFB = 0.7V 10 µA
Error Amp Min Sink current VFB = 0.9V -10 µA
2.7 3.0 3.3
VIN UVLO Threshold V
–40°C to +85C 2.4 3.6
VIN UVLO Hysteresis 0.35 V
Soft-Start Time (5) 0V < VFB < 0.8V 0.5 ms
RFREQ = 95kΩ 0.8 1 1.2
Oscillator Frequency MHz
–40°C to +85C 0.7 1.3
Minimum Switch On Time 100 ns
Shutdown Supply Current VEN < 0.3V 12 20 µA
Quiescent Supply Current No load, VFB = 0.9V 140 µA
Thermal Shutdown Hysteresis = 20C 150 C
Minimum Off Time 100 ns
Minimum On Time (5) 100 ns
1.4 1.55 1.7
EN Up Threshold V
–40°C to +85C 1.3 1.8
EN Threshold Hysteresis 320 mV
Note:
5) Derived from bench characterization. Not tested in production.

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 MP4560 – 2A, 2MHz, 55V STEP-DOWN CONVERTER

PIN FUNCTIONS
QFN SOIC8
Name Description
Pin # Pin #
Switch Node. This is the output from the high-side switch. A low VF Schottky rectifier
1, 2 1 SW to ground is required. The rectifier must be close to the SW pins to reduce switching
spikes.
Enable Input. Pulling this pin below the specified threshold shuts the chip down.
3 2 EN
Pulling it up above the specified threshold or leaving it floating enables the chip.
Compensation. This node is the output of the GM error amplifier. Control loop
4 3 COMP
frequency compensation is applied to this pin.
Feedback. This is the input to the error amplifier. An external resistive divider
5 4 FB connected between the output and GND is compared to the internal +0.8V reference
to set the regulation voltage.
GND, Ground. It should be connected as close as possible to the output capacitor avoiding
6 5 Exposed the high current switch paths. Connect exposed pad to GND plane for optimal thermal
pad performance.
Switching Frequency Program Input. Connect a resistor from this pin to ground to set
7 6 FREQ
the switching frequency.
Input Supply. This supplies power to all the internal control circuitry, both BS
8, 9 7 VIN regulators and the high-side switch. A decoupling capacitor to ground must be placed
close to this pin to minimize switching spikes.
Bootstrap. This is the positive power supply for the internal floating high-side
10 8 BST
MOSFET driver. Connect a bypass capacitor between this pin and SW pin.

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 MP4560 – 2A, 2MHz, 55V STEP-DOWN CONVERTER

TYPICAL PERFORMANCE CHARACTERISTICS

VIN = 12V, VOUT =3.3V, C1 = 4.7µF, C2 = 22µF, L1 = 10µH and TA = +25C, unless otherwise noted.
Efficiency @VOUT=2.5V Efficiency @VOUT=5V Output Voltage Ripple
L1=15uH, fs=500kHz IOUT=0.1A
90 100
Vin=12V Vin=12V
80 90
VOUT
80 AC Coupled
70
70 Vin=55V 10mV/div
EFFICIENCY (%)

EFFICIENCY (%)
60 Vin=48V
60
50
50
40 Vsw
40
10V/div
30 30
20 20 IL
10 10 500mA/div

0 0
0 0.5 1 1.5 2 0 0.5 1 1.5 2

OUTPUT CURRENT (A) OUTPUT CURRENT (A)

Output Voltage Ripple Output Voltage Ripple

IOUT=1A IOUT=2A

VOUT VOUT
AC Coupled AC Coupled
10mV/div 10mV/div

Vsw Vsw
10V/div 10V/div

IL IL
1A/div 2A/div

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 MP4560 – 2A, 2MHz, 55V STEP-DOWN CONVERTER

TYPICAL PERFORMANCE CHARACTERISTICS

VIN = 12V, VOUT =3.3V, C1 = 4.7µF, C2 = 22µF, L1 = 10µH and TA = +25C, unless otherwise noted.

Strart up Strart up Strart up

IOUT=0.1A IOUT=1A IOUT=2A

VEN VEN VEN

2V/div 2V/div 2V/div

VOUT VOUT VOUT

2V/div 2V/div 2V/div

Vsw Vsw Vsw

10V/div 10V/div 10V/div
IL
1A/div IL IL
1A/div 2A/div
10ms/div 4ms/div 4ms/div

Shut down Shut down Shut down

IOUT=0.1A IOUT=1A IOUT=2A

VEN VEN VEN

2V/div 2V/div 2V/div

VOUT VOUT VOUT

2V/div 2V/div 2V/div

Vsw Vsw Vsw

10V/div 10V/div 10V/div
IL
1A/div IL IL
1A/div 2A/div
1ms/div

Short Circuit Entry Short Circuit Steady State Short Circuit Recovery
IOUT=0.1A to Short IOUT=Short to 0A

VOUT VOUT VOUT

2V/div 2V/div 2V/div

Vsw Vsw Vsw

10V/div 10V/div 10V/div

IL IL IL
1A/div 1A/div 2A/div

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 MP4560 – 2A, 2MHz, 55V STEP-DOWN CONVERTER

BLOCK DIAGRAM
VIN

REFERENCE INTERNAL
EN
UVLO REGULATORS
BST

ISW --
LOGIC
0.5ms SS SS +

SW

FB -- COMP
SS
0V8 +

OSCILLATOR

COMP GND FREQ

Figure 1—Functional Block Diagram

the power MOSFET does not reach the COMP

OPERATION set current value, the power MOSFET remains
The MP4560 is a programmable frequency, on, saving a turn-off operation.
non-synchronous, step-down switching regulator
Pulse Skipping Mode
with an integrated high-side high voltage power
Under light load condition the switching
MOSFET. It provides a single highly efficient
frequency stretches down zero to reduce the
solution with current mode control for fast loop
switching loss and driving loss.
response and easy compensation. It features a
wide input voltage range, internal soft-start Error Amplifier
control and precision current limiting. Its very low The error amplifier compares the FB pin voltage
operational quiescent current makes it suitable with the internal reference (REF) and outputs a
for battery powered applications. current proportional to the difference between the
two. This output current is then used to charge
PWM Control Mode
the external compensation network to form the
At moderate to high output current, the MP4560
COMP voltage, which is used to control the
operates in a fixed frequency, peak current
power MOSFET current.
control mode to regulate the output voltage. A
PWM cycle is initiated by the internal clock. The During operation, the minimum COMP voltage is
power MOSFET is turned on and remains on clamped to 0.9V and its maximum is clamped to
until its current reaches the value set by the 2.0V. COMP is internally pulled down to GND in
COMP voltage. When the power switch is off, it shutdown mode. COMP should not be pulled up
remains off for at least 100ns before the next beyond 2.6V.
cycle starts. If, in one PWM period, the current in

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 MP4560 – 2A, 2MHz, 55V STEP-DOWN CONVERTER

Internal Regulator Floating Driver and Bootstrap Charging

Most of the internal circuitries are powered from The floating power MOSFET driver is powered by
the 2.6V internal regulator. This regulator takes an external bootstrap capacitor. This floating
the VIN input and operates in the full VIN range. driver has its own UVLO protection. This UVLO’s
When VIN is greater than 3.0V, the output of the rising threshold is 2.2V with a hysteresis of
regulator is in full regulation. When VIN is lower 150mV. The driver’s UVLO is soft-start related. In
than 3.0V, the output decreases. case the bootstrap voltage hits its UVLO, the
soft-start circuit is reset. To prevent noise, there
Enable Control
is 20µs delay before the reset action. When
The MP4560 has a dedicated enable control pin
bootstrap UVLO is gone, the reset is off and then
(EN). With high enough input voltage, the chip
soft-start process resumes.
can be enabled and disabled by EN which has
positive logic. Its falling threshold is a precision The bootstrap capacitor is charged and regulated
1.2V, and its rising threshold is 1.5V (300mV to about 5V by the dedicated internal bootstrap
higher). regulator. When the voltage between the BST
and SW nodes is lower than its regulation, a
When floating, EN is pulled up to about 3.0V by
PMOS pass transistor connected from VIN to
an internal 1µA current source so it is enabled.
BST is turned on. The charging current path is
To pull it down, 1µA current capability is needed.
from VIN, BST and then to SW. External circuit
When EN is pulled down below 1.2V, the chip is should provide enough voltage headroom to
put into the lowest shutdown current mode. facilitate the charging.
When EN is higher than zero but lower than its
As long as VIN is sufficiently higher than SW, the
rising threshold, the chip is still in shutdown
bootstrap capacitor can be charged. When the
mode but the shutdown current increases slightly.
power MOSFET is ON, VIN is about equal to SW
Under-Voltage Lockout (UVLO) so the bootstrap capacitor cannot be charged.
Under-voltage lockout (UVLO) is implemented to When the external diode is on, the difference
protect the chip from operating at insufficient between VIN and SW is largest, thus making it
supply voltage. The UVLO rising threshold is the best period to charge. When there is no
about 3.0V while its falling threshold is a current in the inductor, SW equals the output
consistent 2.6V. voltage VOUT so the difference between VIN and
Internal Soft-Start VOUT can be used to charge the bootstrap
The soft-start is implemented to prevent the capacitor.
converter output voltage from overshooting At higher duty cycle operation condition, the time
during startup and short circuit recovery. When period available to the bootstrap charging is less
the chip starts, the internal circuitry generates a so the bootstrap capacitor may not be sufficiently
soft-start voltage (SS) ramping up from 0V to charged.
2.6V. When it is lower than the internal reference
In case the internal circuit does not have
(REF), SS overrides REF so the error amplifier
sufficient voltage and the bootstrap capacitor is
uses SS as the reference. When SS is higher
not charged, extra external circuitry can be used
than REF, REF regains control.
to ensure the bootstrap voltage is in the normal
Thermal Shutdown operational region. Refer to External Bootstrap
Thermal shutdown is implemented to prevent the Diode in Application section.
chip from operating at exceedingly high
The DC quiescent current of the floating driver is
temperatures. When the silicon die temperature
about 20µA. Make sure the bleeding current at
is higher than its upper threshold, it shuts down
the SW node is higher than this value, such that:
the whole chip. When the temperature is lower
than its lower threshold, the chip is enabled again. VO
IO   20A
(R1  R2)

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 MP4560 – 2A, 2MHz, 55V STEP-DOWN CONVERTER

Current Comparator and Current Limit Startup and Shutdown

The power MOSFET current is accurately sensed If both VIN and EN are higher than their
via a current sense MOSFET. It is then fed to the appropriate thresholds, the chip starts. The
high speed current comparator for the current reference block starts first, generating stable
mode control purpose. The current comparator reference voltage and currents, and then the
takes this sensed current as one of its inputs. internal regulator is enabled. The regulator
When the power MOSFET is turned on, the provides stable supply for the remaining
comparator is first blanked till the end of the turn- circuitries.
on transition to avoid noise issues. The
While the internal supply rail is up, an internal
comparator then compares the power switch
timer holds the power MOSFET OFF for about
current with the COMP voltage. When the
50µs to blank the startup glitches. When the
sensed current is higher than the COMP voltage,
internal soft-start block is enabled, it first holds its
the comparator output is low, turning off the
SS output low to ensure the remaining circuitries
power MOSFET. The cycle-by-cycle maximum
are ready and then slowly ramps up.
current of the internal power MOSFET is
internally limited. Three events can shut down the chip: EN low,
VIN low and thermal shutdown. In the shutdown
Short Circuit Protection
procedure, power MOSFET is turned off first to
When the output is shorted to the ground, the
avoid any fault triggering. The COMP voltage and
switching frequency is folded back and the
the internal supply rail are then pulled down.
current limit is reduced to lower the short circuit
current. When the voltage of FB is at zero, the Programmable Oscillator
current limit is reduced to about 50% of its full The MP4560 oscillating frequency is set by an
current limit. When FB voltage is higher than external resistor, RFREQ from the FREQ pin to
0.4V, current limit reaches 100%. ground. The value of RFREQ can be calculated
from:
In short circuit FB voltage is low, the SS is pulled
down by FB and SS is about 100mV above FB. 100000
In case the short circuit is removed, the output RFREQ (kΩ) = -5
fS (kHz)
voltage will recover at the SS pace. When FB is
high enough, the frequency and current limit To get fSW=500kHz, RFREQ=195kΩ.
return to normal values.

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 MP4560 – 2A, 2MHz, 55V STEP-DOWN CONVERTER

APPLICATION INFORMATION Inductor

The inductor is required to supply constant
COMPONENT SELECTION current to the output load while being driven by
Setting the Output Voltage the switched input voltage. A larger value
The output voltage is set using a resistive voltage inductor will result in less ripple current that will
divider from the output voltage to FB pin. The result in lower output ripple voltage. However, the
voltage divider divides the output voltage down to larger value inductor will have a larger physical
the feedback voltage by the ratio: size, higher series resistance, and/or lower
saturation current.
R2
VFB =VOUT  A good rule for determining the inductance to use
R1+R2
is to allow the peak-to-peak ripple current in the
Thus the output voltage is: inductor to be approximately 30% of the
maximum switch current limit. Also, make sure
R1+R2 that the peak inductor current is below the
VOUT =VFB 
R2 maximum switch current limit. The inductance
For example, the value for R2 can be 10kΩ. With value can be calculated by:
this value, R1 can be determined by: VOUT VOUT
L1=  (1- )
R1=12.5  (VOUT -0.8)(KΩ) fs  ΔIL VIN

For example, for a 3.3V output voltage, R2 is Where VOUT is the output voltage, VIN is the input
10kΩ, and R1 is 31.6kΩ. voltage, fS is the switching frequency, and ∆IL is
the peak-to-peak inductor ripple current.
Choose an inductor that will not saturate under
the maximum inductor peak current. The peak
inductor current can be calculated by:
VOUT  V 
ILP  ILOAD   1  OUT 
2  fS  L1  VIN 

Where ILOAD is the load current.

Table 1 lists a number of suitable inductors from
various manufacturers. The choice of which style
inductor to use mainly depends on the price vs.
size requirements and any EMI requirement.

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Table 1—Inductor Selection Guide

Inductance Max DCR Current Rating Dimensions
Part Number
(µH) (Ω) (A) L x W x H (mm3)
Wurth Electronics
7447789004 4.7 0.033 2.9 7.3x7.3x3.2
744066100 10 0.035 3.6 10x10x3.8
744771115 15 0.025 3.75 12x12x6
744771122 22 0.031 3.37 12x12x6
TDK
RLF7030T-4R7 4.7 0.031 3.4 7.3x6.8x3.2
SLF10145T-100 10 0.0364 3 10.1x10.1x4.5
SLF12565T-150M4R2 15 0.0237 4.2 12.5x12.5x6.5
SLF12565T-220M3R5 22 0.0316 3.5 12.5x12.5x6.5
Toko
FDV0630-4R7M 4.7 0.049 3.3 7.7x7x3
919AS-100M 10 0.0265 4.3 10.3x10.3x4.5
919AS-160M 16 0.0492 3.3 10.3x10.3x4.5
919AS-220M 22 0.0776 3 10.3x10.3x4.5

Output Rectifier Diode Choose a diode whose maximum reverse voltage

The output rectifier diode supplies the current to rating is greater than the maximum input voltage,
the inductor when the high-side switch is off. To and whose current rating is greater than the
reduce losses due to the diode forward voltage maximum load current. Table 2 lists example
and recovery times, use a Schottky diode. Schottky diodes and manufacturers.

Table 2—Diode Selection Guide

Voltage/
Diodes Current Manufacturer
Rating
B290-13-F 90V, 2A Diodes Inc.
B380-13-F 80V, 3A Diodes Inc.
CMSH2-100M 100V, 2A Central Semi
CMSH3-100MA 100V, 3A Central Semi

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 MP4560 – 2A, 2MHz, 55V STEP-DOWN CONVERTER

Input Capacitor VOUT  V 

ΔVOUT   1  OUT   R ESR
The input current to the step-down converter is fS  L  VIN 
discontinuous, therefore a capacitor is required to
supply the AC current to the step-down converter The characteristics of the output capacitor also
while maintaining the DC input voltage. Use low affect the stability of the regulation system. The
ESR capacitors for the best performance. MP4560 can be optimized for a wide range of
Ceramic capacitors are preferred, but tantalum or capacitance and ESR values.
low-ESR electrolytic capacitors may also suffice. Compensation Components
For simplification, choose the input capacitor with MP4560 employs current mode control for easy
RMS current rating greater than half of the compensation and fast transient response. The
maximum load current. The input capacitor (C1) system stability and transient response are
can be electrolytic, tantalum or ceramic. controlled through the COMP pin. COMP pin is
the output of the internal error amplifier. A series
When using electrolytic or tantalum capacitors, a capacitor-resistor combination sets a pole-zero
small, high quality ceramic capacitor, i.e. 0.1μF, combination to control the characteristics of the
should be placed as close to the IC as possible. control system. The DC gain of the voltage
When using ceramic capacitors, make sure that feedback loop is given by:
they have enough capacitance to provide
sufficient charge to prevent excessive voltage VFB
A VDC  R LOAD  G CS  A VEA 
ripple at input. The input voltage ripple caused by VOUT
capacitance can be estimated by:
Where AVEA is the error amplifier voltage gain,
I V  V  400V/V; GCS is the current sense
VIN  LOAD  OUT  1  OUT 
fS  C1 VIN  VIN  transconductance, 5.6A/V; RLOAD is the load
resistor value.
Output Capacitor
The output capacitor (C2) is required to maintain The system has two poles of importance. One is
the DC output voltage. Ceramic, tantalum, or low due to the compensation capacitor (C3), the
ESR electrolytic capacitors are recommended. output resistor of error amplifier. The other is due
Low ESR capacitors are preferred to keep the to the output capacitor and the load resistor.
output voltage ripple low. The output voltage These poles are located at:
ripple can be estimated by: G EA
f P1 
VOUT  V   1  2 π C 3  A VEA
VOUT   1  OUT    R ESR  
f S  L  VIN  
 8  f S  C2 
 1
fP2 
Where L is the inductor value and RESR is the 2 π C 2  R LOAD
equivalent series resistance (ESR) value of the
output capacitor. Where, GEA is the error amplifier
transconductance, 120μA/V.
In the case of ceramic capacitors, the impedance
at the switching frequency is dominated by the The system has one zero of importance, due to
capacitance. The output voltage ripple is mainly the compensation capacitor (C3) and the
caused by the capacitance. For simplification, the compensation resistor (R3). This zero is located
output voltage ripple can be estimated by: at:

VOUT  V  1
ΔVOUT   1  OUT  f Z1 
8  fS
2
 L  C2  VIN 
2 π C 3  R 3
In the case of tantalum or electrolytic capacitors,
the ESR dominates the impedance at the
switching frequency. For simplification, the output
ripple can be approximated to:

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 MP4560 – 2A, 2MHz, 55V STEP-DOWN CONVERTER

The system may have another zero of 2. Choose the compensation capacitor (C3) to
importance, if the output capacitor has a large achieve the desired phase margin. For
capacitance and/or a high ESR value. The zero, applications with typical inductor values, setting
due to the ESR and capacitance of the output the compensation zero, fZ1, below one forth of the
capacitor, is located at: crossover frequency provides sufficient phase
margin. Determine the C3 value by the following
1
f ESR  equation:
2π  C2  R ESR
4
C3 
In this case, a third pole set by the compensation 2 π R 3  f C
capacitor (C5) and the compensation resistor (R3)
is used to compensate the effect of the ESR zero 3. Determine if the second compensation
on the loop gain. This pole is located at: capacitor (C5) is required. It is required if the
ESR zero of the output capacitor is located at
1 less than half of the switching frequency, or the
fP3 
2 π C 5  R 3 following relationship is valid:
The goal of compensation design is to shape the 1 f
converter transfer function to get a desired loop  S
2π  C2  R ESR 2
gain. The system crossover frequency where the
feedback loop has the unity gain is important. If this is the case, then add the second
Lower crossover frequencies result in slower line compensation capacitor (C5) to set the pole fP3 at
and load transient responses, while higher the location of the ESR zero. Determine the C5
crossover frequencies could cause system value by the equation:
unstable. A good rule of thumb is to set the
crossover frequency to approximately one-tenth C 2  R ESR
C5 
of the switching frequency. R3

Table 3—Compensation Values for Typical High Frequency Operation

Output Voltage/Capacitor Combinations The switching frequency of MP4560 can be
programmed up to 2MHz by an external resistor.
VOUT C2 R3 C3 C6
L (µH) The minimum on time of MP4560 is about 100ns
(V) (µF) (kΩ) (pF) (pF)
1.8 4.7 33 32.4 680 None (typ). Pulse skipping operation can be seen more
2.5 4.7 - 6.8 22 26.1 680 None easily at higher switching frequency due to the
3.3 6.8 -10 22 68.1 220 None minimum on time.
5 15 - 22 33 47.5 330 None Since the internal bootstrap circuitry has higher
12 10 22 16 470 2 impedance, which may not be adequate to
To optimize the compensation components for charge the bootstrap capacitor during each
conditions not listed in Table 3, the following (1-D)×Ts charging period, an external bootstrap
procedure can be used. charging diode is strongly recommended if the
switching frequency is about 2MHz (see External
1. Choose the compensation resistor (R3) to set Bootstrap Diode section for detailed
the desired crossover frequency. Determine the implementation information).
R3 value by the following equation:
With higher switching frequencies, the inductive
2 π C 2  f C VOUT reactance (XL) of capacitor comes to dominate,
R3 
G EA  G CS VFB so that the ESL of input/output capacitor

Where fC is the desired crossover frequency.

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 MP4560 – 2A, 2MHz, 55V STEP-DOWN CONVERTER

determines the input/output ripple voltage at The bootstrap diode can be a low cost one such
higher switching frequency. As a result of that, as IN4148 or BAT54.
high frequency ceramic capacitor is strongly
5V
recommended as input decoupling capacitor and
output filtering capacitor for such high frequency
operation. BST

Layout becomes more important when the device MP4560

switches at higher frequency. It is essential to
SW
place the input decoupling capacitor, catch diode
and the MP4560 (VIN pin, SW pin and PGND) as
close as possible, with traces that are very short Figure 2—External Bootstrap Diode
and fairly wide. This can help to greatly reduce At no load or light load, the converter may
the voltage spike on SW node, and lower the EMI operate in pulse skipping mode in order to
noise level as well. maintain the output voltage in regulation. Thus
Try to run the feedback trace as far from the there is less time to refresh the BS voltage. In
inductor and noisy power traces as possible. It is order to have enough gate voltage under such
often a good idea to run the feedback trace on operating conditions, the difference of (VIN –VOUT
the side of the PCB opposite of the inductor with should be greater than 3V. For example, if the
a ground plane separating the two. The VOUT is set to 3.3V, the VIN needs to be higher
compensation components should be placed than 3.3V+3V=6.3V to maintain enough BST
closed to the MP4560. Do not place the voltage at no load or light load. To meet this
compensation components close to or under high requirement, EN pin can be used to program the
dv/dt SW node, or inside the high di/dt power input UVLO voltage to VOUT+3V.
loop. If you have to do so, the proper ground
plane must be in place to isolate those. Switching
loss is expected to be increased at high switching
frequency. To help to improve the thermal
conduction, a grid of thermal vias can be created
right under the exposed pad. It is recommended
that they be small (15mil barrel diameter) so that
the hole is essentially filled up during the plating
process, thus aiding conduction to the other side.
Too large a hole can cause ‘solder wicking’
problems during the reflow soldering process.
The pitch (distance between the centers) of
several such thermal vias in an area is typically
40mil.
External Bootstrap Diode
An external bootstrap diode may enhance the
efficiency of the regulator. In below cases, an
external BST diode is recommended from the 5V
to BST pin:
 There is a 5V rail available in the system;
 VIN is no greater than 5V;
 VOUT is between 3.3V and 5V;
This diode is also recommended for high duty
cycle operation (when VOUT/VIN > 65%)
applications.

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 MP4560 – 2A, 2MHz, 55V STEP-DOWN CONVERTER

TYPICAL APPLICATION CIRCUITS

C4
100nF

10
VIN 8,9 BST 1,2 VOUT
VIN SW 1.8V
D1

3 5
EN EN FB
MP4560

7 4
FREQ COMP
C3
GND 680pF
C5
6 NS

Figure 3—1.8V Output Typical Application Schematic

C4
100nF

10
VIN 8,9 BST 1,2 VOUT
VIN SW 5V
D1

3 5
EN EN FB
MP4560

7 4
FREQ COMP
C3
GND 330pF
C5
6 NS

Figure 4—5V Output Typical Application Schematic

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 MP4560 – 2A, 2MHz, 55V STEP-DOWN CONVERTER

PCB LAYOUT GUIDE 2) Bypass ceramic capacitors are suggested

to be put close to the VIN Pin.
PCB layout is very important to achieve stable
operation. It is highly recommended to duplicate 3) Ensure all feedback connections are short
EVB layout for optimum performance. and direct. Place the feedback resistors
and compensation components as close to
If change is necessary, please follow these the chip as possible.
guidelines and take Figure 5 for reference.
4) Route SW away from sensitive analog
1) Keep the path of switching current short areas such as FB.
and minimize the loop area formed by Input
cap, high-side MOSFET and external 5) Connect IN, SW, and especially GND
switching diode. respectively to a large copper area to cool
the chip to improve thermal performance
and long-term reliability.

MP4560

MP4560 Typical Application Circuit

TOP Layer Bottom Layer

MP4560DN Layout Guide

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 MP4560 – 2A, 2MHz, 55V STEP-DOWN CONVERTER

TOP Layer Bottom Layer

MP4560DQ Layout Guide

Figure 5―MP4560 Typical Application Circuit and PCB Layout Guide

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 MP4560 – 2A, 2MHz, 55V STEP-DOWN CONVERTER

PACKAGE INFORMATION
3mm x 3mm QFN10 (EXPOSED PAD)
2.90 0.30 1.45 PIN 1 ID
3.10 0.50 1.75 SEE DETAIL A
PIN 1 ID
MARKING
0.18
10 1
0.30

2.90 2.25
PIN 1 ID 2.55
3.10 0.50
INDEX AREA
BSC

6 5

TOP VIEW BOTTOM VIEW

PIN 1 ID OPTION A PIN 1 ID OPTION B

R0.20 TYP. R0.20 TYP.
0.80
1.00
0.20 REF
0.00
0.05

SIDE VIEW DETAIL A

NOTE:
2.90
1) ALL DIMENSIONS ARE IN MILLIMETERS.
0.70 1.70 2) EXPOSED PADDLE SIZE DOES NOT INCLUDE MOLD FLASH.
3) LEAD COPLANARITY SHALL BE 0.10 MILLIMETER MAX.
0.25 4) DRAWING CONFORMS TO JEDEC MO-229, VARIATION VEED-5.
5) DRAWING IS NOT TO SCALE.

2.50
0.50

RECOMMENDED LAND PATTERN

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 MP4560 – 2A, 2MHz, 55V STEP-DOWN CONVERTER

SOIC8 (EXPOSED PAD)

0.189(4.80) 0.124(3.15)
0.197(5.00) 0.136(3.45)
8 5

0.150(3.80) 0.228(5.80) 0.089(2.26)

PIN 1 ID 0.157(4.00) 0.244(6.20) 0.101(2.56)

1 4

TOP VIEW BOTTOM VIEW

SEE DETAIL "A"

0.051(1.30)
0.067(1.70)
0.0075(0.19)
SEATING PLANE
0.0098(0.25)
0.000(0.00)
0.013(0.33) 0.006(0.15)
0.020(0.51) SIDE VIEW
0.050(1.27)
BSC

FRONT VIEW 0.010(0.25)

x 45o
0.020(0.50)

GAUGE PLANE
0.010(0.25) BSC

0.024(0.61) 0.050(1.27)
0.016(0.41)
0o-8o 0.050(1.27)
0.063(1.60)
DETAIL "A"

0.103(2.62) 0.213(5.40)
NOTE:
1) CONTROL DIMENSION IS IN INCHES. DIMENSION IN
BRACKET IS IN MILLIMETERS.
2) PACKAGE LENGTH DOES NOT INCLUDE MOLD FLASH,
PROTRUSIONS OR GATE BURRS.
3) PACKAGE WIDTH DOES NOT INCLUDE INTERLEAD FLASH
0.138(3.51) OR PROTRUSIONS.
4) LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING)
SHALL BE 0.004" INCHES MAX.
RECOMMENDED LAND PATTERN 5) DRAWING CONFORMS TO JEDEC MS-012, VARIATION BA.
6) DRAWING IS NOT TO SCALE.

NOTICE: The information in this document is subject to change without notice. Users should warrant and guarantee that third
party Intellectual Property rights are not infringed upon when integrating MPS products into any application. MPS will not
assume any legal responsibility for any said applications.

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