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UNIT- 5
APPLICATION ICs

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CONTENTS

TECHNICAL TERMS

5.1 VOLTAGE REGULATORS

5.2 UNREGULATED POWER SUPPLY

5.3 VOLTAGE REGULATOR ICS - LM317

5.4 LM723/LM723C VOLTAGE REGULATOR

5.5 LM380 POWER AUDIO AMPLIFIER

5.6 VOLTAGE CONTROLLED OSCILLATOR ICL8038

5.7 ANALOG ISOLATION TECHNIQUES

5.7.1 AD210 3-Port Isolator

5.7.2 Motor Control Isolation Amplifier Application

5.7.3 AD215 Two-Port Isolator

5.8 OPTOCOUPLERS [4N25, 4N26, 4N27, 4N28]

5.9 QUESTION BANK

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TECHNICAL TERMS
1. Supply Voltage: The total supply voltage from V+ to V-.

2. Supply Current: The supply current required from the power supply to operate the

device, excluding load currents.

3. Frequency Range: The frequency range at the square wave output through which

circuit operation is guaranteed.

4. Sweep FM Range: The ratio of maximum frequency to minimum frequency which

can be obtained by applying a sweep voltage.

5. FM Linearity: The percentage deviation from the best fit straight line on the control

voltage versus output frequency curve.

6. Output Amplitude: The peak-to-peak signal amplitude appearing at the outputs.

7. Rise and Fall Times: The time required for the square wave output to change from

10% to 90%, or 90% to 10%, of its final value.

8. Triangle Waveform Linearity: The percentage deviation from the best fit straight

line on the rising and falling triangle waveform.

9. Total Harmonic Distortion: The total harmonic distortion at the sine wave output.

10.

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5.1 VOLTAGE REGULATORS

An ideal power supply maintains a constant voltage at its output terminals under all operating
conditions. The output voltage of a practical power supply changes with load generally
dropping as load current increases as shown in Figure 5.1.

Figure 5.1 Voltage Regulator

The terminal voltage when full load current is drawn is called full load voltage (VFL). The no
load voltage is the terminal voltage when zero current is drawn from the supply, that is, the
open circuit terminal voltage.

Power supply performance is measured in terms of percent voltage regulation, which

indicates its ability to maintain a constant voltage. It is defined as

V NL−V FL
V R= ×100 %
V FL

The Thevenin's equivalent of a power supply is shown in Figure 5.2. The Thevenin voltage is
the no-load voltage VNL and the Thevenin resistance is called the output resistance Ro. Let the
full load current be IFL. Therefore, the full load resistance RFL is given by

V FL
R FL=
I FL

From the equivalent circuit, we have

V FL=
( R FL
V
)
R FL −R0 NL

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Figure 5.2 Thevenin's equivalent

and the voltage regulation is given by

V R=
V NL− ( R FL
)V
RFL + R 0 NL
× 100 %

( R FL
)
V
R FL + R0 NL

V R=R 0
[ ]
I FL
V FL
×100 %

It is clear that the ideal power supply has zero output resistance.

5.2 UNREGULATED POWER SUPPLY

An unregulated power supply consists of a transformer (step down), a rectifier and a filter.
These power supplies are not good for some applications where constant voltage is required
irrespective of external disturbances. The main disturbances are:

As the load current varies, the output voltage also varies because of its poor regulation.

The dc output voltage varies directly with ac input supply. The input voltage may vary over a
wide range thus dc voltage also changes.

The dc output voltage varies with the temperature if semiconductor devices are used.

An electronic voltage regulator is essentially a controller used along with unregulated power
supply to stabilize the output dc voltage against three major disturbances

Load current (IL)

Supply voltage (Vi)

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Temperature (T)

Where,
Vi = unregulated dc voltage.
Vo = regulated dc voltage.

Figure 5.3 shows the basic block diagram of voltage regulator

Since the output dc voltage VLo depends on the input unregulated dc voltage Vi, load current
IL and the temperature t, then the change ΔV o in output voltage of a power supply can be
expressed as follows

VO = VO(Vi, IL, T)

Take partial derivative of VO, we get,

∂Vo ∂Vo ∂Vo

∆V o ∆ V i+ ∆ V i ∆ I L+ ∆T
∂Vi ∂ IL ∂T

∆ V o SV ∆ V i+ ∆ V i R L ∆ I L + ST ∆ T

SV gives variation in output voltage only due to unregulated dc voltage. R O gives the output
voltage variation only due to load current. S T gives the variation in output voltage only due to
temperature.

The smaller the value of the three coefficients, the better the regulations of power supply. The
input voltage variation is either due to input supply fluctuations or presence of ripples due to
inadequate filtering.

5.3 VOLTAGE REGULATOR ICS - LM317

The LM317 is an adjustable 3−terminal positive voltage regulator capable of
supplying in excess of 1.5A over an output voltage range of 1.2 V to 37 V. This voltage
regulator is exceptionally easy to use and requires only two external resistors to set the output

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voltage. Further, it employs internal current limiting, thermal shutdown and safe area
compensation, making it essentially blow−out proof. The LM317 serves a wide variety of
applications including local, on card regulation. This device can also be used to make a
programmable output regulator, or by connecting a fixed resistor between the adjustment and
output, the LM317 can be used as a precision current
Features
 Output Current in Excess of 1.5 A
 Output Adjustable between 1.2 V and 37 V
 Internal Thermal Overload Protection
 Internal Short Circuit Current Limiting Constant with Temperature
 Output Transistor Safe−Area Compensation
 Floating Operation for High Voltage Applications
 Available in Surface Mount D2PAK−3, and Standard 3−Lead Transistor Package
 Eliminates stocking many Fixed Voltages
 Pb−Free Packages are Available

Pin 1. Adust Pin 2. Vout Pin 3. Vin

Figure 5.4

**Cin is required if regulator is located an appreciable distance from power supply filter.
**CO is not needed for stability; however, it does improve transient response.
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Figure 5.5
Basic Circuit Operation
The LM317 is a 3−terminal floating regulator. In operation, the LM317 develops and
maintains a nominal 1.25 V reference (Vref) between its output and adjustment terminals. This
reference voltage is converted to a programming current (I PROG) by R1, and this constant
current flows through R2 to ground. [Figure 5.5]
The regulated output voltage is given by:

( )
V out =1.25V 1+
R2
R1
+ I Adj R 2

Since the current from the adjustment terminal (I Adj) represents an error term in the equation,
the LM317 was designed to control IAdj to less than 100 A and keep it constant (Figure 5.6).
To do this, all quiescent operating current is returned to the output terminal. This imposes the
requirement for a minimum load current. If the load current is less than this minimum, the
output voltage will rise. Since the LM317 is a floating regulator, it is only the voltage
differential across the circuit which is important to performance, and operation at high
voltages with respect to ground is possible.

Figure 5.6. Basic Circuit Configuration

Load Regulation
The LM317 is capable of providing extremely good load regulation, but a few precautions are
needed to obtain maximum performance. For best performance, the programming resistor
(R1) should be connected as close to the regulator as possible to minimize line drops which
effectively appear in series with the reference, thereby degrading regulation. The ground end

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of R2 can be returned near the load ground to provide remote ground sensing and improve
load regulation.

External Capacitors
A 0.1 µF disc or 1.0 µF tantalum input bypass capacitor (C in) is recommended to reduce the
sensitivity to input line impedance. The adjustment terminal may be bypassed to ground to
improve ripple rejection. This capacitor (C Adj) prevents ripple from being amplified as the
output voltage is increased. A 10 µF capacitor should improve ripple rejection about 15 dB at
120 Hz in a 10 V application. Although the LM317 is stable with no output capacitance, like
any feedback circuit, certain values of external capacitance can cause excessive ringing. An
output capacitance (CO) in the form of a 1.0 µF tantalum or 25 µF aluminum electrolytic
capacitor on the output swamps this effect and insures stability.

Protection Diodes
When external capacitors are used with any IC regulator it is sometimes necessary to add
protection diodes to prevent the capacitors from discharging through low current points into
the regulator. Figure 5.7 shows the LM317 with the recommended protection diodes for
output voltages in excess of 25 V or high capacitance values (C O > 25 µF, CAdj > 10 µF).
Diode D1 prevents CO from discharging thru the IC during an input short circuit. Diode D 2
protects against capacitor CAdj discharging through the IC during an output short circuit. The
combination of diodes D1 and D2 prevents CAdj from discharging through the IC during an
input short circuit.

Figure 5.7. Power Supply with Adjustable Current Limit and Output Voltage

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5.4 LM723/LM723C VOLTAGE REGULATOR

Features
• 150 mA output current without external pass
• Input voltage 40V max transistor
• Output voltage adjustable from 2V to 37V
• Output currents in excess of 10A possible by
• Can be used as either a linear or a switching adding external transistors regulator

Description
The LM723/LM723C (Figure 5.8) is a voltage regulator designed primarily for series
regulator applications. By itself, it will supply output currents up to 150 mA; but external
transistors can be added to provide any desired load current. The circuit features extremely
low standby current drain, and provision is made for either linear or fold back current
limiting. The LM723/LM723C is also useful in a wide range of other applications such as a
shunt regulator, a current regulator or a temperature controller. The LM723C is identical to
the LM723 except that the LM723C has its performance guaranteed over a 0°C to +70°C
temperature range, instead of −55°C to +125°C.

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Figure 5.8.Top View

Equivalent Circuit

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Figure 5.9

APPLICATION
Voltage Regulator

Figure 5.10 Voltage Regulator

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External Pass Transistor

Figure 5.11 External Pass Transistor

Fold back Current Limiting

Figure 5.12 Fold back Current Limiting

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Floating Regulator

Figure 5.13 Positive Floating Regulator

Switching Regulator

Figure 5.14 Positive Switching Regulator

Shunt Regulator

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Figure 5.16 Shunt Regulator

5.5 LM380 POWER AUDIO AMPLIFIER

General Description
The LM380 is a power audio amplifier as shown in Fig 5.17,18,19 for consumer application.
In order to hold system cost to a minimum, gain is internally fixed at 34 dB. A unique input
stage allows inputs to be ground referenced. The output is automatically self centering to one
half the supply voltage. The output is short circuit proof with internal thermal limiting. The
package outline is standard dual-in-line. A copper lead frame is used with the center three
pins on either side comprising a heat sink. This makes the device easy to use in standard p-c
layout. Uses include simple phonograph amplifiers, intercoms, line drivers, teaching machine
outputs, alarms, ultrasonic drivers, TV sound systems, AM-FM radio, small servo drivers,
power converters, etc. A selected part for more power on higher supply voltages is available
as the LM384.
Features
 Wide supply voltage range
 Low quiescent power drain
 Voltage gain fixed at 50
 High peak current capability

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 Input referenced to GND

 High input impedance
 Low distortion
 Quiescent output voltage is at one-half of the supply voltage
 Standard dual-in-line package

Figure 5.17

Figure 5.18

LM380 circuit description: It is connected of 4 stages, (i) PNP emitter follower (ii)
Different amplifier (iii) Common emitter (iv) Emitter follower

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Figure 5.19

(i) PNP Emitter follower: The input stage is emitter follower composed of PNP transistors
Q1 & Q2 which drives the PNP Q3-Q4 differential pair.
The choice of PNP input transistors Q1 & Q2 allows the input to be referenced to ground i.e.,
the input can be direct coupled to either the inverting & non-inverting terminals of the
amplifier.
(ii) Differential Amplifier: The current in the PNP differential pair Q 3-Q4 is established by
Q7, R3 & +V. The current mirror formed by transistor Q 7, Q8 & associated resistors then
establishes the collector current of Q9. Transistor Q5 & Q6 constitute of collector loads for
the PNP differential pair. The output of the differential amplifier is taken at the junction of Q 4
& Q6 transistors & is applied as an input to the common emitter voltage gain.
(iii) Common Emitter: Common Emitter amplifier stage is formed by transistor Q 9 with D1,
D2 & Q8 as a current source load. The capacitor C between the base & collector of Q 9
provides internal compensation & helps to establish the upper cutoff frequency of 100 KHz.
Since Q7 & Q8 form a current mirror, the current through D 1 & D2 is approximately the same
as the current through R3. D1 & D2 are temperature compensating diodes for transistors Q 10 &
Q11 in that D1 & D2 have the same characteristics as the base-emitter junctions of Q 11.
Therefore the current through Q10 & (Q11-Q12) is approximately equal to the current through
diodes D1 & D2.

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(iv) (Output stage) - Emitter follower: Emitter follower formed by NPN transistor Q 10 & Q11.
The combination of PNP transistor Q11 & NPN transistor Q12 has the power capability of
NPN transistors but the characteristics of a PNP transistor.
The negative dc feedback applied through R 5 balances the differential amplifier so that the dc
output voltage is stabilized at +V/2; To decouple the input stage from the supply voltage +V,
by pass capacitor in order of micro farad should be connected between the by pass terminal
(pin 1) & ground (pin 7). The overall internal gain of the amplifier is fixed at 50. However
gain can be increased by using positive feedback.
(i) Audio Power Amplifier:

Figure 5.20

The circuit shown in Figure 5.20 is a very simple audio power amplifier.

The main component of this circuit is the LM380 audio amplifier. The simplicity of this
circuit is made possible by the LM380's minimal requirements for external components, since
it is already internally equipped with the necessary biasing, compensation, and gain circuits
for audio amplification. The circuit in Figure 1 uses the LM380 in non-inverting mode, with
the inverting input left open (the inverting input may also be tied to ground, either directly or
through a resistor or capacitor). C2 is used to decouple Vcc from ground. The optional RC
circuit at the output (pin 8) is used for added stability, i.e., to eliminate oscillations in an RF-
sensitive application.

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Figure 5.21

Figure 5.21 shows how the circuit in Figure 5.22 can have a variable gain simply by
connecting a potentiometer across the inputs of the LM380. Rv is varied to adjust the gain of
the amplifier.

5.6 VOLTAGE CONTROLLED OSCILLATOR ICL8038

The ICL8038 (Figure5.23) waveform generator is a monolithic integrated circuit capable of
producing high accuracy sine, square, triangular, saw tooth and pulse waveforms with a
minimum of external components. The frequency (or repetition rate) can be selected
externally from 0.001Hz to more than 300kHz using either resistors or capacitors, and
frequency modulation and sweeping can be accomplished with an external voltage. The
ICL8038 is fabricated with advanced monolithic technology, using Schottky barrier diodes
and thin film resistors, and the output is stable over a wide range of temperature and supply
variations. These devices may be interfaced with phase locked loop circuitry to reduce
temperature drift to less than 250ppm/°C.

Features
• Low Frequency Drift with Temperature . . . . . 250ppm/°C
• Low Distortion . . . . . . . . . . . . . . . 1% (Sine Wave Output)
• High Linearity . . . . . . . . . . .0.1% (Triangle Wave Output)
• Wide Frequency Range . . . . . . . . . . . .0.001Hz to 300kHz
• Variable Duty Cycle . . . . . . . . . . . . . . . . . . . . . 2% to 98%
• High Level Outputs. . . . . . . . . . . . . . . . . . . . . . TTL to 28V
• Simultaneous Sine, Square, and Triangle Wave Outputs
• Easy to Use - Just a Handful of External Components Required

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Pin out Diagram

Figure 5.23

Functional Diagram

Figure 5.24. Test circuit

An external capacitor C is charged and discharged by two current sources. Current source #2
is switched on and off by a flip-flop, while current source #1 is on continuously. Assuming
that the flip-flop is in a state such that current source #2 is off, and the capacitor is charged
with a current I, the voltage across the capacitor rises linearly with time. When this voltage
reaches the level of comparator #1 (set at 2/3 of the supply voltage), the flip-flop is triggered,
changes states, and releases current source #2. This current source normally carries a current

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2I, thus the capacitor is discharged with a net-current I and the voltage across it drops linearly
with time. When it has reached the level of comparator #2 (set at 1/3 of the supply voltage),
the flip-flop is triggered into its original state and the cycle starts again. Four waveforms are
readily obtainable from this basic generator circuit. With the current sources set at I and 2I
respectively, the charge and discharge times are equal. Thus a triangle waveform is created
across the capacitor and the flip-flop produces a square wave. Both waveforms are fed to
buffer stages and are available at pins 3 and 9. The levels of the current sources can, however,
be selected over a wide range with two external resistors. Therefore, with the two currents set
at values different from I and 2I, an asymmetrical saw tooth appears at Terminal 3 and pulses
with a duty cycle from less than 1% to greater than 99% are available at Terminal 9. The sine
wave is created by feeding the triangle wave into a nonlinear network (sine converter). This
network provides decreasing shunt impedance as the potential of the triangle moves toward
the two extremes.

Waveform Timing
The symmetry of all waveforms can be adjusted with the external timing resistors. Two
possible ways to accomplish this are shown in Figure 5.27. Best results are obtained by
keeping the timing resistors RA and RB separate (A). RA controls the rising portion of the
triangle and sine wave and the 1 state of the square wave. The magnitude of the triangle
waveform is set at 1/3 VSUPPLY; therefore the rising portion of the triangle is,
R A ×C
T 1=
0.66

The falling portion of the triangle and sine wave and the 0 state of the square wave is
R A RB C
T 2=
0.66(R A −R B )

Thus a 50% duty cycle is achieved when RA = RB. If the duty cycle is to be varied over a
small range about 50% only, the connection shown in Figure 3B is slightly more convenient.
A 1kΩ potentiometer may not allow the duty cycle to be adjusted through 50% on all devices.
If a 50% duty cycle is required, a 2kΩ or 5kΩ potentiometer should be used. With two
separate timing resistors, the frequency is given by:
1
f=
T 1 +T 2

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Figure 5.25a. Square wave duty cycle - 50% Figure 5.25b. Square wave duty cycle - 80%

Figure 5.25 Phase relationship of waveforms

Figure 5.26a Figure 5.26b

Figure 5.27 Possible connections for the external timing resistors

Neither time nor frequency is dependent on supply voltage, even though none of the voltages
are regulated inside the integrated circuit. This is due to the fact that both currents and
thresholds are direct, linear functions of the supply voltage and thus their effects cancel.
Reducing Distortion
To minimize sine wave distortion the 82kΩ resistor between pins 11 and 12 is best made
variable. With this arrangement distortion of less than 1% is achievable. To reduce this even
further, two potentiometers can be connected as shown in Figure 5.28; this configuration
allows a typical reduction of sine wave distortion close to 0.5%.

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Figure 5.28 Connection to achieve minimum sine wave distortion

Selecting RA, RB and C

For any given output frequency, there is a wide range of RC combinations that will work,
however certain constraints are placed upon the magnitude of the charging current for
optimum performance. At the low end, currents of less than 1μA are undesirable because
circuit leakages will contribute significant errors at high temperatures. At higher currents (I >
5mA), transistor betas and saturation voltages will contribute increasingly larger errors.
Optimum performance will, therefore, be obtained with charging currents of 10μA to 1mA. If
pins 7 and 8 are shorted together, the magnitude of the charging current due to RA can be
calculated from:
I =R1 ׿ ¿

R1 and R2 are shown in the Detailed Schematic. A similar calculation holds for R B. The
capacitor value should be chosen at the upper end of its possible range.

Waveform out Level Control and Power Supplies

The waveform generator can be operated either from a single power supply (10V to 30V) or a
dual power supply (±5V to ±15V). With a single power supply the average levels of the
triangle and sine wave are at exactly one-half of the supply voltage, while the square wave
alternates between V+ and ground. A split power supply has the advantage that all
waveforms move symmetrically about ground. The square wave output is not committed. A
load resistor can be connected to a different power supply, as long as the applied voltage

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remains within the breakdown capability of the waveform generator (30V). In this way, the
square wave output can be made TTL compatible (load resistor connected to +5V) while the
waveform generator itself is powered from a much higher voltage.
Frequency Modulation and Sweeping
The frequency of the waveform generator is a direct function of the DC voltage at Terminal 8
(measured from V+). By altering this voltage, frequency modulation is performed. For small
deviations (e.g. ±10%) the modulating signal can be applied directly to pin 8, merely
providing DC decoupling with a capacitor as shown in Figure 5.29. An external resistor
between pins 7 and 8 is not necessary, but it can be used to increase input impedance from
about 8kΩ (pins 7 and 8 connected together), to about (R + 8kΩ). For larger FM deviations or
for frequency sweeping, the modulating signal is applied between the positive supply voltage
and pin 8 (Figure 5B). In this way the entire bias for the current sources is created by the
modulating signal, and a very large (e.g. 1000:1) sweep range is created (f = Minimum at
VSWEEP = 0, i.e., Pin 8 = V+). Care must be taken, however, to regulate the supply voltage; in
this configuration the charge current is no longer a function of the supply voltage (yet the
trigger thresholds still are) and thus the frequency becomes dependent on the supply voltage.
The potential on Pin 8 may be swept down from V+ by (1/3 VSUPPLY - 2V).

Figure 5.29 Connections for frequency sweep

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Figure 5.30 Strobe tone burst generator

To obtain a 1000:1 Sweep Range on the ICL8038 the voltage across external resistors RA
and RB must decrease to nearly zero. This requires that the highest voltage on control Pin 8
exceed the voltage at the top of RA and RB by a few hundred mV. The Circuit of Figure 8
achieves this by using a diode to lower the effective supply voltage on the ICL8038. The
large resistor on pin 5 helps reduce duty cycle variations with sweep. The linearity of input
sweep voltage versus output frequency can be significantly improved by using an op amp as
shown in Figure 5.32.

Figure 5.31 Waveform generator used as stable VCO in a phase-locked loop

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Figure 5.32 Linear voltage controlled oscillator

Use in Phase Locked Loops

Its high frequency stability makes the ICL8038 an ideal building block for a phase locked
loop as shown in Figure 5.31.In this application the remaining functional blocks, the phase
detector and the amplifier can be formed by a number of available ICs (e.g., MC4344,
NE562). In order to match these building blocks to each other, two steps must be taken. First,
two different supply voltages are used and the square wave output is returned to the supply of
the phase detector. This assures that the VCO input voltage will not exceed the capabilities of
the phase detector. If a smaller VCO signal is required, a simple resistive voltage divider is
connected between pin 9 of the waveform generator and the VCO input of the phase detector.
Second, the DC output level of the amplifier must be made compatible to the DC level
required at the FM input of the waveform generator (pin 8, 0.8V +). The simplest solution here
is to provide a voltage divider to V+ (R1, R2 as shown) if the amplifier has a lower output
level, or to ground if its level is higher. The divider can be made part of the low-pass filter.
This application not only provides for a free-running frequency with very low temperature
drift, but is also has the unique feature of producing a large reconstituted sine wave signal
with a frequency identical to that at the input.

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5.7 ANALOG ISOLATION TECHNIQUES

There are many applications where it is desirable, or even essential, for a sensor to have no
direct ("galvanic") electrical connection with the system to which it is supplying data. This
might be in order to avoid the possibility of dangerous voltages or currents from one half of
the system doing damage in the other, or to break an intractable ground loop. Such a system
is said to be "isolated", and the arrangement that passes a signal without galvanic connections
is known as an isolation barrier.

Just as interference, or unwanted information, may be coupled by electric or magnetic

fields, or by electromagnetic radiation, these phenomena may be used for the transmission of
wanted information in the design of isolated systems. The most common isolation amplifiers
use transformers, which exploit magnetic fields, and another common type uses small high
voltage capacitors, exploiting electric fields.

Optoisolators, which consist of an LED and a photocell, provide isolation by using

light, a form of electromagnetic radiation. Different isolators have differing performance:
some are sufficiently linear to pass high accuracy analog signals across an isolation barrier.
With others, the signal may need to be converted to digital form before transmission for
accuracy is to be maintained.

Transformers are capable of analog accuracy of 12-16 bits and bandwidths up to

several hundred kHz, but their maximum voltage rating rarely exceeds 10 kV, and is often
much lower.

Capacitively-coupled isolation amplifiers have lower accuracy, perhaps 12-bits

maximum, lower bandwidth, and lower voltage ratings—but they are low cost. Optical
isolators are fast and cheap, and can be made with very high voltage ratings (4 -7 kV is one of
the more common ratings), but they have poor analog linearity, and are not usually suitable
for direct coupling of precision analog signals.

5.7.1 AD210 3-Port Isolator

In a basic two-port form of isolator, the output and power circuits are not isolated
from one another. A three-port isolator (input, power, output) as shown figure 5.33 below.

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Note that in this diagram, the input circuits, output circuits, and power source are all isolated
from one another. This figure represents the circuit architecture of a self-contained isolator,
the AD210.
An isolator of this type requires power from a two-terminal dc power supply (PWR, PWR
COM). An internal oscillator (50 kHz) converts the dc power to ac, which is transformer-
coupled to the shielded input section, then converted to dc for the input stage and the
auxiliary power output.

Figure 5.33

The ac carrier is also modulated by the input stage amplifier output, transformer-coupled to
the output stage, demodulated by a phase-sensitive demodulator (using the carrier as the
reference), filtered, and buffered using isolated dc power derived from the carrier.
The AD210 allows the user to select gains from 1 to 100, using external resistors with the
input section op amp. Bandwidth is 20 kHz, and voltage isolation is 2500 V rms (continuous)
and ± 3500 V peak (continuous). The AD210 is a 3-port isolation amplifier, thus the power
circuitry is isolated from both the input and the output stages and may therefore be connected
to either (or to neither), without change in functionality. It uses transformer isolation to
achieve 3500 V isolation with 12-bit accuracy.

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5.7.2 Motor Control Isolation Amplifier Application

A typical isolation amplifier application using the AD210 as shown figure 5.34 below.
The AD210 is used with an AD620 instrumentation amplifier in a current-sensing system for
motor control. The input of the AD210, being isolated, can be directly connected to a 110 or
230 V power line without protection being necessary. The input section's isolated ±15 V
powers the AD620, which senses the voltage drop in a small value current sensing resistor.
The AD210 input stage op amp is simply connected as a unity-gain follower, which
minimizes its error contribution. The 110 or 230 V rms common-mode voltage is ignored by
this isolated system.

Figure 5.34 Motor Control Current Sensing Application

Within this system the AD620 preamp is used as the system scaling control point, and will
produce and output voltage proportional to motor current, as scaled by the sensing resistor
value and gain as set by the AD620's RG. The AD620 also improves overall system accuracy,
as the AD210 VOS is 15 mV, versus the AD620's 30 µV (with less drift also). Note that if
higher dc offset and drift are acceptable, the AD620 may be omitted and the AD210
connected at a gain of 100.
Due to the nature of this type of carrier-operated isolation system, there will be certain
operating situations where some residual ac carrier component will be superimposed upon the

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recovered output dc signal. When this occurs, a low impedance passive RC filter section
following the output stage may be used (if the following stage has a high input impedance,
i.e., non-loading to this filter). Note that will be the case for many high input impedance
sampling ADCs, which appear essentially as a small capacitor. A 150 Ω resistance and 1 nF
capacitor will provide a corner frequency of about 1 kHz. Note also that the capacitor should
be a film type for low errors, such as polypropylene.

5.7.3 AD215 Two-Port Isolator

The AD215 is a high speed, two-port isolation amplifier, designed to isolate and
amplify wide bandwidth analog signals. The innovative circuit and transformer design of the
AD215 ensures wide-band dynamic characteristics, while preserving dc performance
specifications. An AD215 block diagram as shown figure 5.35.

Figure 5.35 AD215 Low Distortion Isolation Amplifier

The AD215 provides complete galvanic isolation between the input and output of the device,
which also includes the user-available front-end isolated bipolar power supply. The
functionally complete design, powered by a ±15 V dc supply on the output side, eliminates
the need for a user supplied isolated dc/dc converter. This permits the designer to minimize
circuit overhead and reduce overall system design complexity and component costs.
The design of the AD215 emphasizes maximum flexibility and ease of use in a broad range
of applications where fast analog signals must be measured under high common-mode
voltage (CMV) conditions.

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The AD215 has a ±10 V input/output range, a specified gain range of 1 V/V to 10 V/V, a
buffered output with offset trim and a user-available isolated front end power supply which
produces ±15 V dc at ±10 mA.

5.8 OPTOCOUPLERS [4N25, 4N26, 4N27, 4N28]

Figure 5.36

PIN 1. ANODE
2. CATHODE
3. NO CONNECTION
4. EMITTER
5. COLLECTOR
6. BASE

Description
The general purpose optocouplers consist of a gallium arsenide infrared emitting diode riving
a silicon phototransistor in a 6-pin dual in-line package as shown figure 5.36.

Features
• Also available in white package by specifying -M suffix, eg. 4N25-M
• UL recognized (File # E90700)
• VDE recognized (File # 94766)
- Add option V for white package (e.g., 4N25V-M)
- Add option 300 for black package (e.g., 4N25.300)

Operation

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Both optocouplers and optoisolators allow the transfer of signals and data from one
system to another within a piece of electronic equipment without a direct electrical
connection. This is done optically by using a beam of light to an optical receiver in a single
package with a light-conducting medium between the emitter and detector. This allows the
total electrical isolation of electronic circuits while transmitting information from one voltage
potential to another. In all optocouplers and optoisolators, input signals are converted to a
pulse of light from an LED. This pulse of light is transmitted to a silicon photosensor.
The photosensor can be either analog or digital depending on the type of input signal to be
transferred across the device. When an application requires an analog signal, such as for 4 to
20 mA, the photosensor can be either a photodiode or phototransistor. Both of these devices
provide an analog output signal that can be used for a variety of analog applications.
An analog response is required for those applications where the amount of signal is critical to
the operation of the system. The amount of current on the output of the device referenced to
the amount of light into the LED is called the current transfer ratio (CTR), the output current
divided by the input current. CTR values may vary from 10% to over 5,000%, depending on
the gain of the system. Typically the lower the CTR, the faster the rise and fall times.

Applications
• Power supply regulators
• Digital logic inputs
• Microprocessor inputs

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QUESTION BANK
PART-A
1. What is a voltage regulator?
2. Give the classification of voltage regulators:
3. What is a linear voltage regulator?
4. What is a switching regulator?
9. Define line regulation.
10. Define load regulation.
11. What is meant by current limiting?
12. Give the drawbacks of linear regulators:
13. What is the advantage of switching regulators?
14. What is an opto-coupler IC?
15. What are the types of optocouplers?
16. Give two examples of IC optocouplers?
17. Mention the advantages of opto-couplers:
18. Mention the advantages of opto-couplers:
19. What is an isolation amplifier?
. What are the features of isolation amplifier?
21. What is LM380?
22. What are the features of MA78s40?
PART-B
1. Explain i) Oscillation amplifier.ii) Voltage regulator (16)
2. Draw and explain the functional block diagram of a 723 regulator. (16)
3. Draw the block diagram of the function generator in IC 8038 (or) any other equivalent and
explain its operation. (16)
4. Write an explanatory note on opto-couplers. (16)
5. Explain in detail about the 380 power amplifier. (16)

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