MIC27600

36V, 7A Hyper Speed Control™

Synchronous DC-DC Buck Regulator

SuperSwitcher II™

General Description Features

The Micrel MIC27600 is a constant-frequency, synchronous • Hyper Speed Control™ architecture enables
buck regulator featuring a unique digitally-modified adaptive - High Delta V operation (VIN = 36V and VOUT = 0.8V)
on-time control architecture. The MIC27600 operates over an - Small output capacitance
input supply range of 4.5V to 36V and provides a regulated • 4.5V to 36V voltage input
output of up to 7A of output current. The output voltage is
• Adjustable output from 0.8V to 5.5V (VHSD ≤ 28V)
adjustable down to 0.8V with a guaranteed accuracy of ±1%,
and the device operates at a switching frequency of 300kHz. • Adjustable output from 0.8V to 3.6V (VHSD ≤ 36V)
Micrel’s Hyper Speed Control™ architecture allows for ultra- • ±1% FB accuracy
fast transient response while reducing the output capacitance • Any Capacitor™ Stable - Zero-ESR to high-ESR
and also makes (High VIN)/(Low VOUT) operation possible. • 7A output current capability, up to 95% efficiency
This digitally modified adaptive tON ripple control architecture • 300kHz switching frequency
combines the advantages of fixed-frequency operation and • Internal compensation, 6ms Internal soft-start
fast transient response in a single device.
• Foldback current-limit and “hiccup” mode short-circuit
The MIC27600 offers a full suite of protection features to protection
ensure protection of the IC during fault conditions. These
• Thermal shutdown
include undervoltage lockout to ensure proper operation
under power-sag conditions, internal soft-start to reduce • Supports safe startup into a pre-biased load
inrush current, foldback current limit, “hiccup” mode short- • –40°C to +125°C junction temperature range
circuit protection and thermal shutdown. • 28-pin 5mm × 6mm MLF® package
All support documentation can be found on Micrel’s web
site at: www.micrel.com. Applications
• Distributed power systems
• Communications/networking infrastructure
• Set-top box, gateways and routers
• Printers, scanners, graphic cards and video cards
___________________________________________________________________________________________________________
Typical Application

Efficiency (VIN = 28V)

vs. Output Current
95
5.0V
90 3.3V
2.5V
85
EFFICIENCY (%)

1.8V
1.5V
80 1.2V
1.0V
75 0.9V
0.8V
70

65

60
0 1 2 3 4 5 6 7 8 9
OUTPUT CURRENT (A)

Hyper Speed Control, SuperSwitcher II and Any Capacitor are trademarks of Micrel, Inc.
MLF and MicroLeadFrame are registered trademarks of Amkor Technology, Inc.
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com

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Ordering Information
Part Number Voltage Switching Frequency Junction Temperature Range Package Lead Finish
MIC27600YJL Adjustable 300kHz −40°C to +125°C 28-pin 5mm × 6mm MLF® Pb-Free

Pin Configuration

28-Pin 5mm × 6mm MLF® (YJL)

Pin Description
Pin
Pin Name Pin Function
Number
13, 14, 15, High-Side N-internal MOSFET Drain Connection (Input): The PVIN operating voltage range is from
16, 17, 18, PVIN 4.5V to 36V. Input capacitors between the PVIN pins and the power ground (PGND) are required and
19 keep the connection short.
Enable (Input): A logic level control of the output. The EN pin is CMOS-compatible. Logic high or
24 EN floating = enable, logic low = shutdown. In the off state, the VDD supply current of the device is
reduced (typically 0.7mA). Do not pull the EN pin above the VDD supply.
Feedback (Input): Input to the transconductance amplifier of the control loop. The FB pin is regulated
25 FB to 0.8V. A resistor divider connecting the feedback to the output is used to adjust the desired output
voltage.
Signal ground. SGND must be connected directly to the ground planes. Do not route the SGND pin to
26 SGND
the PGND Pad on the top layer, see PCB layout guidelines for details.
VDD Bias (Input): Power to the internal reference and control sections of the MIC27600. The VDD
operating voltage range is from 4.5V to 5.5V. A 2.2µF ceramic capacitor from the VDD pin to the
27 VDD
PGND pin must be placed next to the IC. VDD must be powered up at the same time or after VIN to
make the soft-start function correctly.
Power Ground. PGND is the ground path for the MIC27600 buck converter power stage. The PGND
2, 5, 6, 7, pin connects to the sources of low-side N-Channel internal MOSFETs, the negative terminals of input
PGND
8, 21 capacitors, and the negative terminals of output capacitors. The loop for the power ground should be
as small as possible and separate from the signal ground (SGND) loop.
Current Sense (Input): High current output driver return. The CS pin connects directly to the switch
node. Due to the high-speed switching on this pin, the CS pin should be routed away from sensitive
22 CS
nodes. CS pin also senses the current by monitoring the voltage across the low-side internal
MOSFET during OFF-time.

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Pin Description (Continued)

Pin
Pin Name Pin Function
Number
Boost (Output): Bootstrapped voltage to the high-side N-channel internal MOSFET driver. A Schottky
20 BST diode is connected between the VDD pin and the BST pin. A boost capacitor of 0.1μF is connected
between the BST pin and the SW pin.
4, 9, 10, Switch Node (Output): Internal connection for the high-side MOSFET source and low-side MOSFET
SW
11, 12 drain.
23 VIN Power Supply Voltage (Input): Requires bypass capacitor to SGND.
1, 3, 28 NC No Connect.

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Absolute Maximum Ratings(1, 2) Operating Ratings(3)

PVIN to PGND................................................ −0.3V to +38V Supply Voltage (PVIN, VIN)................................. 4.5V to 36V
VIN to PGND ....................................................−0.3V to PVIN Bias Voltage (VDD)............................................ 4.5V to 5.5V
VDD to PGND ................................................... −0.3V to +6V Enable Input (VEN) ................................................. 0V to VDD
VSW, VCS to PGND .............................. −0.3V to (PVIN +0.3V) Junction Temperature (TJ) ........................ −40°C to +125°C
VBST to VSW ........................................................ −0.3V to 6V Maximum Power Dissipation......................................Note 4
VBST to PGND .................................................. −0.3V to 44V Package Thermal Resistance(4)
VEN to PGND ...................................... −0.3V to (VDD + 0.3V) 5mm x 6mm MLF® (θJA) ....................................36°C/W
VFB to PGND....................................... −0.3V to (VDD + 0.3V)
PGND to SGND ........................................... −0.3V to +0.3V
Junction Temperature .............................................. +150°C
Storage Temperature (TS).........................−65°C to +150°C
Lead Temperature (soldering, 10sec)........................ 260°C

Electrical Characteristics(5)
PVIN = VIN = 12V, VDD = 5V; VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate −40°C ≤ TJ ≤ +125°C.
Parameter Condition Min. Typ. Max. Units
Power Supply Input
Input Voltage Range (VIN, PVIN) 4.5 36 V
VDD Bias Voltage
Operating Bias Voltage (VDD) 4.5 5 5.5 V
Under-Voltage Lockout Trip Level VDD Rising 2.4 2.7 3.2 V
UVLO Hysteresis 50 mV
Quiescent Supply Current VFB = 1.5V 1.4 3 mA
VDD = VBST = 5.5V, VIN = 36V 0.7 2 mA
Shutdown Supply Current
SW = unconnected, VEN = 0V
Reference
0°C ≤ TJ ≤ 85°C (±1.0%) 0.792 0.8 0.808
Feedback Reference Voltage V
−40°C ≤ TJ ≤ 125°C (±1.5%) 0.788 0.8 0.812
Load Regulation IOUT = 0A to 7A 0.2 %
Line Regulation VIN = (VOUT + 3.0V) to 36V 0.1 %
FB Bias Current VFB = 0.8V 5 nA
DC-DC Converter
3.0V ≤ VHSD ≤ 28V 0.8 5.5
Output Voltage Range V
3.0V ≤ VHSD ≤ 36V 0.8 3.6
Enable Control
EN Logic Level High 4.5V < VDD < 5.5V 1.2 0.85 V
EN Logic Level Low 4.5V < VDD < 5.5V 0.78 0.4 V
EN Bias Current VEN = 0V 50 µA
Notes:
1. Exceeding the absolute maximum rating may damage the device.
2. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5kΩ in series with 100pF.
3. The device is not guaranteed to function outside operating range.
4. PD(MAX) = (TJ(MAX) – TA)/ θJA, where θJA depends upon the printed circuit layout. See “Applications Information.”
5. Specification for packaged product only.

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Electrical Characteristics(5) (Continued)

PVIN = VIN = 12V, VDD = 5V; VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate −40°C ≤ TJ ≤ +125°C.
Parameter Condition Min. Typ. Max. Units
Oscillator
Switching Frequency (6) 225 300 375 kHz
(7)
Maximum Duty Cycle VFB = 0V 87 %
Minimum Duty Cycle VFB > 0.8V 0 %
Minimum Off-time 360 ns
Soft-Start
Soft-Start time 6 ms
Short Circuit Protection
Current-Limit Threshold VFB = 0.8V 7.7 15 A
Short-Circuit Current VFB = 0V 6 A
Internal FETs
Top-MOSFET RDS (ON) ISW = 1A 25 mΩ
Bottom-MOSFET RDS (ON) ISW = 1A 10 mΩ
SW Leakage Current VIN = 36V, VSW = 36V, VEN = 0V, VBST = 41V 60 µA
VIN Leakage Current VIN = 36V, VSW = 0V, VEN = 0V, VBST = 41V 25 µA
Thermal Protection
Over-Temperature Shutdown TJ Rising 155 °C
Over-Temperature Shutdown
10 °C
Hysteresis
Notes:
6. Measured in test mode.
7. The maximum duty-cycle is limited by the fixed mandatory off-time tOFF of typically 360ns.

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Typical Characteristics

VIN Operating Supply Current VIN Shutdown Current VDD Operating Supply Current
vs. Input Voltage vs. Input Voltage vs. Input Voltage
20 20 10

SHUTDOWN CURRENT (µA)

SUPPLY CURRENT (mA)

SUPPLY CURRENT (mA)

16 16 8

12 12 6

8 VOUT = 3.3V 8 4
IOUT = 0A VOUT = 3.3V
VDD = 5V VDD = 5V 2 VDD= 5V
4 4
SWITCHING VEN = 0V SWITCHING

0 0 0
5 10 15 20 25 30 35 40 5 10 15 20 25 30 35 40 5 10 15 20 25 30 35 40

INPUT VOLTAGE (V) INPUT VOLTAGE (V) INPUT VOLTAGE (V)

Feedback Voltage Total Regulation Current Limit

vs. Input Voltage vs. Input Voltage vs. Input Voltage
0.808 1.0% 20
VOUT = 3.3V
FEEDBACK VOLTAGE (V)

TOTAL REGULATION (%)

0.8% VDD = 5V

CURRENT LIMIT (A)

0.804 IOUT = 0A to 7A 15

0.6%
0.800 10
0.4%
VOUT = 3.3V
0.796 VDD = 5V 5
0.2% VOUT = 3.3V
IOUT = 0A
VDD = 5V
0.792 0.0% 0
5 10 15 20 25 30 35 40 5 10 15 20 25 30 35 40 5 10 15 20 25 30 35 40
INPUT VOLTAGE (V) INPUT VOLTAGE (V) INPUT VOLTAGE (V)

Switching Frequency VDD Operating Supply Current VDD Shutdown Current

vs. Input Voltage vs. Temperature vs. Temperature
400 10 1
VOUT = 3.3V
SWITCHING FREQUENCY (kHz)

SUPPLY CURRENT (mA)

SUPPLY CURRENT (mA)

VDD = 5V 8 0.8
350
IOUT = 0A
6 0.6
300
VIN = 28V
4 0.4 VIN = 28V
VOUT = 3.3V
VDD = 5V IOUT = 0A
250
2 0.2 VDD = 5V
IOUT = 0A
SWITCHING VEN = 0V
200 0 0
5 10 15 20 25 30 35 40 -50 -20 10 40 70 100 130 -50 -20 10 40 70 100 130
INPUT VOLTAGE (V) TEMPERATURE (°C) TEMPERATURE (°C)

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Typical Characteristics (Continued)

VDD UVLO Threshold VIN Operating Supply Current VIN Shutdown Current
vs. Temperature vs. Temperature vs. Temperature
2.8 20 20

SUPPLY CURRENT (mA)

SUPPLY CURRENT (µA)

2.7 Rising 16 16
VDD THRESHOLD (V)

2.6 12 12
Falling

VIN = 28V
2.5 8 8
VOUT = 3.3V
VIN = 28V
VDD = 5V
2.4 4 4 VDD = 5V
IOUT = 0A
IOUT = 0A
SWITCHING
2.3 0 0
-50 -20 10 40 70 100 130 -50 -20 10 40 70 100 130 -50 -20 10 40 70 100 130
TEMPERATURE (°C) TEMPERATURE (°C) TEMPERATURE (°C)

Current Limit Feedback Voltage Load Regulation

vs. Temperature vs. Temperature vs. Temperature
25 0.808 1.0%
VIN = 28V VIN = 28V
FEEBACK VOLTAGE (V)

LOAD REGULATION (%)

20 VOUT = 3.3V VOUT = 3.3V
0.8%
CURRENT LIMIT (A)

0.804 VDD = 5V VDD = 5V

IOUT = 0A IOUT = 0A to 7A
15 0.6%
0.800
10 0.4%
VIN = 28V
0.796
5 VOUT = 3.3V 0.2%
VDD = 5V
0 0.792 0.0%
-50 -20 10 40 70 100 130 -50 -20 10 40 70 100 130 -50 -20 10 40 70 100 130
TEMPERATURE (°C) TEMPERATURE (°C) TEMPERATURE (°C)

Line Regulation Switching Frequency EN Bias Current

vs. Temperature vs. Temperature vs. Temperature
0.5% 400 100
VIN = 28V
SWITCHING FREQUENCY (kHz)

EN BIAS CURRENT (µA)

VIN = 5.5V to 36V

VOUT = 3.3V
LINE REGULATION (%)

0.4% VOUT = 3.3V

80
350 VDD = 5V
VDD = 5V
IOUT = 0A
0.3% 60
300
0.2% 40 VIN = 28V

250 VOUT = 3.3V

0.1% 20 VDD = 5V
VEN = 0V
0.0% 200 0
-50 -20 10 40 70 100 130 -50 -20 10 40 70 100 130 -50 -20 10 40 70 100 130

TEMPERATURE (°C) TEMPERATURE (°C) TEMPERATURE (°C)

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Typical Characteristics (Continued)

Efficiency Feedback Voltage Line Regulation
vs. Output Current vs. Output Current vs. Output Current
100 0.808 0.5%
95 12VIN
VIN = 6V to 36V

FEEDBACK VOLTAGE (V)

LINE REGULATION (%)

90 0.4% VOUT = 3.3V
28VIN 0.804 VDD = 5V
85
EFFICIENCY (%)

80 VOUT = 3.3V 0.3%

36VIN
75 VDD = 5V 0.800
70 0.2%
65 VIN = 28V
0.796
60 VOUT = 3.3V 0.1%
55 VDD = 5V
50 0.792 0.0%
0 1 2 3 4 5 6 7 0 1 2 3 4 5 6 7 0 1 2 3 4 5 6 7
OUTPUT CURRENT (A) OUTPUT CURRENT (A) OUTPUT CURRENT (A)

Die Temperature* (VIN = 12V) Die Temperature* (VIN = 28V) Die Temperature* (VIN = 36V)
vs. Output Current vs. Output Current vs. Output Current
80 80 80

DIE TEMPERATURE (°C)

DIE TEMPERATURE (°C)
DIE TEMPERATURE (°C)

60 60 60

40 40 40

VIN = 12V VIN = 28V VIN = 36V

20 20 20
VOUT = 3.3V VOUT = 3.3V VOUT = 3.3V
VDD= 5V VDD= 5V VDD = 5V
0 0 0
0 1 2 3 4 5 6 7 0 1 2 3 4 5 6 7 0 1 2 3 4 5 6 7

OUTPUT CURRENT (A) OUTPUT CURRENT (A) OUTPUT CURRENT (A)

Efficiency (VIN = 12V) Efficiency (VIN = 28V) Efficiency (VIN = 36V)

vs. Output Current vs. Output Current vs. Output Current
100 95 95
5.0V
90 3.3V
90 3.3V
95 5.0V
2.5V 2.5V
3.3V 85
85 1.8V
2.5V
EFFICIENCY (%)
EFFICIENCY (%)

EFFICIENCY (%)

90 1.8V 80 1.5V
1.5V 1.2V
1.8V 80 1.2V 75 1.0V
1.5V
85 1.0V 0.9V
1.2V 70
75 0.9V 0.8V
1.0V
0.8V
80 0.9V 65
0.8V 70
60
75 65
55
70 60 50
0 1 2 3 4 5 6 7 8 9 0 1 2 3 4 5 6 7 8 9 0 1 2 3 4 5 6 7 8 9
OUTPUT CURRENT (A) OUTPUT CURRENT (A) OUTPUT CURRENT (A)

Die Temperature* : The temperature measurement was taken at the hottest point on the MIC27600 case mounted on a 5 square inch PCB, see
Thermal Measurement section. Actual results will depend upon the size of the PCB, ambient temperature and proximity to other heat emitting
components.

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Typical Characteristics (Continued)

Thermal Derating* Thermal Derating*
vs. Ambient Temperature vs. Ambient Temperature
8 8
0.8V 0.8V
7 7
1.2V 1.2V
OUTPUT CURRENT (A)

OUTPUT CURRENT (A)

6 6
2.5V
5 5 2.5V
3.3V
3.3V
4 4
5V 5V
3 3

2 2
VIN = 12V VIN = 24V
1 1
VOUT = 0.8, 1.2, 2.5, 3.3, 5V VOUT = 0.8, 1.2, 2.5, 3.3, 5V
0 0
85 95 105 115 125 75 85 95 105 115 125

AMBIENT TEMPERATURE (°C) AMBIENT TEMPERATURE (°C)

Die Temperature* : The temperature measurement was taken at the hottest point on the MIC27600 case mounted on a 5 square inch PCB, see
Thermal Measurement section. Actual results will depend upon the size of the PCB, ambient temperature and proximity to other heat emitting
components.

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Functional Characteristics

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Functional Characteristics (Continued)

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Functional Characteristics (Continued)

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Functional Diagram

Figure 1. MIC27600 Block Diagram

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Functional Description where tS = 1/300kHz = 3.33μs. It is not recommended to

use MIC27600 with a OFF-time close to tOFF(min) during
The MIC27600 is an adaptive ON-time synchronous steady-state operation. Also, as VOUT increases, the
step-down DC-DC regulator. It is designed to operate internal ripple injection will increase and reduce the line
over a wide input voltage range from, 4.5V to 36V, and regulation performance. Therefore, the maximum output
provides a regulated output voltage at up to 7A of output voltage of the MIC27600 should be limited to 5.5V.
current. A digitally modified adaptive ON-time control Please refer to “Setting Output Voltage” subsection in
scheme is employed in to obtain a constant switching Application Information for more details.
frequency and to simplify the control compensation.
The actual ON-time and resulting switching frequency
Over-current protection is implemented without the use
will vary with the part-to-part variation in the rise and fall
of an external sense resistor. The device includes an
times of the internal MOSFETs, the output load current,
internal soft-start function which reduces the power
and variations in the VDD voltage. Also, the minimum tON
supply input surge current at start-up by controlling the
results in a lower switching frequency in high VIN to VOUT
output voltage rise time.
applications, such as 26V to 1.0V. The minimum tON
Theory of Operation measured on the MIC27600 evaluation board is about
184ns. During load transients, the switching frequency is
Figure 1 illustrates the block diagram for the control loop
changed due to the varying OFF-time.
of the MIC27600. The output voltage is sensed by the
MIC27600 feedback pin FB via the voltage divider R1 To illustrate, the control loop operation will be analyzed
and R2, and compared to a 0.8V reference voltage VREF in both steady-state and load transient scenarios. For
at the error comparator through a low gain easy analysis, the gain of the gm amplifier is assumed to
transconductance (gm) amplifier. If the feedback voltage be 1. With this assumption, the inverting input of the
decreases and the output of the gm amplifier is below error comparator is the same as the feedback voltage.
0.8V, then the error comparator will trigger the control Figure 2 shows the MIC27600 control loop timing during
logic and generate an ON-time period. The ON-time steady-state operation. During steady-state, the gm
period length is predetermined by the “FIXED tON amplifier senses the feedback voltage ripple, which is
ESTIMATION” circuitry: proportional to the output voltage ripple and the inductor
current ripple, to trigger the ON-time period. The ON-
time is predetermined by the tON estimator. The
VOUT termination of the OFF-time is controlled by the feedback
t ON(estimated) = Eq. 1
VIN × 300kHz voltage. At the valley of the feedback voltage ripple,
which occurs when VFB falls below VREF, the OFF period
ends and the next ON-time period is triggered through
where VOUT is the output voltage and VIN is the power the control logic circuitry.
stage input voltage.
At the end of the ON-time period, the internal high-side
driver turns off the high-side MOSFET and the low-side
driver turns on the low-side MOSFET. The OFF-time
period length depends upon the feedback voltage in
most cases. When the feedback voltage decreases and
the output of the gm amplifier is below 0.8V, the ON-time
period is triggered and the OFF-time period ends. If the
OFF-time period determined by the feedback voltage is
less than the minimum OFF-time tOFF(min), which is about
360ns, then the MIC27600 control logic will apply the
tOFF(min) instead. tOFF(min) is required to maintain enough
energy in the boost capacitor (CBST) to drive the high-
side MOSFET. The maximum duty cycle is obtained
from the 360ns tOFF(min):

t S − t OFF(min) 360ns
D max = = 1− Eq. 2
tS tS Figure 2. MIC27600 Control Loop Timing

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Figure 3 shows the operation of the MIC27600 during a Soft-Start

load transient. The output voltage drops due to the
Soft-start reduces the power supply input surge current
sudden load increase, which causes the VFB to be less
at startup by controlling the output voltage rise time. The
than VREF. This will cause the error comparator to trigger
input surge appears while the output capacitor is
an ON-time period. At the end of the ON-time period, a
charged up. A slower output rise time will draw a lower
minimum OFF-time tOFF(min) is generated to charge CBST
input surge current.
since the feedback voltage is still below VREF. Then, the
next ON-time period is triggered due to the low feedback The MIC27600 implements an internal digital soft-start
voltage. Therefore, the switching frequency changes by making the 0.8V reference voltage VREF ramp from 0
during the load transient, but returns to the nominal fixed to 100% in about 6ms with 9.7mV steps. Therefore, the
frequency once the output has stabilized at the new load output voltage is controlled to increase slowly by a stair-
current level. With the varying duty cycle and switching case VFB ramp. Once the soft-start cycle ends, the
frequency, the output recovery time is fast and the related circuitry is disabled to reduce current
output voltage deviation is small in MIC27600 converter. consumption. VDD must be powered up at the same time
or after VIN to make the soft-start function correctly.

Current Limit
The MIC27600 uses the RDS(ON) of the internal low-side
power MOSFET to sense over-current conditions. This
method will avoid adding cost, board space and power
losses taken by a discrete current sense resistor. The
low-side MOSFET is used because it displays much
lower parasitic oscillations during switching than the
high-side MOSFET.
In each switching cycle of the MIC27600 converter, the
inductor current is sensed by monitoring the low-side
MOSFET in the OFF period. If the peak inductor current
is greater than 15A, then the MIC27600 turns off the
high-side MOSFET and a soft-start sequence is
triggered. This mode of operation is called “hiccup
Figure 3. MIC27600 Load Transient Response mode” and its purpose is to protect the downstream load
in case of a hard short. The current-limit threshold has a
foldback characteristic related to the feedback voltage,
Unlike true current-mode control, the MIC27600 uses the
as shown in Figure 4.
output voltage ripple to trigger an ON-time period. The
output voltage ripple is proportional to the inductor
current ripple if the ESR of the output capacitor is large
enough. The MIC27600 control loop has the advantage Peak Inductor Current
of eliminating the need for slope compensation. vs. Feedback Voltage
20.0
In order to meet the stability requirements, the
PEAK INDUCTOR CURENT (A)

MIC27600 feedback voltage ripple should be in phase

16.0
with the inductor current ripple and large enough to be
sensed by the gm amplifier and the error comparator.
12.0
The recommended feedback voltage ripple is
20mV~100mV. If a low-ESR output capacitor is selected,
then the feedback voltage ripple may be too small to be 8.0 VIN = 12V

sensed by the gm amplifier and the error comparator. VOUT = 0V

Also, the output voltage ripple and the feedback voltage 4.0
ripple are not necessarily in phase with the inductor
current ripple if the ESR of the output capacitor is very 0.0
low. In these cases, ripple injection is required to ensure 0.0 0.2 0.4 0.6 0.8 1.0
proper operation. Please refer to “Ripple Injection” FEEDBACK VOLTAGE (V)
subsection in Application Information for more details
about the ripple injection technique. Figure 4. MIC27600 Current Limit Foldback Characteristic

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the power stroke (high-side switching) cycle, i.e. ΔBST =

Internal MOSFET Gate Drive
10mA x 3.33μs/0.1μF = 333mV. When the low-side
Figure 1 (Block Diagram) shows a bootstrap circuit, MOSFET is turned back on, CBST is recharged through
consisting of D1 (a Schottky diode is recommended) and D1. A small resistor RG, which is in series with CBST, can
CBST. This circuit supplies energy to the high-side drive be used to slow down the turn-on time of the high-side
circuit. Capacitor CBST is charged, while the low-side N-channel MOSFET.
MOSFET is on, and the voltage on the SW pin is
The drive voltage is derived from the VDD supply voltage.
approximately 0V. When the high-side MOSFET driver is
The nominal low-side gate drive voltage is VDD and the
turned on, energy from CBST is used to turn the MOSFET
nominal high-side gate drive voltage is approximately
on. As the high-side MOSFET turns on, the voltage on
VDD – VDIODE, where VDIODE is the voltage drop across
the SW pin increases to approximately VIN. Diode D1 is
D1. An approximate 30ns delay between the high-side
reverse biased and CBST floats high while continuing to
and low-side driver transitions is used to prevent current
keep the high-side MOSFET on. The bias current of the
from simultaneously flowing unimpeded through both
high-side driver is less than 10mA so a 0.1μF to 1μF is
MOSFETs.
sufficient to hold the gate voltage with minimal droop for

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Application Information but the increase in core loss will reduce the efficiency of
the power supply. This is especially noticeable at low
Inductor Selection output power. The winding resistance decreases
efficiency at the higher output current levels. The
Values for inductance, peak, and RMS currents are
winding resistance must be minimized although this
required to select the output inductor. The input and
usually comes at the expense of a larger inductor. The
output voltages and the inductance value determine the
power dissipated in the inductor is equal to the sum of
peak-to-peak inductor ripple current. Generally, higher
the core and copper losses. At higher output loads, the
inductance values are used with higher input voltages.
core losses are usually insignificant and can be ignored.
Larger peak-to-peak ripple currents will increase the
At lower output currents, the core losses can be a
power dissipation in the inductor and MOSFETs. Larger
significant contributor. Core loss information is usually
output ripple currents will also require more output
available from the magnetics vendor. Copper loss in the
capacitance to smooth out the larger ripple current.
inductor is calculated by Equation 7:
Smaller peak-to-peak ripple currents require a larger
inductance value and therefore a larger and more
2
expensive inductor. A good compromise between size, PINDUCTOR(Cu) = IL(RMS) × RWINDING Eq. 7
loss and cost is to set the inductor ripple current to be
equal to 20% of the maximum output current. The
The resistance of the copper wire, RWINDING, increases
inductance value is calculated by Equation 3:
with the temperature. The value of the winding
resistance used should be at the operating temperature:
VOUT × (VIN(max) − VOUT )
L= Eq. 3
VIN(max) × fsw × 20% × IOUT(max) PWINDING(Ht) = RWINDING(20°C) × (1 + 0.0042 × (TH – T20°C))
Eq. 8
where:
fSW = switching frequency, 300kHz where:
20% = ratio of AC ripple current to DC output current TH = temperature of wire under full load
VIN(max) = maximum power stage input voltage T20°C = ambient temperature
The peak-to-peak inductor current ripple is: RWINDING(20°C) = room temperature winding resistance
(usually specified by the manufacturer)

VOUT × (VIN(max) − VOUT ) Output Capacitor Selection

ΔIL(pp) = Eq. 4
VIN(max) × fsw × L The type of the output capacitor is usually determined by
its equivalent series resistance (ESR). Voltage and RMS
current capability are two other important factors for
The peak inductor current is equal to the average output selecting the output capacitor. Recommended capacitor
current plus one half of the peak-to-peak inductor current types are ceramic, low-ESR aluminum electrolytic, OS-
ripple. CON and POSCAP. The output capacitor’s ESR is
usually the main cause of the output ripple. The output
capacitor ESR also affects the control loop from a
IL(pk) =IOUT(max) + 0.5 × ΔIL(pp) Eq. 5
stability point of view. The maximum value of ESR is
calculated:
The RMS inductor current is used to calculate the I2R
losses in the inductor.
ΔVOUT(pp)
ESR COUT ≤ Eq. 9
ΔIL(PP)
2
2 ΔIL(PP)
IL(RMS) = IOUT(max) + Eq. 6
12 where:
ΔVOUT(pp) = peak-to-peak output voltage ripple
Maximizing efficiency requires the proper selection of ΔIL(PP) = peak-to-peak inductor current ripple
core material and minimizing the winding resistance. The
high frequency operation of the MIC27600 requires the
use of ferrite materials for all but the most cost sensitive
applications. Lower cost iron powder cores may be used

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Micrel, Inc. MIC27600

The total output ripple is a combination of the ESR and peak inductor current, so:
output capacitance. The total ripple is calculated in
Equation 10:
ΔVIN = IL(pk) × CESR Eq. 13

2
⎛ ⎞ The input capacitor must be rated for the input current
ΔVOUT(pp) = ⎜⎜
C
ΔIL(PP)
× f × 8 ⎟ (
⎟ + ΔIL(PP) × ESR C )
OUT
2
ripple. The RMS value of input capacitor current is
⎝ OUT SW ⎠ determined at the maximum output current. Assuming
Eq. 10 the peak-to-peak inductor current ripple is low:

where: ICIN(RMS) ≈ IOUT(max) × D × (1 − D) Eq. 14

D = duty cycle
COUT = output capacitance value
The power dissipated in the input capacitor is:
fSW = switching frequency

PDISS(CIN) = ICIN(RMS)2 × CESR Eq. 15

As described in the “Theory of Operation” subsection in
Functional Description, the MIC27600 requires at least
20mV peak-to-peak ripple at the FB pin to make the gm Ripple Injection
amplifier and the error comparator behave properly. Also,
the output voltage ripple should be in phase with the The VFB ripple required for proper operation of the
inductor current. Therefore, the output voltage ripple MIC27600 gm amplifier and error comparator is 20mV to
caused by the output capacitors value should be much 100mV. However, the output voltage ripple is generally
smaller than the ripple caused by the output capacitor designed as 1% to 2% of the output voltage. For a low
ESR. If low-ESR capacitors, such as ceramic capacitors, output voltage, such as a 1V, the output voltage ripple is
are selected as the output capacitors, a ripple injection only 10mV to 20mV, and the feedback voltage ripple is
method should be applied to provide the enough less than 20mV. If the feedback voltage ripple is so small
feedback voltage ripple. Please refer to the “Ripple that the gm amplifier and error comparator can’t sense it,
Injection” subsection for more details. then the MIC27600 will lose control and the output
voltage is not regulated. In order to have some amount
The voltage rating of the capacitor should be 20% of VFB ripple, a ripple injection method is applied for low
greater for aluminum electrolytic or OS-CON. The output output voltage ripple applications.
capacitor RMS current is calculated in Equation 11:
The applications are divided into three situations
according to the amount of the feedback voltage ripple:
ΔIL(PP) 1) Enough ripple at the feedback voltage due to the large
ICOUT (RMS) = Eq. 11
12 ESR of the output capacitors.
As shown in Figure 5a, the converter is stable without
any ripple injection. The feedback voltage ripple is:
The power dissipated in the output capacitor is:

R2
2
PDISS(COUT ) = ICOUT (RMS) × ESR COUT Eq. 12 ΔVFB(pp) = × ESR COUT × ΔIL (pp) Eq. 16
R1 + R2
Input Capacitor Selection
where ΔIL(pp) is the peak-to-peak value of the inductor
The input capacitor for the power stage input VIN should
current ripple.
be selected for ripple current rating and voltage rating.
Tantalum input capacitors may fail when subjected to 2) Inadequate ripple at the feedback voltage due to the
high inrush currents, caused by turning the input supply small ESR of the output capacitors.
on. A tantalum input capacitor’s voltage rating should be The output voltage ripple is fed into the FB pin through a
at least two times the maximum input voltage to feedforward capacitor Cff in this situation, as shown in
maximize reliability. Aluminum electrolytic, OS-CON, and Figure 5b. The typical Cff value is between 1nF and
multilayer polymer film capacitors can handle the higher 100nF.
inrush currents without voltage de-rating. The input
voltage ripple will primarily depend on the input
capacitor’s ESR. The peak input current is equal to the

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Micrel, Inc. MIC27600

With the feedforward capacitor, the feedback voltage The injected ripple is:
ripple is very close to the output voltage ripple:
1
ΔVFB(pp) = VIN × K div × D × (1 - D) × Eq. 18
ΔVFB(pp) ≈ ESR × ΔIL (pp) Eq. 17 fSW × τ

3) Virtually no ripple at the FB pin voltage due to the very

low ESR of the output capacitors. R1//R2
K div = Eq. 19
R inj + R1//R2

where
VIN = Power stage input voltage
D = duty cycle
fSW = switching frequency
τ = (R1//R2//Rinj) × Cff

In Equations 18 and 19, it is assumed that the time

constant associated with Cff must be much greater than
Figure 5a. Enough Ripple at FB
the switching period:

1 T
= << 1 Eq. 20
fSW × τ τ

If the voltage divider resistors R1 and R2 are in the kΩ

range, a Cff of 1nF to 100nF can easily satisfy the large
time constant requirements. Also, a 100nF injection
capacitor Cinj is used in order to be considered as short
for a wide range of the frequencies.
The process of sizing the ripple injection resistor and
capacitors is:
Figure 5b. Inadequate Ripple at FB
Step 1. Select Cff to feed all output ripples into the
feedback pin and make sure the large time constant
assumption is satisfied. Typical choice of Cff is 1nF to
100nF if R1 and R2 are in kΩ range.
Step 2. Select Rinj according to the expected feedback
voltage ripple using Equation 21:

ΔVFB(pp) fSW × τ
K div = × Eq. 21
VIN D × (1 − D)

Then the value of Rinj is obtained as:

Figure 5c. Invisible Ripple at FB

1
In this situation, the output voltage ripple is less than R inj = (R1//R2) × ( − 1) Eq. 22
K div
20mV. Therefore, additional ripple is injected into the FB
pin from the switching node SW via a resistor Rinj and a
capacitor Cinj, as shown in Figure 5c. Step 3. Select Cinj as 100nF, which could be considered
as short for a wide range of the frequencies.

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Micrel, Inc. MIC27600

Setting Output Voltage

The MIC27600 requires two resistors to set the output
voltage as shown in Figure 6.

Figure 7. Internal Ripple Injection

Figure 6. Voltage-Divider Configuration Thermal Measurements

Measuring the IC’s case temperature is recommended to
The output voltage is determined by Equation 23: ensure it is within its operating limits. Although this might
seem like a very elementary task, it is easy to get
erroneous results. The most common mistake is to use
R1 the standard thermal couple that comes with a thermal
VOUT = VFB × (1 + ) Eq. 23
R2 meter. This thermal couple wire gauge is large, typically
22 gauge, and behaves like a heatsink, resulting in a
lower case measurement.
where, VFB = 0.8V. A typical value of R1 can be between
3kΩ and 10kΩ. If R1 is too large, it may allow noise to be Two methods of temperature measurement are using a
introduced into the voltage feedback loop. If R1 is too smaller thermal couple wire or an infrared thermometer.
small, it will decrease the efficiency of the power supply, If a thermal couple wire is used, it must be constructed
especially at light loads. Once R1 is selected, R2 can be of 36 gauge wire or higher then (smaller wire size) to
calculated using: minimize the wire heat-sinking effect. In addition, the
thermal couple tip must be covered in either thermal
grease or thermal glue to make sure that the thermal
VFB × R1 couple junction is making good contact with the case of
R2 = Eq. 24
VOUT − VFB the IC. Omega brand thermal couple (5SC-TT-K-36-36)
is adequate for most applications.
Wherever possible, an infrared thermometer is
In addition to the external ripple injection added at the
recommended. The measurement spot size of most
FB pin, internal ripple injection is added at the inverting
infrared thermometers is too large for an accurate
input of the comparator inside the MIC27600, as shown
reading on a small form factor ICs. However, a IR
in Figure 7. The inverting input voltage VINJ is clamped to
thermometer from Optris has a 1mm spot size, which
1.2V. As VOUT increases, the swing of VINJ will be
makes it a good choice for measuring the hottest point
clamped. The clamped VINJ reduces the line regulation
on the case. An optional stand makes it easy to hold the
because it is reflected back as a DC error on the FB
beam on the IC for long periods of time.
terminal. To avoid this line regulation problem, the
maximum output voltage of the MIC27600 should be
limited to 5.5V for up to 28V VIN and 3.6V for VIN higher
than 28V.

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Micrel, Inc. MIC27600

PCB Layout Guidelines Inductor

Warning!!! To minimize EMI and output noise, follow • Keep the inductor connection to the switch node
these layout recommendations. (SW) short.
PCB Layout is critical to achieve reliable, stable and • Do not route any digital lines underneath or close to
efficient performance. A ground plane is required to the inductor.
control EMI and minimize the inductance in power, • Keep the switch node (SW) away from the feedback
signal and return paths. (FB) pin.
The following guidelines should be followed to insure • The CS pin should be connected directly to the SW
proper operation of the MIC27600 converter. pin to accurate sense the voltage across the low-
side MOSFET.
IC
• To minimize noise, place a ground plane underneath
• The 2.2µF ceramic capacitor, which is connected to the inductor.
the VDD pin, must be located right at the IC. The
• The inductor can be placed on the opposite side of
VDD pin is very noise sensitive and placement of the
the PCB with respect to the IC. It does not matter
capacitor is very critical. Use wide traces to connect
whether the IC or inductor is on the top or bottom as
to the VDD and PGND pins.
long as there is enough air flow to keep the power
• The signal ground pin (SGND) must be connected components within their temperature limits. The
directly to the ground planes. Do not route the input and output capacitors must be placed on the
SGND pin to the PGND Pad on the top layer. same side of the board as the IC.
• Place the IC close to the point of load (POL). Output Capacitor
• Use fat traces to route the input and output power • Use a wide trace to connect the output capacitor
lines. ground terminal to the input capacitor ground
• Signal and power grounds should be kept separate terminal.
and connected at only one location. • Phase margin will change as the output capacitor
value and ESR changes. Contact the factory if the
Input Capacitor output capacitor is different from what is shown in
• Place the input capacitor next. the BOM.
• Place the input capacitors on the same side of the • The feedback trace should be separate from the
board and as close to the IC as possible. power trace and connected as close as possible to
• Keep both the PVIN pin and PGND connections the output capacitor. Sensing a long high current
short. load trace can degrade the DC load regulation.
• Place several vias to the ground plane close to the RC Snubber
input capacitor ground terminal. • Place the RC snubber on either side of the board
• Use either X7R or X5R dielectric input capacitors. and as close to the SW pin as possible.
Do not use Y5V or Z5U type capacitors.
• Do not replace the ceramic input capacitor with any
other type of capacitor. Any type of capacitor can be
placed in parallel with the input capacitor.
• If a Tantalum input capacitor is placed in parallel
with the input capacitor, it must be recommended for
switching regulator applications and the operating
voltage must be derated by 50%.
• In “Hot-Plug” applications, a Tantalum or Electrolytic
bypass capacitor must be used to limit the over-
voltage spike seen on the input supply with power is
suddenly applied.

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Micrel, Inc. MIC27600

Evaluation Board Schematic

Figure 8. Schematic of MIC27600 Evaluation Board

(J13, R13, R15 are for testing purposes)

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Micrel, Inc. MIC27600

Bill of Materials
Item Part Number Manufacturer Description Qty.
(1)
C1 B41125A6107M EPCOS 100µF Aluminum Capacitor, SMD, 50V 1
(2)
12105C475KAZ2A AVX
C2, C3 4.7µF Ceramic Capacitor, X7R, Size 1210, 50V 2
GRM32ER71H475KA88L Murata(3)
12106D107MAT2A AVX
C13 100µF Ceramic Capacitor, X5R, Size 1210, 6.3V 1
GRM32ER60J107ME20L Murata
06035C104KAT2A AVX
C6, C7, C10 GRM188R71H104KA93D Murata 0.1µF Ceramic Capacitor, X7R, Size 0603, 50V 3
(4)
C1608X7R1H104K TDK
0805ZC225MAT2A AVX
C8, C9 GRM21BR71A225KA01L Murata 2.2µF Ceramic Capacitor, X7R, Size 0805, 10V 2
C2012X7R1A225K TDK
06035C102KAT2A AVX
C11 GRM188R71H102KA01D Murata 1nF Ceramic Capacitor, X7R, Size 0603, 50V 1
C1608X7R1H102K TDK
06035C223KAZ2A AVX
C12 GRM188R71H223K Murata 22nF Ceramic Capacitor, X7R, Size 0603, 50V 1
C1608X7R1H223K TDK
C4, C5 Open
C14, C15 Open
SD101AWS-7 Diodes Inc(6)
D1 Small Signal Schottky Diode 1
SD101AWS-V Vishay(7)
D2 CMDZ5L6 Central Semi(8) 5.6V Zener Diode 1
L1 HCF1305-4R0-R Cooper Bussmann(9) 4.0µH Inductor, 15A Saturation Current 1
Q1 FCX619 ZETEX 50V NPN Transistor 1
R1 CRCW06034R75FKEA Vishay Dale 4.75Ω Resistor, Size 0603, 1% 1
R2, R16 CRCW08051R21FKEA Vishay Dale 1.21Ω Resistor, Size 0805, 1% 2
R3, R4 CRCW060310K0FKEA Vishay Dale 10kΩ Resistor, Size 0603, 1% 2
R5 CRCW060380K6FKEA Vishay Dale 80.6kΩ Resistor, Size 0603, 1% 1
R6 CRCW060340K2FKEA Vishay Dale 40.2kΩ Resistor, Size 0603, 1% 1
R7 CRCW060320K0FKEA Vishay Dale 20kΩ Resistor, Size 0603, 1% 1
Notes:
1. EPCOS: www.epcos.com.
2. AVX: www.avx.com.
3. Murata: www.murata.com.
4. TDK: www.tdk.com.
5. SANYO: www.sanyo.com.
6. Diode Inc.: www.diodes.com.
7. Vishay: www.vishay.com.
8. Central Semi: www.centralsemi.com.
9. Cooper Bussmann: www.cooperbussmann.com.

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Micrel, Inc. MIC27600

Bill of Materials (Continued)

Item Part Number Manufacturer Description Qty.
R8 CRCW060311K5FKEA Vishay Dale 11.5kΩ Resistor, Size 0603, 1% 1
R9 CRCW06038K06FKEA Vishay Dale 8.06kΩ Resistor, Size 0603, 1% 1
R10 CRCW06034K75FKEA Vishay Dale 4.75kΩ Resistor, Size 0603, 1% 1
R11 CRCW06033K24FKEA Vishay Dale 3.24kΩ Resistor, Size 0603, 1% 1
R12 CRCW06031K91FKEA Vishay Dale 1.91kΩ Resistor, Size 0603, 1% 1
R13 CRCW06030000FKEA Vishay Dale 0Ω Resistor, Size 0603, 5% 1
R14 CRCW06035K23FKEA Vishay Dale 5.23kΩ Resistor, Size 0603, 1% 1
R15 CRCW060349R9FKEA Vishay Dale 49.9Ω Resistor, Size 0603, 1% 1
(10)
U1 MIC27600YJL Micrel. Inc. 26V/7A Synchronous Buck DC-DC Regulator 1
Note:
10. Micrel, Inc.: www.micrel.com.

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Micrel, Inc. MIC27600

PCB Layout

Figure 9. MIC27600 Evaluation Board Top Layer

Figure 10. MIC27600 Evaluation Board Mid-Layer 1 (Ground Plane)

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Micrel, Inc. MIC27600

PCB Layout (Continued)

Figure 11. MIC27600 Evaluation Board Mid-Layer 2

Figure 12. MIC27600 Evaluation Board Bottom Layer

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Micrel, Inc. MIC27600

Recommended Land Pattern

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Micrel, Inc. MIC27600

Package Information

28-Lead 5mm x 6mm MLF® (YJL)

MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA

TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com

Micrel makes no representations or warranties with respect to the accuracy or completeness of the information furnished in this data sheet. This
information is not intended as a warranty and Micrel does not assume responsibility for its use. Micrel reserves the right to change circuitry,
specifications and descriptions at any time without notice. No license, whether express, implied, arising by estoppel or otherwise, to any intellectual
property rights is granted by this document. Except as provided in Micrel’s terms and conditions of sale for such products, Micrel assumes no liability
whatsoever, and Micrel disclaims any express or implied warranty relating to the sale and/or use of Micrel products including liability or warranties
relating to fitness for a particular purpose, merchantability, or infringement of any patent, copyright or other intellectual property right

Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product
can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant
into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A
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indemnify Micrel for any damages resulting from such use or sale.

© 2010 Micrel, Incorporated.

July 2011 28 M9999-070811