LT3436

3A, 800kHz Step-Up

Switching Regulator
U
FEATURES DESCRIPTIO
■ Constant 800kHz Switching Frequency The LT®3436 is an 800kHz monolithic boost switching
■ Wide Operating Voltage Range: 3V to 25V regulator. A high efficiency 3A, 0.1Ω switch is included on
■ High Efficiency 0.1Ω/3A Switch the die together with all the control circuitry required to
■ 1.2V Feedback Reference Voltage complete a high frequency, current-mode switching regu-
■ ±2% Overall Output Voltage Tolerance lator. Current-mode control provides fast transient re-
■ Uses Low Profile Surface Mount External sponse and excellent loop stability.
Components
New design techniques achieve high efficiency at high
■ Low Shutdown Current: 11µA
switching frequencies over a wide operating range. A low
■ Synchronizable from 1MHz to 1.4MHz
■
dropout internal regulator maintains consistent perfor-
Current-Mode Control
mance over a wide range of inputs from 24V systems to Li-
■ Constant Maximum Switch Current Rating
Ion batteries. An operating supply current of 1mA main-
at All Duty Cycles*
■
tains high efficiency, especially at lower output currents.
Available in a Small Thermally Enhanced
Shutdown reduces quiescent current to 11µA. Maximum
TSSOP-16 Package
switch current remains constant at all duty cycles. Syn-
U chronization capability allows an external logic level signal
APPLICATIO S to increase the internal oscillator from 1MHz to 1.4MHz.
■ DSL Modems Full cycle-by-cycle switch current limit protection and ther-
■ Portable Computers mal shutdown are provided. High frequency operation al-
■ Battery-Powered Systems lows the reduction of input and output filtering components
■ Distributed Power and permits the use of tiny chip inductors. The LT3436 is
available in an exposed pad, 16-pin TSSOP package.
, LTC and LT are registered trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
*Protectd by U.S. Patents including 6535042, 6611131, 6498466

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TYPICAL APPLICATIO
Efficiency vs Load Current
5V to 12V Boost Converter
90
3.9µH VIN = 5V
VOUT = 12V
85
B220A
OUTPUT
INPUT 12V
VIN VSW 80
5V
EFFICIENCY (%)

OPEN 0.9A†
4.7µF LT3436 90.9k
CERAMIC OR
HIGH 75
SHDN FB
= ON SYNC GND VC
70
10k 22µF
10nF CERAMIC
1%
470pF 65
4.7k

†MAXIMUM OUTPUT CURRENT IS SUBJECT TO THERMAL DERATING.

60
3436 TA01
0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8
LOAD CURRENT (A)
3436 TA01b

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ABSOLUTE MAXIMUM RATINGS PACKAGE/ORDER INFORMATION
(Note 1)
Input Voltage .......................................................... 25V TOP VIEW ORDER PART NUMBER
GND 1 16 GND
Switch Voltage ......................................................... 35V
SHDN Pin ............................................................... 25V
VIN
SW
2
3
15 NC
LT3436EFE
14 SYNC

FB Pin Current ....................................................... 1mA SW 4

17
13 VC
GND 5 12 FB
SYNC Pin Current .................................................. 1mA GND 6 11 SHDN FE PART MARKING
Operating Junction Temperature Range (Note 2) NC 7 10 NC

LT3436E .......................................... – 40°C to 125°C GND 8 9 GND

3436EFE
Storage Temperature Range ................ – 65°C to 150°C FE PACKAGE
16-LEAD PLASTIC TSSOP
EXPOSED PAD IS GND (PIN 17),
Lead Temperature (Soldering, 10 sec)................. 300°C MUST BE SOLDERED TO PCB
TJMAX = 125°C, θJA = 45°C/W,
θJC(PAD) = 10°C/W

Consult LTC Marketing for parts specified with wider operating temperature ranges.

ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature
range, otherwise specifications are at TA = 25°C. VIN = 15V, VC = 0.8V, SHDN, SYNC and switch open unless otherwise noted.
PARAMETER CONDITION MIN TYP MAX UNITS
Recommended Operating Voltage ● 3 25 V
Maximum Switch Current Limit ● 3 4 6 A
Oscillator Frequency 3.3V < VIN < 25V ● 640 800 960 kHz
Switch On Voltage Drop ISW = 3A ● 330 550 mV
VIN Undervoltage Lockout (Note 3) ● 2.47 2.6 2.73 V
VIN Supply Current ISW = 0A ● 1 1.3 mA
VIN Supply Current/ISW ISW = 3A 15 mA/A
Shutdown Supply Current VSHDN = 0V, VIN = 25V, VSW = 25V 11 25 µA
● 45 µA
Feedback Voltage 3V < VIN < 25V, 0.4V < VC < 0.9V 1.182 1.2 1.218 V
● 1.176 1.224 V
FB Input Current ● 0 – 0.2 – 0.4 µA
FB to VC Voltage Gain 0.4V < VC < 0.9V 150 350
FB to VC Transconductance ∆IVC = ±10µA ● 500 850 1300 µMho
VC Pin Source Current VFB = 1V ● – 85 – 120 – 165 µA
VC Pin Sink Current VFB = 1.4V ● 70 110 165 µA
VC Pin to Switch Current Transconductance 4.8 A/V
VC Pin Minimum Switching Threshold Duty Cycle = 0% 0.3 V
VC Pin 3A ISW Threshold 0.9 V
Maximum Switch Duty Cycle VC = 1.2V, ISW = 350mA 85 90 %
VC = 1.2V, ISW = 1A ● 80 %
SHDN Threshold Voltage ● 1.28 1.35 1.42 V
SHDN Input Current (Shutting Down) SHDN = 60mV Above Threshold ● –7 –10 –13 µA
SHDN Threshold Current Hysteresis SHDN = 100mV Below Threshold 4 7 10 µA
SYNC Threshold Voltage 1.5 2.2 V
SYNC Input Frequency 1 1.4 MHz
SYNC Pin Resistance ISYNC = 1mA 20 kΩ
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 LT3436
ELECTRICAL CHARACTERISTICS
Note 1: Absolute Maximum Ratings are those values beyond which the life Note 3: Minimum input voltage is defined as the voltage where the internal
of a device may be impaired. regulator enters lockout. Actual minimum input voltage to maintain a
Note 2: The LT3436E is guaranteed to meet performance specifications regulated output will depend on output voltage and load current. See
from 0°C to 125°C. Specifications over the – 40°C to 125°C operating Applications Information.
junction temperature range are assured by design, characterization and
correlation with statistical process controls.

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TYPICAL PERFORMANCE CHARACTERISTICS
FB Voltage Switch On Voltage Drop Oscillator Frequency
1.220 500 920
450 TA = 125°C
1.215 890

OSCILLATOR FREQUENCY (kHz)

400
1.210 TA = 25°C 860
SWITCH VOLTAGE (mV)

350
FB VOLTAGE (V)

1.205 300 830

1.200 250 TA = –40°C 800

200
1.195 770
150
1.190 740
100
1.185 50 710

1.180 0 680
–50 –25 0 25 50 75 100 125 0 0.5 1.0 1.5 2.0 2.5 3.0 –50 –25 0 25 50 75 100 125
TEMPERATURE (°C) SWITCH CURRENT (A) TEMPERATURE (°C)
3436 G01 3436 G02 3436 G03

SHDN Threshold SHDN Supply Current SHDN Input Current

1.40 14 –12
TA = 25°C
12 SHDN = 0V
–10
1.38
SHDN INPUT CURRENT (µA)

SHUTTING DOWN
SHDN THRESHOLD (V)

10
VIN CURRENT (µA)

–8
1.36
8
–6
6
1.34
–4 STARTING UP
4
1.32
2 –2

1.30 0 0
–50 –25 0 25 50 75 100 125 0 5 10 15 20 25 30 –50 –25 0 25 50 75 100 125
TEMPERATURE (°C) INPUT VOLTAGE (V) TEMPERATURE (°C)
3436 G04 3436 G05 3436 G06

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TYPICAL PERFOR A CE CHARACTERISTICS
SHDN Supply Current Input Supply Current Current Limit Foldback
300 1200 4.0 40
TA = 25°C TA = 25°C TA = 25°C
VIN = 15V
250 1000 3.5
SWITCH CURRENT

SWITCH PEAK CURRENT (A)

FB INPUT CURRENT (µA)

3.0 30

VIN CURRENT (µA)

VIN CURRENT (µA)

200 800
2.5
MINIMUM
150 600 INPUT
2.0 20
VOLTAGE
1.5
100 400
1.0 10
50 200
0.5

0 0 0 0
0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 0 5 10 15 20 25 30 0 0.2 0.4 0.6 0.8 1.0 1.2
SHDN VOLTAGE (V) INPUT VOLTAGE (V) FEEDBACK VOLTAGE (V)
3436 G07 3436 G08 3436 G09

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PIN FUNCTIONS
GND (Pins 1, 5, 6, 8, 9, 16, 17): Short GND pins 1, 5, 6,8, a predetermined level. Float or pull high to put the regula-
9, 16 and the exposed pad (pin 17) on the PCB. The GND tor in the operating mode.
is the reference for the regulated output, so load regulation FB (Pin 12): The feedback pin is used to set output voltage
will suffer if the “ground” end of the load is not at the same using an external voltage divider that generates 1.2V at the
voltage as the GND of the IC. This condition occurs when pin with the desired output voltage. If required, the current
the load current flows through the metal path between the limit can be reduced during start up when the FB pin is
GND pins and the load ground point. Keep the ground path below 0.5V (see the Current Limit Foldback graph in the
short between the GND pins and the load and use a ground Typical Performance Characteristics section). An imped-
plane when possible. Keep the path between the input ance of less than 5kΩ at the FB pin is needed for this
bypass and the GND pins short. The exposed pad should feature to operate.
be attached to a large copper area to improve thermal
performance. VC (Pin 13): The VC pin is the output of the error amplifier
and the input of the peak switch current comparator. It is
VIN (Pin 2): This pin powers the internal circuitry and normally used for frequency compensation, but can do
internal regulator. Keep the external bypass capacitor double duty as a current clamp or control loop override.
close to this pin. This pin sits at about 0.3V for very light loads and 0.9V at
SW (Pins 3, 4): The switch pin is the collector of the on- maximum load.
chip power NPN switch and has large currents flowing SYNC (Pin 14): The sync pin is used to synchronize the
through it. Keep the traces to the switching components as
internal oscillator to an external signal. It is directly logic
short as possible to minimize radiation and voltage spikes. compatible and can be driven with any signal between
SHDN (Pin 11): The shutdown pin is used to turn off the 20% and 80% duty cycle. The synchronizing range is
regulator and to reduce input drain current to a few equal to initial operating frequency, up to 1.4MHz. See
microamperes. The 1.35V threshold can function as an Synchronization section in Applications Information for
accurate undervoltage lockout (UVLO), preventing the details. When not in use, this pin should be grounded.
regulator from operating until the input voltage has reached
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BLOCK DIAGRAM
The LT3436 is a constant frequency, current-mode boost amplifier commands current to be delivered to the output
converter. This means that there is an internal clock and rather than voltage. A voltage fed system will have low
two feedback loops that control the duty cycle of the power phase shift up to the resonant frequency of the inductor
switch. In addition to the normal error amplifier, there is a and output capacitor, then an abrupt 180° shift will occur.
current sense amplifier that monitors switch current on a The current fed system will have 90° phase shift at a much
cycle-by-cycle basis. A switch cycle starts with an oscilla- lower frequency, but will not have the additional 90° shift
tor pulse which sets the RS flip-flop to turn the switch on. until well beyond the LC resonant frequency. This makes
When switch current reaches a level set by the inverting it much easier to frequency compensate the feedback loop
input of the comparator, the flip-flop is reset and the and also gives much quicker transient response.
switch turns off. Output voltage control is obtained by A comparator connected to the shutdown pin disables the
using the output of the error amplifier to set the switch internal regulator, reducing supply current.
current trip point. This technique means that the error

INPUT

2.5V BIAS INTERNAL

REGULATOR VCC

SLOPE COMP Σ

0.3V

800kHz SW
SYNC S Q1
OSCILLATOR CURRENT RS DRIVER POWER
COMPARATOR FLIP-FLOP CIRCUITRY SWITCH
+ R

CURRENT SENSE
SHUTDOWN – AMPLIFIER VOLTAGE
COMPARATOR GAIN = 40
7µA

+ – +
0.005Ω

1.35V –

SHDN –
FB
3µA

+
ERROR
VC AMPLIFIER
1.2V
gm = 850µMho
GND
3436 F01

Figure 1. Block Diagram

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APPLICATIONS INFORMATION
FB RESISTOR NETWORK to 22µF range. Since the absolute value of capacitance
The suggested resistance (R2) from FB to ground is 10k defines the pole frequency of the output stage, an X7R or
1%. This reduces the contribution of FB input bias current X5R type ceramic, which have good temperature stability,
to output voltage to less than 0.2%. The formula for the is recommended.
resistor (R1) from VOUT to FB is: Tantalum capacitors are usually chosen for their bulk
capacitance properties, useful in high transient load appli-
R2(VOUT − 1. 2) cations. ESR rather than absolute value defines output
R1 =
1.2 − R2(0.2µA) ripple at 800kHz. Values in the 22µF to 100µF range are
generally needed to minimize ESR and meet ripple current
ratings. Care should be taken to ensure the ripple ratings
are not exceeded.
LT3436 VSW
Table 1. Surface Mount Solid Tantalum Capacitor ESR and
OUTPUT
ERROR Ripple Current
AMPLIFIER
+ 1.2V
E Case Size ESR (Max, Ω ) Ripple Current (A)
R1 AVX TPS, Sprague 593D 0.1 to 0.3 0.7 to 1.1
FB
+
– D Case Size
R2 AVX TPS, Sprague 593D 0.1 to 0.3 0.7 to 1.1
10k
C Case Size
3436 F02
AVX TPS 0.2 (typ) 0.5 (typ)
VC GND

Figure 2. Feedback Network

INPUT CAPACITOR
OUTPUT CAPACITOR Unlike the output capacitor, RMS ripple current in the
input capacitor is normally low enough that ripple current
Step-up regulators supply current to the output in pulses. rating is not an issue. The current waveform is triangular,
The rise and fall times of these pulses are very fast. The with an RMS value given by:
output capacitor is required to reduce the voltage ripple
this causes. The RMS ripple current can be calculated 0.29(VIN )(VOUT − VIN )
from: IRIPPLE(RMS) =
(L)( f)(VOUT )
IRIPPLE(RMS) = IOUT (VOUT − VIN ) / VIN At higher switching frequency, the energy storage require-
ment of the input capacitor is reduced so values in the
The LT3436 will operate with both ceramic and tantalum range of 2.2µF to 10µF are suitable for most applications.
output capacitors. Ceramic capacitors are generally cho- Y5V or similar type ceramics can be used since the
sen for their small size, very low ESR (effective series absolute value of capacitance is less important and has no
resistance), and good high frequency operation, reducing significant effect on loop stability. If operation is required
output ripple voltage. Their low ESR removes a useful zero close to the minimum input voltage required by either the
in the loop frequency response, common to tantalum output or the LT3436, a larger value may be necessary.
capacitors. To compensate for this, the VC loop compen- This is to prevent excessive ripple causing dips below the
sation pole frequency must typically be reduced by a factor minimum operating voltage resulting in erratic operation.
of 10. Typical ceramic output capacitors are in the 4.7µF

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APPLICATIONS INFORMATION
INDUCTOR CHOICE AND MAXIMUM OUTPUT The recommended minimum inductance is:
CURRENT
(VIN )2 (VOUT – VIN )
When choosing an inductor, there are 2 conditions that LMIN =
0.4(VOUT )2 (IOUT )(f)
limit the minimum inductance; required output current,
and avoidance of subharmonic oscillation. The maximum The inductor value may need further adjustment for other
output current for the LT3436 in a standard boost con- factors such as output voltage ripple and filtering require-
verter configuration with an infinitely large inductor is: ments. Remember also, inductance can drop significantly
with DC current and manufacturing tolerance.
VIN • η
IOUT (MAX) = 3A The inductor must have a rating greater than its peak
VOUT operating current to prevent saturation resulting in effi-
Where η = converter efficiency (typically 0.87 at high ciency loss. Peak inductor current is given by:
current).
(VOUT )(IOUT ) VIN (VOUT − VIN )
As the value of inductance is reduced, ripple current ILPEAK = +
VIN • η 2VOUT (L)(f)
increases and IOUT(MAX) is reduced. The minimum induc-
tance for a required output current is given by: Also, consideration should be given to the DC resistance
of the inductor. Inductor resistance contributes directly to
VIN (VOUT – VIN ) the efficiency losses in the overall converter.
LMIN =
⎛ (V )(I )⎞ Suitable inductors are available from Coilcraft, Coiltronics,
2VOUT (f)⎜ 3 – OUT OUT ⎟
⎝ VIN • η ⎠ Dale, Sumida, Toko, Murata, Panasonic and other manu-
factures.
The second condition, avoidance of subharmonic oscilla- Table 2
tion, must be met if the operating duty cycle is greater than PART VALUE ISAT(DC) DCR HEIGHT
50%. The slope compensation circuit within the LT3436 NUMBER (µH) (Amps) (Ω) (mm)
prevents subharmonic oscillation for inductor ripple cur- Coilcraft
rents of up to 1.4AP-P, defining the minimum inductor DO1608C-222 2.2 2.4 0.07 2.9
value to be: Sumida
CDRH3D16-1R5 1.5 1.6 0.043 1.8
VIN (VOUT – VIN ) CDRH4D18-1R0 1.0 1.7 0.035 2.0
LMIN =
1.4VOUT (f) CDC5D23-2R2 2.2 2.2 0.03 2.5
CR43-1R4 1.4 2.5 0.056 3.5
These conditions define the absolute minimum induc- CDRH5D28-2R6 2.6 2.6 0.013 3.0
tance. However, it is generally recommended that to CDRH6D38-3R3 3.3 3.5 0.02 4.0
prevent excessive output noise, and difficulty in obtaining CDRH6D28-3R0 3.0 3.0 0.024 3.0
stability, the ripple current is no more than 40% of the Toko
average inductor current. Since inductor ripple is: (D62F)847FY-2R4M 2.4 2.5 0.037 2.7
(D73LF)817FY-2R2M 2.2 2.7 0.03 3.0
VIN (VOUT – VIN )
IP −P RIPPLE =
VOUT (L)(f)

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APPLICATIONS INFORMATION
CATCH DIODE shutdown pin can be used. The threshold voltage of the
shutdown pin comparator is 1.35V. A 3µA internal current
The suggested catch diode (D1) is a B220A Schottky. It is
source defaults the open pin condition to be operating (see
rated at 2A average forward current and 20V reverse
Typical Performance Graphs). Current hysteresis is added
voltage. Typical forward voltage is 0.5V at 2A. The diode
above the SHDN threshold. This can be used to set voltage
conducts current only during switch off time. Peak reverse
hysteresis of the UVLO using the following:
voltage is equal to regulator output voltage. Average
forward current in normal operation is equal to output
VH − VL
current. R1 =
7µA
SHUTDOWN AND UNDERVOLTAGE LOCKOUT 1.35V
R2 =
Figure 4 shows how to add undervoltage lockout (UVLO) (VH − 1.35V) + 3µA
to the LT3436. Typically, UVLO is used in situations where R1
the input supply is current limited, or has a relatively high
source resistance. A switching regulator draws constant VH – Turn-on threshold
power from the source, so source current increases as VL – Turn-off threshold
source voltage drops. This looks like a negative resistance
Example: switching should not start until the input is
load to the source and can cause the source to current limit
above 4.75V and is to stop if the input falls below 3.75V.
or latch low under low source voltage conditions. UVLO
prevents the regulator from operating at source voltages VH = 4.75V
where these problems might occur.
VL = 3.75V

LT3436 4.75V − 3.75V

R1 = = 143k
INPUT IN
7µA 7µA
1.35V 1.35V
R2 = = 50.4k
(4.75V − 1.35V) + 3µA
R1 3µA
VCC
SHDN

143k
C1 R2
GND
3436 F04 Keep the connections from the resistors to the SHDN pin
short and make sure that the interplane or surface capaci-
Figure 4. Undervoltage Lockout tance to the switching nodes are minimized. If high resis-
tor values are used, the SHDN pin should be bypassed with
An internal comparator will force the part into shutdown a 1nF capacitor to prevent coupling problems from the
below the minimum VIN of 2.6V. This feature can be used switch node.
to prevent excessive discharge of battery-operated sys-
tems. If an adjustable UVLO threshold is required, the

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APPLICATIONS INFORMATION
SYNCHRONIZATION high speed switching current path, shown in Figure 5,
must be kept as short as possible. This is implemented in
The SYNC pin, is used to synchronize the internal oscilla-
the suggested layout of Figure 6. Shortening this path will
tor to an external signal. The SYNC input must pass from
also reduce the parasitic trace inductance of approxi-
a logic level low, through the maximum synchronization
mately 25nH/inch. At switch off, this parasitic inductance
threshold with a duty cycle between 20% and 80%. The
produces a flyback spike across the LT3436 switch. When
input can be driven directly from a logic level output. The
operating at higher currents and output voltages, with
synchronizing range is equal to initial operating frequency
poor layout, this spike can generate voltages across the
up to 1.4MHz. This means that minimum practical sync
frequency is equal to the worst-case high self-oscillating LT3436 that may exceed its absolute maximum rating. A
frequency (960kHz), not the typical operating frequency of ground plane should always be used under the switcher
800kHz. Caution should be used when synchronizing circuitry to prevent interplane coupling and overall noise.
above 1.1MHz because at higher sync frequencies the The VC and FB components should be kept as far away as
amplitude of the internal slope compensation used to possible from the switch node. The LT3436 pinout has
prevent subharmonic switching is reduced. Higher induc- been designed to aid in this. The ground for these compo-
tor values will tend to eliminate this problem. See Fre- nents should be separated from the switch current path.
quency Compensation section for a discussion of an Failure to do so will result in poor stability or subharmonic
entirely different cause of subharmonic switching before like oscillation.
assuming that the cause is insufficient slope compensa- Board layout also has a significant effect on thermal
tion. Application Note 19 has more details on the theory resistance. The exposed pad is the copper plate that runs
of slope compensation. under the LT3436 die. This is the best thermal path for heat
out of the package. Soldering the pad onto the board will
LAYOUT CONSIDERATIONS reduce die temperature and increase the power capability
As with all high frequency switchers, when considering of the LT3436. Provide as much copper area as possible
layout, care must be taken to achieve optimal electrical, around this pad. Adding multiple solder filled feedthroughs
thermal and noise performance. For maximum efficiency, under and around the pad to the ground plane will also
switch rise and fall times are typically in the nanosecond help. Similar treatment to the catch diode and inductor
range. To prevent noise both radiated and conducted, the terminations will reduce any additional heating effects.

L1
C3 D1
VOUT
SW
LT3436
HIGH
FREQUENCY
VIN SWITCHING C1 LOAD
PATH

GND

3436 F05

Figure 5. High Speed Switching Path

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APPLICATIONS INFORMATION

L1
3.9µH
D1
B220A
OUTPUT
INPUT 12V
VIN VSW
5V C3 0.8A†
OPEN LT3436 R1
4.7µF
OR 90.9k
CERAMIC
HIGH SHDN FB
= ON SYNC GND VC

R2 C1
C2 10k 22µF
10nF C4 1% CERAMIC
R3 470pF
4.7k

†MAXIMUM OUTPUT CURRENT IS SUBJECT TO THERMAL DERATING.

INPUT

L1 GND

C3 R3

C4 C2
KEEP FB AND VC
COMPONENTS
AWAY FROM
HIGH FREQUENCY,
HIGH INPUT
COMPONENTS
D1

U1 R2 R1
MINIMIZE
LT3436, GND
C1
C1, D1 LOOP

VOUT

KELVIN SENSE PLACE FEEDTHROUGHS SOLDER EXPOSED

VOUT AROUND GROUND PIN FOR GROUND PAD
GOOD THERMAL CONDUCTIVITY TO BOARD

Figure 6. Typical Application and Suggested Layout (Topside Only Shown)

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APPLICATIONS INFORMATION
The inductor must have a rating greater than its peak thermal resistance number and add in worst-case ambient
operating current to prevent saturation resulting in effi- temperature:
ciency loss. Peak inductor current is given by:
TJ = TA + θJA (PTOT)
(VOUT )(IOUT ) VIN (VOUT − VIN ) If a true die temperature is required, a measurement of
ILPEAK = + the SYNC to GND pin resistance can be used. The SYNC
VIN • η 2VOUT (L)(f)
pin resistance across temperature must first be cali-
Also, consideration should be given to the DC resistance brated, with no device power, in an oven. The same
of the inductor. Inductor resistance contributes directly to measurement can then be used in operation to indicate the
the efficiency losses in the overall converter. die temperature.

THERMAL CALCULATIONS FREQUENCY COMPENSATION

Power dissipation in the LT3436 chip comes from four Loop frequency compensation is performed on the output
sources: switch DC loss, switch AC loss, drive current, and of the error amplifier (VC pin) with a series RC network.
input quiescent current. The following formulas show how The main pole is formed by the series capacitor and the
to calculate each of these losses. These formulas assume output impedance (≈500kΩ) of the error amplifier. The
continuous mode operation, so they should not be used pole falls in the range of 2Hz to 20Hz. The series resistor
for calculating efficiency at light load currents. creates a “zero” at 1kHz to 5kHz, which improves loop
stability and transient response. A second capacitor, typi-
(VOUT − VIN ) cally one-tenth the size of the main compensation capaci-
DC, duty cycle = tor, is sometimes used to reduce the switching frequency
VOUT
ripple on the VC pin. VC pin ripple is caused by output
(V )(I )
ISW = OUT OUT voltage ripple attenuated by the output divider and multi-
VIN plied by the error amplifier. Without the second capacitor,
VC pin ripple is:
Switch loss:
(
PSW = (DC )(ISW )2 (RSW ) + 17n(ISW ) VOUT ( f)) 1.2(VRIPPLE)(gm)(RC)
VC Pin Ripple =
VIN loss: (VOUT)
(VIN )(ISW )(DC ) VRIPPLE = Output ripple (VP–P)
PVIN = + 1mA(VIN )
50 gm = Error amplifier transconductance
RSW = Switch resistance (≈ 0.16Ω hot) (≈850µmho)
RC = Series resistor on VC pin
Example: VIN = 5V, VOUT = 12V and IOUT = 0.8A VOUT = DC output voltage

Total power dissipation = 0.34 + 0.31 + 0.11 + 0.005 = To prevent irregular switching, VC pin ripple should be
0.77W kept below 50mVP–P. Worst-case VC pin ripple occurs at
Thermal resistance for LT3436 package is influenced by maximum output load current and will also be increased if
the presence of internal or backside planes. With a full poor quality (high ESR) output capacitors are used. The
plane under the package, thermal resistance will be about addition of a 150pF capacitor on the VC pin reduces
40°C/W. To calculate die temperature, use the appropriate switching frequency ripple to only a few millivolts. A low
value for RC will also reduce VC pin ripple, but loop phase
margin may be inadequate.

3436fa

11
LT3436
U
TYPICAL APPLICATIO S
Load Disconnects in Shutdown
D3
1N4148

L1 C6 D2
3.9µH 0.1µF 1N4148

D1 C5 R4
B220A 0.1µF 1M
VOUT
VIN VIN VSW 12V
5V Q1
C3 C1 C7 0.8A
4.7µF LT3436 Si2306DS
4.7µF 22µF
SHDN FB R1
OFF ON 90.9k
SYNC GND VC

R2
C2 10k
10nF 1%
R3 C4
4.7k 470pF

LT3436 • TA02

3V to 20VIN 5VOUT SEPIC with Either Two Inductors or a Transformer

L1 D1
CDRH6D28-100 B220A
VIN VOUT
3V TO 20V + C1 C6
5V
C7 C5 R1
OPT 1µF, X5R, 25V OPT OPT
VIN SW 31.6K
CERAMIC 1%
SHDN SHDN FB
LT3436 C3
10nF C2
SYNC SYNC VC L2
CDRH6D28-100 22µF
GND GND X5R
R2 10V
C1 C4 10K CERAMIC
R3 1%
4.7µF 470pF 2.2k
X5R
25V
CERAMIC
GND GND

OPTION: REPLACE L1, L2 WITH TRANSFORMER CTX5-1A, CTX8-1A, CTX10-2A 3436 TA02b

Maximum Load Current

Increases with Input Voltage Efficiency
2.0 100
1.8 90
MAXIMUM LOAD CURRENT (A)

1.6 80 12VIN
1.4 70
EFFICIENCY (%)

3.3VIN 5VIN
1.2 60
1.0 50
0.8 40
0.6 30
0.4 20
0.2 10
0 0
0 2 4 6 8 10 12 14 16 18 20 0 500 1.0k 1.5k 2.0k
VIN (V) LOAD CURRENT (mA)
3436 TA02c 3436 TA02d

3436fa

12
 LT3436
U
TYPICAL APPLICATIO S
4V-9VIN to 5VOUT SEPIC Converter**

VIN**
4V TO 9V

L1A*
15µH D1
VIN B220A
ON • VOUT†
SHDN VSW
OFF R2 5V
LT3436 C2 31.6k
4.7µF 1%
+ C1 FB
•
4.7µF
20V
GND VC + C3
L1B* 47µF
15µH 10V
R1 R3
2.2k C5 10k
470pF 1%
C4
15nF
LT3436 • TA03
†MAX I
OUT
* COILTRONICS CTX15-4 IOUT VIN
** INPUT VOLTAGE MAY BE GREATER OR 0.84A 4V
LESS THAN OUTPUT VOLTAGE 1.03A 5V
1.18A 6V
1.29A 7V
1.50A 9V

Boost Converter Drives Luxeon III 1A 3.6V White LED with 70% Efficiency

0.05Ω
1% 1A CONSTANT CURRENT LXHL-PW09 EMITTER
VIN
VOUT = VIN + VLED
3.3V TO 4.2V
49.9Ω
L1 UPS120 1%

VIN

VIN SW +
SHDN FB LT1783
LED ON LT3436 –
SYNC VC VOUT Q2
GND GND
4.7µF
X5R Q1 78.7k
6.3V 0.1µF 22µF
CERAMIC X5R
10V
8.2k 4.99k 1.21k CERAMIC
1%
GND
3436 TA03b

Q1: MMBT2222A
Q2: FMMT3906
L1: CDRH6D28-3R0

3436fa

13
LT3436
U
TYPICAL APPLICATIO S
Single Li-Ion Cell to 5V
D1
L1 B220A
4.7µH
VOUT
5V
VIN R1
ON VSW 31.6k
OFF SHDN 1%
LT3436
+ SINGLE + C1 FB + C4
Li-Ion 47µF
10µF GND VC
CELL 10V
R2
10k
C2
1%
3.3nF
R3 C3
1.5k 470pF

LT3436 • TA04

IOUT VIN
1.5A 2.7V
1.86A 3.3V
2.0A 3.6V

SEPIC Converter Drives 5W LumiLEDs Luxeon V White LEDs at 70% Efficiency

D1
B130A
VOUT
CCOUP
2.2µF, X5R, 25V D2
L1 CERAMIC L2
VIN
3.6V TO 17V VIN
LED ON
VIN SW + 700mA

SHDN FB LT1783
LT3436 –
SYNC VC R5
C1 23.7k
GND GND
4.7µF
X5R R7
25V Q1
C4 124k C2
CERAMIC 0.1µF 22µF
VOUT
R4 R2 X5R
8.2k R6 1k 0.068Ω 16V
4.99k 1% 1% CERAMIC
GND
3436 TA04b

Q1: DIODES, INC. MMBT2222A

L1: CDRH6D28 10µH 1.7A
L2: CDRH4D28 10µH 1A
D2: LUMILEDS LXHL-PW03 EMITTER OR LXHL-LW6C STAR

3436fa

14
 LT3436
U
PACKAGE DESCRIPTION
FE Package
16-Lead Plastic TSSOP (4.4mm)
(Reference LTC DWG # 05-08-1663)
Exposed Pad Variation BB

4.90 – 5.10*
3.58 (.193 – .201)
(.141)
3.58
(.141)
16 1514 13 12 1110 9

6.60 ±0.10
2.94
4.50 ±0.10 (.116)
SEE NOTE 4 2.94 6.40
(.116) (.252)
0.45 ±0.05 BSC

1.05 ±0.10

0.65 BSC
RECOMMENDED SOLDER PAD LAYOUT 1 2 3 4 5 6 7 8
1.10
4.30 – 4.50* (.0433)
(.169 – .177) 0.25 MAX
REF
0° – 8°

0.65
0.09 – 0.20 0.50 – 0.75 (.0256) 0.05 – 0.15
(.0035 – .0079) (.020 – .030) BSC (.002 – .006)
0.195 – 0.30
FE16 (BB) TSSOP 0204
(.0077 – .0118)
TYP
NOTE:
1. CONTROLLING DIMENSION: MILLIMETERS 4. RECOMMENDED MINIMUM PCB METAL SIZE
MILLIMETERS FOR EXPOSED PAD ATTACHMENT
2. DIMENSIONS ARE IN
(INCHES) *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
3. DRAWING NOT TO SCALE SHALL NOT EXCEED 0.150mm (.006") PER SIDE

3436fa

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 15
LT3436
U
TYPICAL APPLICATIO
High Voltage Laser Power Supply
0.01µF 1800pF
5kV 10kV 47k
5W

1800pF
10kV

L1 11 8 HV DIODES

5 4 1 3 LASER
2

+
2.2µF
Q1 0.47µF Q2

150Ω
L2
MUR405 10µH

VSW 10k 10k

VIN
VIN FB 1N4002
12V TO 25V
+ LT3436 0.1µF VIN (ALL) L1 = TBD
2.2µF 190Ω Q1, Q2 = ZETEX ZTX849
1% 0.47µF = WIMA 3X 0.15µF TYPE MKP-20
VC GND
HV DIODES = SEMTECH-FM-50
+ LASER = HUGHES 3121H-P
10µF LT3436 • TA05 COILTRONICS (407) 241-7876

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3436fa

Linear Technology Corporation LT/LWI/LT 0505 REV A • PRINTED IN USA

16 1630 McCarthy Blvd., Milpitas, CA 95035-7417

(408) 432-1900 ● FAX: (408) 434-0507 ●
www.linear.com © LINEAR TECHNOLOGY CORPORATION 2003
Mouser Electronics

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