a CMOS 300 MSPS Quadrature

Complete-DDS
AD9854
FEATURES Multiple Power-Down Functions
300 MHz Internal Clock Rate Single-Ended or Differential Input Reference Clock
FSK, BPSK, PSK, CHIRP, AM Operation Small 80-Lead LQFP Packaging
Dual Integrated 12-Bit D/A Converters
APPLICATIONS
Ultrahigh-Speed Comparator, 3 ps RMS Jitter
Agile, Quadrature L.O. Frequency Synthesis
Excellent Dynamic Performance: 80 dB SFDR @ 100 MHz
Programmable Clock Generator
(1 MHz) AOUT
FM Chirp Source for Radar and Scanning Systems
4
to 20
Programmable Reference Clock Multiplier
Test and Measurement Equipment
Dual 48-Bit Programmable Frequency Registers
Commercial and Amateur RF Exciter
Dual 14-Bit Programmable Phase Offset Registers
12-Bit Amplitude Modulation and Programmable
Shaped On/Off Keying Function GENERAL DESCRIPTION
Single Pin FSK and BPSK Data Interface The AD9854 digital synthesizer is a highly integrated device
PSK Capability Via I/O Interface that uses advanced DDS technology, coupled with two internal
Linear or Nonlinear FM Chirp Functions with Single high-speed, high-performance quadrature D/A converters to
Pin Frequency “Hold” Function form a digitally programmable I and Q synthesizer function. When
Frequency-Ramped FSK referenced to an accurate clock source, the AD9854 generates
<25 ps RMS Total Jitter in Clock Generator Mode highly stable, frequency-phase amplitude-programmable sine
Automatic Bidirectional Frequency Sweeping and cosine outputs that can be used as an agile L.O. in com-
SIN(x)/x Correction munications, radar, and many other applications. The AD9854’s
Simplified Control Interface innovative high-speed DDS core provides 48-bit frequency
10 MHz Serial, 2-Wire or 3-Wire SPI-Compatible or resolution (1 microHertz tuning resolution with 300 MHz
100 MHz Parallel 8-Bit Programming SYSCLK). Maintaining 17 bits assures excellent SFDR. The
3.3 V Single Supply AD9854’s circuit architecture allows the generation of
(continued on page 14)

FUNCTIONAL BLOCK DIAGRAM

SYSTEM CLOCK DDS CORE DIGITAL MULTIPLIERS

4
–20
INV.
REF REF CLK I SINC MUX
MUX

REFERENCE 12-BIT “I”

ACCUMULATOR

ACCUMULATOR

CLK MULTI- FILTER ANALOG

CLOCK IN DAC
FREQUENCY

12
CONVERTER

BUFFER OUT
PHASE-TO-
AMPLITUDE

PLIER 12
PHASE

48 48 17 17
ACC 1

ACC 2

SYSTEM
CLOCK DAC RSET
DIFF/SINGLE MUX
SELECT SYSTEM
14 INV. 12-BIT
MUX

CLOCK 48
MUX

Q SINC “Q” DAC OR ANALOG

FILTER CONTROL OUT
12 12
D 12 DAC
FSK/BPSK/HOLD E 3 12
M MUX MUX MUX
DATA IN U
X DELTA PROGRAMMABLE
SYSTEM AMPLITUDE AND ANALOG
FREQUENCY CLOCK
RATE TIMER RATE CONTROL IN
2
COMPARATOR
48 SYSTEM 48 48 14
CLOCK 14 12 12
CLOCK
DELTA FREQUENCY FREQUENCY 1ST 14-BIT PHASE/ 2ND 14-BIT PHASE/ I AND Q 12-BIT 12-BIT DC OUT
FREQUENCY TUNING TUNING OFFSET WORD OFFSET WORD AM MODULATION CONTROL
WORD WORD 1 WORD 2
MODE SELECT PROGRAMMING REGISTERS SHAPED
ON/OFF
SYSTEM KEYING
CK SYSTEM AD9854
CLOCK 2 CLOCK BUS
Q
D
BIDIRECTIONAL GND
INT INTERNAL I/O PORT BUFFERS
INTERNAL/EXTERNAL PROGRAMMABLE
I/O UPDATE UPDATE CLOCK +VS
CLOCK EXT

READ WRITE SERIAL/ 6-BIT ADDRESS 8-BIT MASTER

PARALLEL OR SERIAL PARALLEL RESET
SELECT PROGRAMMING LOAD
LINES
REV. B
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties that One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
may result from its use. No license is granted by implication or otherwise Tel: 781/329-4700 www.analog.com
under any patent or patent rights of Analog Devices. Fax: 781/326-8703 © Analog Devices, Inc., 2002
AD9854
TABLE OF CONTENTS
FEATURES . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 GENERAL OPERATION OF THE
GENERAL DESCRIPTION . . . . . . . . . . . . . . . . . . . . . . . . . 1 SERIAL INTERFACE . . . . . . . . . . . . . . . . . . . . . . . . . . 26
FUNCTIONAL BLOCK DIAGRAM . . . . . . . . . . . . . . . . . 1 Instruction Byte . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
TABLE OF CONTENTS . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 Serial Interface Port Pin Description . . . . . . . . . . . . . . . . 27
SPECIFICATIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 Notes on Serial Port Operation . . . . . . . . . . . . . . . . . . . . 27
EXPLANATION OF TEST LEVELS . . . . . . . . . . . . . . . . . 5 MSB/LSB TRANSFERS . . . . . . . . . . . . . . . . . . . . . . . . . . . 28
Test Level . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 Control Register Description . . . . . . . . . . . . . . . . . . . . . . 28
ABSOLUTE MAXIMUM RATINGS . . . . . . . . . . . . . . . . . 5 POWER DISSIPATION AND
ORDERING GUIDE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 THERMAL CONSIDERATIONS . . . . . . . . . . . . . . . . . 29
PIN FUNCTION DESCRIPTIONS . . . . . . . . . . . . . . . . . . 6 THERMAL IMPEDANCE . . . . . . . . . . . . . . . . . . . . . . . . . 30
TYPICAL PERFORMANCE CHARACTERISTICS . . . . . 8 JUNCTION TEMPERATURE CONSIDERATIONS . . . . 30
PIN CONFIGURATION . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 EVALUATION OF OPERATING CONDITIONS . . . . . . 31
TYPICAL APPLICATIONS . . . . . . . . . . . . . . . . . . . . . . . 12 THERMALLY ENHANCED PACKAGE
OVERVIEW . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14 MOUNTING GUIDELINES . . . . . . . . . . . . . . . . . . . . 31
DESCRIPTION OF AD9854 MODES OF OPERATION . . 14 EVALUATION BOARD . . . . . . . . . . . . . . . . . . . . . . . . . . 32
Single-Tone (Mode 000) . . . . . . . . . . . . . . . . . . . . . . . . . 14 EVALUATION BOARD INSTRUCTIONS . . . . . . . . . . . 32
Unramped FSK (Mode 001) . . . . . . . . . . . . . . . . . . . . . . 15 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32
Ramped FSK (Mode 010) . . . . . . . . . . . . . . . . . . . . . . . . 15 GENERAL OPERATING INSTRUCTIONS . . . . . . . . . . 32
Chirp (Mode 011) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 Attach REFCLK . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33
Basic FM Chirp Programming Steps . . . . . . . . . . . . . . . . 19 Clock Input, J25 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33
BPSK (Mode 100) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21 Three-State Control . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33
USING THE AD9854 . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21 Programming . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33
Internal and External Update Clock . . . . . . . . . . . . . . . . . 21 Low-Pass Filter Testing . . . . . . . . . . . . . . . . . . . . . . . . . . 33
Shaped On/Off Keying . . . . . . . . . . . . . . . . . . . . . . . . . . . 21 Observing the Unfiltered IOUT1 and the
I and Q DACs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22 Unfiltered IOUT2 DAC Signals . . . . . . . . . . . . . . . . . . . 33
Control DAC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23 Observing the Filtered IOUT1 and the Filtered IOUT2 . . . . 33
Inverse SINC Function . . . . . . . . . . . . . . . . . . . . . . . . . . 23 Observing the Filtered IOUT and the Filtered IOUTB . . . . . 33
REFCLK Multiplier . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23 To Connect the High-Speed Comparator . . . . . . . . . . . . 34
PROGRAMMING THE AD9854 . . . . . . . . . . . . . . . . . . . 24 Single-Ended Configuration . . . . . . . . . . . . . . . . . . . . . . . 34
Parallel I/O Operation . . . . . . . . . . . . . . . . . . . . . . . . . . . 26 USING THE PROVIDED SOFTWARE . . . . . . . . . . . . . . 34
Serial Port I/O Operation . . . . . . . . . . . . . . . . . . . . . . . . . 26 OUTLINE DIMENSIONS . . . . . . . . . . . . . . . . . . . . . . . . . 41
REVISION HISTORY . . . . . . . . . . . . . . . . . . . . . . . . . . . . 42

–2– REV. B
 AD9854
SPECIFICATIONS (VS = 3.3 V  5%, RSET = 3.9 k external reference clock frequency = 30 MHz with
REFCLK Multiplier enabled at 10
for AD9854ASQ, external reference clock frequency = 20 MHz with REFCLK Multiplier
enabled at 10
for AD9854AST unless otherwise noted.)
Test AD9854ASQ AD9854AST
Parameter Temp Level Min Typ Max Min Typ Max Unit
1
REF CLOCK INPUT CHARACTERISTICS
Internal System Clock Frequency Range
REFCLK Multiplier Enabled Full VI 20 300 20 200 MHz
REFCLK Multiplier Disabled Full VI DC 300 DC 200 MHz
External REF Clock Frequency Range
REFCLK Multiplier Enabled Full VI 5 75 5 50 MHz
REFCLK Multiplier Disabled Full VI DC 300 DC 200 MHz
Duty Cycle 25°C IV 45 50 55 45 50 55 %
Input Capacitance 25°C IV 3 3 pF
Input Impedance 25°C IV 100 100 kΩ
Differential Mode Common-Mode Voltage Range
Minimum Signal Amplitude2 25°C IV 800 800 mV p-p
Common-Mode Range 25°C IV 1.6 1.75 1.9 1.6 1.75 1.9 V
VIH (Single-Ended Mode) 25°C IV 2.3 2.3 V
VIL (Single-Ended Mode) 25°C IV 1 1 V
DAC STATIC OUTPUT CHARACTERISTICS
Output Update Speed Full I 300 200 MSPS
Resolution 25°C IV 12 12 Bits
I and Q Full-Scale Output Current 25°C IV 5 10 20 5 10 20 mA
I and Q DAC DC Gain Imbalance3 25°C I –0.5 +0.15 +0.5 –0.5 +0.15 +0.5 dB
Gain Error 25°C I –6 +2.25 –6 +2.25 % FS
Output Offset 25°C I 2 2 µA
Differential Nonlinearity 25°C I 0.3 1.25 0.3 1.25 LSB
Integral Nonlinearity 25°C I 0.6 1.66 0.6 1.66 LSB
Output Impedance 25°C IV 100 100 kΩ
Voltage Compliance Range 25°C I –0.5 +1.0 –0.5 +1.0 V
DAC DYNAMIC OUTPUT CHARACTERISTICS
I and Q DAC Quad. Phase Error 25°C IV 0.2 1 0.2 1 Degrees
DAC Wideband SFDR
1 MHz to 20 MHz AOUT 25°C V 58 58 dBc
20 MHz to 40 MHz AOUT 25°C V 56 56 dBc
40 MHz to 60 MHz AOUT 25°C V 52 52 dBc
60 MHz to 80 MHz AOUT 25°C V 48 48 dBc
80 MHz to 100 MHz AOUT 25°C V 48 48 dBc
100 MHz to 120 MHz AOUT 25°C V 48 dBc
DAC Narrowband SFDR
10 MHz AOUT (± 1 MHz) 25°C V 83 83 dBc
10 MHz AOUT (± 250 kHz) 25°C V 83 83 dBc
10 MHz AOUT (± 50 kHz) 25°C V 91 91 dBc
41 MHz AOUT (± 1 MHz) 25°C V 82 82 dBc
41 MHz AOUT (± 250 kHz) 25°C V 84 84 dBc
41 MHz AOUT (± 50 kHz) 25°C V 89 89 dBc
119 MHz AOUT (± 1 MHz) 25°C V 71 dBc
119 MHz AOUT (± 250 kHz) 25°C V 77 dBc
119 MHz AOUT (± 50 kHz) 25°C V 83 dBc
Residual Phase Noise
(AOUT = 5 MHz, Ext. CLK = 30 MHz,
REFCLK Multiplier Engaged at 10×)
1 kHz Offset 25°C V 140 140 dBc/Hz
10 kHz Offset 25°C V 138 138 dBc/Hz
100 kHz Offset 25°C V 142 142 dBc/Hz
(AOUT = 5 MHz, Ext. CLK = 300 MHz,
REFCLK Multiplier Bypassed)
1 kHz Offset 25°C V 142 142 dBc/Hz
10 kHz Offset 25°C V 148 148 dBc/Hz
100 kHz Offset 25°C V 152 152 dBc/Hz

REV. B –3–
AD9854–SPECIFICATIONS
Test AD9854ASQ AD9854AST
Parameter Temp Level Min Typ Max Min Typ Max Unit
Pipeline Delays4, 5, 6
DDS Core (Phase Accumulator and 25°C IV 33 33 SysClk Cycles
Phase to Amp Converter)
Frequency Accumulator 25°C IV 26 26 SysClk Cycles
Inverse Sinc Filter 25°C IV 16 16 SysClk Cycles
Digital Multiplier 25°C IV 9 9 SysClk Cycles
DAC 25°C IV 1 1 SysClk Cycles
I/O Update Clock (INT MODE) 25°C IV 2 2 SysClk Cycles
I/O Update Clock (EXT MODE) 25°C IV 3 3 SysClk Cycles
MASTER RESET DURATION 25°C IV 10 10 SysClk Cycles
COMPARATOR INPUT CHARACTERISTICS
Input Capacitance 25°C V 3 3 pF
Input Resistance 25°C IV 500 500 kΩ
Input Current 25°C I ±1 ±5 ±1 ±5 µA
Hysteresis 25°C IV 10 20 10 20 mV p-p
COMPARATOR OUTPUT CHARACTERISTICS
Logic “1” Voltage, High Z Load Full VI 3.1 3.1 V
Logic “0” Voltage, High Z Load Full VI 0.16 0.16 V
Output Power, 50 Ω Load, 120 MHz Toggle Rate 25°C I 9 11 9 11 dBm
Propagation Delay 25°C IV 3 3 ns
Output Duty Cycle Error7 25°C I –10 ±1 +10 –10 ±1 +10 %
Rise/Fall Time, 5 pF Load 25°C V 2 2 ns
Toggle Rate, High Z Load 25°C IV 300 350 300 350 MHz
Toggle Rate, 50 Ω Load 25°C IV 375 400 375 400 MHz
Output Cycle-to-Cycle Jitter8 25°C IV 4.0 4.0 ps rms
COMPARATOR NARROWBAND SFDR9
10 MHz (± 1 MHz) 25°C V 84 84 dBc
10 MHz (± 250 kHz) 25°C V 84 84 dBc
10 MHz (± 50 kHz) 25°C V 92 92 dBc
41 MHz (± 1 MHz) 25°C V 76 76 dBc
41 MHz (± 250 kHz) 25°C V 82 82 dBc
41 MHz (± 50 kHz) 25°C V 89 89 dBc
119 MHz (± 1 MHz) 25°C V 73 dBc
119 MHz (± 250 kHz) 25°C V 73 dBc
119 MHz (± 50 kHz) 25°C V 83 dBc
CLOCK GENERATOR OUTPUT JITTER9
5 MHz AOUT 25°C V 23 23 ps rms
40 MHz AOUT 25°C V 12 12 ps rms
100 MHz AOUT 25°C V 7 7 ps rms
PARALLEL I/O TIMING CHARACTERISTICS
TASU (Address Setup Time to WR Signal Active) Full IV 8.0 7.5 8.0 7.5 ns
TADHW (Address Hold Time to WR Signal Inactive) Full IV 0 0 ns
TDSU (Data Setup Time to WR Signal Inactive) Full IV 3.0 1.6 3.0 1.6 ns
TDHD (Data Hold Time to WR Signal Inactive) Full IV 0 0 ns
TWRLOW (WR Signal Minimum Low Time) Full IV 2.5 1.8 2.5 1.8 ns
TWRHIGH (WR Signal Minimum High Time) Full IV 7 7 ns
TWR (Minimum Write Time) Full IV 10.5 10.5 ns
TADV (Address to Data Valid Time) Full V 15 15 15 15 ns
TADHR (Address Hold Time to RD Signal Inactive) Full IV 5 5 ns
TRDLOV (RD Low-to-Output Valid) Full IV 15 15 ns
TRDHOZ (RD High-to-Data Three-State) Full IV 10 10 ns
SERIAL I/O TIMING CHARACTERISTICS
TPRE (CS Setup Time) Full IV 30 30 ns
TSCLK (Period of Serial Data Clock) Full IV 100 100 ns
TDSU (Serial Data Setup Time) Full IV 30 30 ns
TSCLKPWH (Serial Data Clock Pulsewidth High) Full IV 40 40 ns
TSCLKPWL (Serial Data Clock Pulsewidth Low) Full IV 40 40 ns
TDHLD (Serial Data Hold Time) Full IV 0 0 ns
TDV (Data Valid Time) Full V 30 30 ns

–4– REV. B
 AD9854
Test AD9854ASQ AD9854AST
Parameter Temp Level Min Typ Max Min Typ Max Unit
CMOS LOGIC INPUTS10
Logic “1” Voltage 25°C I 2.2 2.2 V
Logic “0” Voltage 25°C I 0.8 0.8 V
Logic “1” Current 25°C IV ±5 ± 12 µA
Logic “0” Current 25°C IV ±5 ± 12 µA
Input Capacitance 25°C V 3 3 pF
11
POWER SUPPLY
+VS Current12 25°C I 1050 1210 755 865 mA
+VS Current13 25°C I 710 816 515 585 mA
+VS Current14 25°C I 600 685 435 495 mA
PDISS12 25°C I 3.475 4.190 2.490 3.000 W
PDISS13 25°C I 2.345 2.825 1.700 2.025 W
PDISS14 25°C I 1.975 2.375 1.435 1.715 W
PDISS Power-Down Mode 25°C I 1 50 1 50 mW
NOTES
1
The reference clock inputs are configured to accept a 1 V p-p (typical) dc offset square or sine waves centered at one-half the applied V DD or a 3 V TTL-level pulse input.
2
An internal 800 mV p-p differential voltage swing equates to 400 mV p-p applied to both REFCLK input pins.
3
The I and Q gain imbalance is digitally adjustable to less than 0.01 dB.
4
Pipeline delays of each individual block are fixed; however, if the 8 top MSBS of a tuning word are all zeros, the delay will appear longer. This is due to insufficient
phase accumulation per a system CLK period to produce enough LSB amplitude to the D/A converter.
5
If a feature like the Inverse Sinc, which has 16 Pipeline delays, can be bypassed, the total delay will be reduced by that amount.
6
The I/O Update CLK transfers data from the I/O Port Buffers to the Programming Registers. This transfer takes system clocks to perform.
7
Change in duty cycle from 1 MHz to 100 MHz with 1 V p-p sine wave input and 0.5 V threshold.
8
Represents comparator’s inherent cycle-to-cycle jitter contribution. Input signal is a 1 V, 40 MHz square wave. Measurement device Wavecrest DTS – 2075.
9
Comparator input originates from analog output section via external 7-pole elliptic LPF. Single-ended input, 0.5 V p-p. Comparator output terminated in 50 Ω.
10
Avoid overdriving digital inputs. (Refer to equivalent circuits in Figure 1.)
11
Simultaneous operation at the maximum ambient temperature of 85 °C and the maximum internal clock frequency of 200 MHz for the 80-lead LQFP, or 300 MHz
for the thermally enhanced 80-lead LQFP may cause the maximum die junction temperature of 150 °C to be exceeded. Refer to the Power Dissipation section
and Thermal Considerations for derating and thermal management information.
12
All functions engaged.
13
All functions except inverse sinc engaged.
14
All functions except inverse sinc and digital multipliers engaged. ABSOLUTE MAXIMUM RATINGS*
Specifications subject to change without notice. Maximum Junction Temperature . . . . . . . . . . . . . . . . 150°C
VS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 V
EXPLANATION OF TEST LEVELS Digital Inputs . . . . . . . . . . . . . . . . . . . . . . . . . –0.7 V to +VS
Test Level Digital Output Current . . . . . . . . . . . . . . . . . . . . . . . . . 5 mA
I – 100% Production Tested. Storage Temperature . . . . . . . . . . . . . . . . . . –65°C to +150°C
III – Sample Tested Only. Operating Temperature . . . . . . . . . . . . . . . . . –40°C to +85°C
IV – Parameter is guaranteed by design and characterization Lead Temperature (Soldering, 10 sec) . . . . . . . . . . . . 300°C
testing. Maximum Clock Frequency (ASQ) . . . . . . . . . . . . . 300 MHz
V – Parameter is a typical value only. Maximum Clock Frequency (AST) . . . . . . . . . . . . . 200 MHz
VI – Devices are 100% production tested at 25°C and θJA (ASQ) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16°C/W
guaranteed by design and characterization testing θJA (AST) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38°C/W
for industrial operating temperature range. θJC (ASQ) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2°C/W
*Absolute Maximum Ratings are limiting values, to be applied individually, and
beyond which the serviceability of the circuit may be impaired. Functional
operability under any of these conditions is not necessarily implied. Exposure of
absolute maximum rating conditions for extended periods of time may affect device
reliability.

ORDERING GUIDE

Model Temperature Range Package Description Package Option

AD9854ASQ –40°C to +85°C Thermally Enhanced 80-Lead LQFP SQ-80
AD9854AST –40°C to +85°C 80-Lead LQFP ST-80
AD9854/PCB 0°C to 70°C Evaluation Board

CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection. Although WARNING!
the AD9854 features proprietary ESD protection circuitry, permanent damage may occur on
devices subjected to high-energy electrostatic discharges. Therefore, proper ESD precautions are
ESD SENSITIVE DEVICE
recommended to avoid performance degradation or loss of functionality.

REV. B –5–
AD9854
PIN FUNCTION DESCRIPTIONS

Pin
No. Pin Name Function
1–8 D7–D0 Eight-Bit Bidirectional Parallel Programming Data Inputs. Used only in parallel programming mode.
9, 10, 23, DVDD Connections for the Digital Circuitry Supply Voltage. Nominally 3.3 V more positive than AGND
24, 25, 73, and DGND.
74, 79, 80
11, 12, 26, DGND Connections for Digital Circuitry Ground Return. Same potential as AGND.
27, 28, 72,
75, 76, 77,
78
13, 35, 57, NC No Internal Connection
58, 63
14–19 A5–A0 Six-Bit Parallel Address Inputs for Program Registers. Used only in parallel programming mode. A0, A1,
and A2 have a second function when the serial programming mode is selected. See immediately below.
(17) A2/IO RESET Allows an IO RESET of the serial communications bus that is unresponsive due to improper program-
ming protocol. Resetting the serial bus in this manner does not affect previous programming nor
does it invoke the “default” programming values seen in the Table IV. Active HIGH.
(18) A1/SDO Unidirectional Serial Data Output for Use in 3-Wire Serial Communication Mode.
(19) A0/SDIO Bidirectional Serial Data Input/Output for Use in 2-Wire Serial Communication Mode.
20 I/O UD CLK Bidirectional I/O Update CLK. Direction is selected in control register. If selected as an input,
a rising edge will transfer the contents of the I/O Port Buffers to the Programming Registers. If I/O UD is
selected as an output (default), an output pulse (low to high) of eight system clock cycle duration
indicates that an internal frequency update has occurred.
21 WRB/SCLK Write Parallel Data to I/O Port Buffers. Shared function with SCLK. Serial clock signal associated with
the serial programming bus. Data is registered on the rising edge. This pin is shared with WRB when the
parallel mode is selected. Mode dependent on Pin 70 (5/p select).
22 RDB/CSB Read Parallel Data from Programming Registers. Shared function with CSB. Chip-select signal associated
with the serial programming bus. Active LOW. This pin is shared with RDB when the parallel mode is selected.
29 FSK/BPSK/ Multifunction Pin According to the Mode of Operation Selected in the Programming Control Register.
HOLD If in the FSK mode logic low selects F1, logic high selects F2. If in the BPSK mode, logic low selects
Phase 1, logic high selects Phase 2. If in the Chirp mode, logic high engages the HOLD function causing
the frequency accumulator to halt at its current location. To resume or commence Chirp, logic low is asserted.
30 SHAPED Must First Be Selected in the Programming Control Register to Function. A logic high will cause the
KEYING I and Q DAC outputs to ramp up from zero-scale to full-scale amplitude at a preprogrammed rate.
Logic low causes the full-scale output to ramp down to zero-scale at the preprogrammed rate.
31, 32, 37, AVDD Connections for the Analog Circuitry Supply Voltage. Nominally 3.3 V more positive than AGND
38, 44, 50, and DGND.
54, 60, 65
33, 34, 39, AGND Connections for Analog Circuitry Ground Return. Same potential as DGND.
40, 41, 45,
46, 47, 53,
59, 62, 66,
67
36 VOUT Internal High-Speed Comparator’s Noninverted Output Pin. Designed to drive 10 dBm to 50 Ω load
as well as standard CMOS logic levels.
42 VINP Voltage Input Positive. The internal high-speed comparator’s noninverting input.
43 VINN Voltage Input Negative. The internal high-speed comparator’s inverting input.
48 IOUT1 Unipolar Current Output of the I or Cosine DAC. (Refer to Figure 1)
49 IOUT1B Complementary Unipolar Current Output of the I or Cosine DAC.
51 IOUT2B Complementary Unipolar Current Output of the Q or Sine DAC.
52 IOUT2 Unipolar Current Output of the Q or Sine DAC. This DAC can be programmed to accept external
12-bit data in lieu of internal sine data. This allows the AD9854 to emulate the AD9852 control
DAC function.

–6– REV. B
 AD9854
Pin
No. Pin Name Function
55 DACBP Common Bypass Capacitor Connection for Both I and Q DACs. A 0.01 µF chip cap from this pin to
AVDD improves harmonic distortion and SFDR slightly. No connect is permissible (slight SFDR
degradation).
56 DAC RSET Common Connection for Both I and Q DACs to Set the Full-Scale Output Current. RSET = 39.9/IOUT.
Normal RSET range is from 8 kΩ (5 mA) to 2 kΩ (20 mA).
61 PLL FILTER This pin provides the connection for the external zero compensation network of the REFCLK
Multiplier’s PLL loop filter. The zero compensation network consists of a 1.3 kΩ resistor in series
with a 0.01 µF capacitor. The other side of the network should be connected to AVDD as close as
possible to Pin 60. For optimum phase noise performance, the REFCLK Multiplier can be bypassed
by setting the “Bypass PLL” bit in control register 1E.
64 DIFF CLK Differential REFCLK Enable. A high level of this pin enables the differential clock inputs, REFCLK
ENABLE and REFCLKB (Pins 69 and 68 respectively).
68 REFCLKB The Complementary (180 Degrees Out-of-Phase) Differential Clock Signal. User should tie this pin
high or low when single-ended clock mode is selected. Same signal levels as REFCLK.
69 REFCLK Single-Ended (CMOS logic levels required) Reference Clock Input or One of Two Differential Clock
Signals. In Differential Ref Clock mode, both inputs can be CMOS logic levels or have greater than
400 mV p-p square or sine waves centered about 1.6 V dc.
70 S/P SELECT Selects Between Serial Programming Mode (Logic LOW) and Parallel Programming Mode
(Logic High).
71 MASTER Initializes the serial/parallel programming bus to prepare for user programming; sets programming
RESET registers to a “do-nothing” state defined by the default values seen in the Table IV. Active on logic
high. Asserting MASTER RESET is essential for proper operation upon power-up.

PIN CONFIGURATION
DIFF CLK ENABLE
MASTER RESET

S/P SELECT

PLL FILTER
REFCLKB
REFCLK
DGND
DGND

DGND
DGND

DGND

AGND

AGND

AGND
DVDD
DVDD

DVDD
DVDD

AVDD

NC

80 79 78 77 76 75 74 73 72 71 70 69 68 67 66 65 64 63 62 61

60 AVDD
D7 1 PIN 1
IDENTIFIER 59 AGND
D6 2
58 NC
D5 3
57 NC
D4 4
56 DAC RSET
D3 5
55 DACBP
D2 6
54 AVDD
D1 7
D0 8 53 AGND

52 IOUT2
DVDD 9
AD9854 51 IOUT2B
DVDD 10
TOP VIEW
(Not to Scale) 50 AVDD
DGND 11
80-LEAD LQFP 14
14
1.4 49 IOUT1B
DGND 12
48 IOUT1
NC 13
47 AGND
A5 14
46 AGND
A4 15
A3 16 45 AGND

A2/IO RESET 17 44 AVDD

A1/SDO 18 43 VINN

42 VINP
A0/SDIO 19
41 AGND
I/O UD CLK 20

21 22 23 24 25 26 27 28 29 30 31 32 33 34 35 36 37 38 39 40
WRB/SCLK

RDB/CSB

DVDD
DVDD

DVDD
DGND

DGND
DGND

AVDD

AVDD

AGND

AGND

NC

VOUT

AVDD

AVDD

AGND

AGND
FSK/BPSK/HOLD
SHAPED KEYING

NC = NO CONNECT

REV. B –7–
AD9854
DVDD

AVDD

AVDD DIGITAL
AVDD IN

IOUT IOUTB COMPARATOR

OUT VINP/ AVOID OVERDRIVING
MUST TERMINATE OUTPUTS VINN DIGITAL INPUTS. FORWARD
FOR CURRENT FLOW. DO BIASING ESD DIODES MAY
NOT EXCEED THE OUTPUT COUPLE DIGITAL NOISE
VOLTAGE COMPLIANCE RATING. ONTO POWER PINS.

a. DAC Outputs b. Comparator Output c. Comparator Input d. Digital Input

Figure 1. Equivalent Input and Output Circuits

Typical Performance Characteristics

TPCs 1–6 indicate the wideband harmonic distortion performance of the AD9854 from 19.1 MHz to 119.1 MHz Fundamental
Output, Reference Clock = 30 MHz, REFCLK Multiplier = 10. Each graph plotted from 0 MHz to 150 MHz (Nyquist).

0 0

–10 –10

–20 –20

–30 –30

–40 –40

–50 –50

–60 –60

–70 –70

–80 –80

–90 –90

–100 –100
START 0Hz 15MHz/ STOP 150MHz START 0Hz 15MHz/ STOP 150MHz

TPC 1. Wideband SFDR, 19.1 MHz TPC 3. Wideband SFDR, 59.1 MHz

0 0

–10 –10

–20 –20

–30 –30

–40 –40

–50 –50

–60 –60

–70 –70

–80 –80

–90 –90

–100 –100
START 0Hz 15MHz/ STOP 150MHz START 0Hz 15MHz/ STOP 150MHz

TPC 2. Wideband SFDR, 39.1 MHz TPC 4. Wideband SFDR, 79.1 MHz

–8– REV. B
 Typical Performance Characteristics–AD9854
0 0

–10 –10

–20 –20

–30 –30

–40 –40

–50 –50

–60 –60

–70 –70

–80 –80

–90 –90

–100 –100
START 0Hz 15MHz/ STOP 150MHz CENTER 39.1MHz 5kHz/ SPAN 50kHz

TPC 5. Wideband SFDR, 99.1 MHz TPC 8. Narrowband SFDR, 39.1 MHz, 50 kHz BW,
300 MHz REFCLK with REFCLK Multiplier Bypassed
TPCs 7–10 show the trade-off in elevated noise floor, increased phase noise, and discrete spurious energy when the internal REFCLK
Multiplier circuit is engaged. Plots with wide (1 MHz) and narrow (50 kHz) spans are shown.

0 0

–10 –10

–20 –20

–30 –30

–40 –40

–50 –50

–60 –60

–70 –70

–80 –80

–90 –90

–100 –100
START 0Hz 15MHz/ STOP 150MHz CENTER 39.1MHz 100kHz/ SPAN 1MHz

TPC 6. Wideband SFDR, 119.1 MHz TPC 9. Narrowband SFDR, 39.1 MHz, 1 MHz BW,
30 MHz REFCLK with REFCLK Multiply = 10 ×

0 0

–10 –10

–20 –20

–30 –30

–40 –40

–50 –50

–60 –60

–70 –70

–80 –80

–90 –90

–100 –100
CENTER 39.1MHz 100kHz/ SPAN 1MHz CENTER 39.1MHz 5kHz/ SPAN 50kHz

TPC 7. Narrowband SFDR, 39.1 MHz, 1 MHz BW, TPC 10. Narrowband SFDR, 39.1 MHz, 50 kHz BW,
300 MHz REFCLK with REFCLK Multiply Bypassed 30 MHz REFCLK with REFCLK Multiplier = 10 ×

REV. B –9–
AD9854
Compare the noise floor of TPCs 8 and 10 to TPCs 11 and 12. The improvement seen in TPCs 8 and 10 is a direct result of sam-
pling the fundamental at a higher rate. Sampling at a higher rate spreads the quantization noise of the DAC over a wider bandwidth,
which effectively lowers the noise floor.
0 0

–10 –10

–20 –20

–30 –30

–40 –40

–50 –50

–60 –60

–70 –70

–80 –80

–90 –90

–100 –100
CENTER 39.1MHz 5kHz/ SPAN 50kHz CENTER 39.1MHz 5kHz/ SPAN 50kHz

TPC 11. Narrowband SFDR, 39.1 MHz, 50 kHz BW, TPC 14. Narrowband SFDR, 39.1 MHz, 50 kHz BW,
100 MHz REFCLK with REFCLK Multiplier Bypassed 20 MHz REFCLK with REFCLK Multiplier = 10 ×

TPCs 13 and 14 show the narrowband performance of the AD9854 when operating with a 20 MHz reference clock and the REFCLK
Multiplier enabled at 10× vs. a 200 MHz reference clock with REFCLK Multiplier bypassed.
0 –100

–10
–110
–20
–120
PHASE NOISE – dBc/Hz

–30
AOUT = 80MHz
–40 –130

–50
–140
–60

–70 –150

–80
–160 AOUT = 5MHz
–90

–100 –170
CENTER 39.1MHz 5kHz/ SPAN 50kHz 1.0E + 01 1.0E + 02 1.0E + 03 1.0E + 04 1.0E + 05 1.0E + 06
FREQUENCY – Hz

TPC 12. Narrowband SFDR, 39.1 MHz, 50 kHz BW, TPC 15a. Residual Phase Noise, 300 MHz REFCLK with
10 MHz REFCLK with REFCLK Multiplier = 10 × REFCLK Multiplier Bypassed

0 –90

–10
–100
–20

–30 –110 AOUT = 80MHz

PHASE NOISE – dBc/Hz

–40
–120
–50
–130
–60

–70 –140

–80
–150
–90 AOUT = 5MHz

–100 –160
CENTER 39.1MHz 5kHz/ SPAN 50kHz 1.0E + 01 1.0E + 02 1.0E + 03 1.0E + 04 1.0E + 05 1.0E + 06
FREQUENCY – Hz

TPC 13. Narrowband SFDR, 39.1 MHz, 50 kHz BW, TPC 15b. Residual Phase Noise, 30 MHz REFCLK with
200 MHz REFCLK with REFCLK Multiplier Bypassed REFCLK Multiplier = 10 ×

–10– REV. B
 AD9854
55

54 REF1 RISE
1.174ns
53
C1 FALL
1.286ns
SFDR – dBc

52

51

50

49

48
0 5 10 15 20 25 CH1 500mV M 500ps CH1 980mV
DAC CURRENT – mA

TPC 16. SFDR vs. DAC Current, 59.1 AOUT, 300 MHz TPC 19. Comparator Rise/Fall Times
REFCLK with REFCLK Multiplier Bypassed

620 1200
MINIMUM COMPARATOR
INPUT DRIVE
VCM = 0.5V
615 1000
SUPPLY CURRENT – mA

610 AMPLITUDE – mV p-p 800

605 600

600 400

595 200

590 0
0 20 40 60 80 100 120 140 0 100 200 300 400 500
FREQUENCY – MHz FREQUENCY – MHz

TPC 17. Supply Current vs. Output Frequency; Variation TPC 20. Comparator Toggle Voltage Requirement
Is Minimal as a Percentage and Heavily Dependent on
Tuning Word

RISE TIME
1.04ns

JITTER
[10.6ps RMS]

–33ps 0ps +33ps

500ps/DIV 232mV/DIV 50 INPUT

TPC 18. Typical Comparator Output Jitter, 40 MHz AOUT,

300 MHz REFCLK with REFCLK Multiplier Bypassed

REV. B –11–
AD9854
TYPICAL APPLICATIONS

LPF I BASEBAND I BASEBAND

COS COS
LPF CHANNEL LPF
RF/IF AD9854
INPUT AD9854 SELECT RF OUTPUT
REFCLK LPF FILTERS REFCLK LPF
SIN SIN

LPF Q BASEBAND Q BASEBAND

a. Quadrature Downconversion b. Direct Conversion Quadrature Upconverter

Figure 2. Quadrature Up/Down Conversion Applications

I 8
I/Q MIXER
Rx DUAL DIGITAL Rx BASEBAND
AND
RF IN 8-/10-BIT DEMODULATOR DIGITAL DATA
LOW-PASS Q 8
ADC OUT
FILTER
VCA AGC
ADC CLOCK FREQUENCY
LOCKED TO Tx CHIP/ ADC ENCODE
SYMBOL/PN RATE

AD9854
48
CLOCK
REFERENCE GENERATOR CHIP/SYMBOL/PN
CLOCK RATE DATA

Figure 3. Chip Rate Generator in Spread Spectrum Application

BANDPASS
FILTER AMPLIFIER

IOUT
AD9854
50 50
REFERENCE
RF
CLOCK
FREQUENCY
OUT
PHASE LOOP
COMPARATOR VCO
FILTER

AD9854 FINAL OUTPUT

SPECTRUM SPECTRUM FILTER

FUNDAMENTAL
REF CLK IN
FC – FO FC + FO FC + FO AD9854
IMAGE IMAGE IMAGE DAC OUT DDS
PROGRAMMABLE
"DIVIDE-BY-N" FUNCTION
FCLK BANDPASS (WHERE N = 248/TUNING WORD)
FILTER
TUNING
WORD

Figure 4. Using an Aliased Image to Generate a High Figure 5. Programmable “Fractional Divide-by-N”
Frequency Synthesizer

–12– REV. B
 AD9854
REF
CLOCK RF AD8346 QUADRATURE
FREQUENCY 36dB MODULATOR
AD9854 FILTER OUT TYPICAL
DDS PHASE LOOP SSB COSINE (DC TO 70MHz)
COMPARATOR VCO
FILTER REJECTION LO
50
90
VOUT
AD9854
TUNING
WORD DIVIDE-BY-N
 PHASE
SPLITTER
0.8 TO
2.5GHz QUADRATURE
DDS
0
Figure 6a. Agile High-Frequency Synthesizer LO
SINE (DC TO 70MHz)

DDS – LO LO DDS
+ LO
NOTES
FLIP DDS QUADRATURE SIGNALS TO SELECT ALTERNATE SIDEBAND. ADJUST DDS
SINE OR COSINE SIGNAL AMPLITUDE FOR GREATEST SIDEBAND SUPPRESSION.
DDS DAC OUTPUTS MUST BE LOW-PASS FILTERED PRIOR TO USE WITH THE AD8346.
(REFER TO THE TECHNICAL NOTE AT WEBSITE [WWW.ANALOG.COM/DDS])

Figure 6b. Single-Sideband Upconversion

DIFFERENTIAL
TRANSFORMER-COUPLED
OUTPUT
REFERENCE IOUT
CLOCK FILTER
DDS 50
AD9854 IOUT

50
1:1 TRANSFORMER
I.E, MINI-CIRCUITS T1–1T

Figure 7a. Differential Output Connection for Reduction of Common-Mode Signals

COMPARATORS

AOUT = 100MHz
REFERENCE SIN
CLOCK LPF

AD9854
LPF CLOCK OUT = 200MHz
COS

Figure 7b. Clock Frequency Doubler

AD9854 LOW-PASS
FILTER
8-BIT PARALLEL OR “I” DAC
PROCESSOR/ NOTES
SERIAL PROGRAMMING 1 IOUT = APPROX 20mA MAX WHEN RSET = 2k
CONTROLLER DATA AND CONTROL LOW-PASS
FPGA, ETC. SIGNALS FILTER SWITCH POSTION 1 PROVIDES COMPLEMENTARY
“Q” DAC OR 2 SINUSOIDAL SIGNALS TO THE COMPARATOR
“CONTROL TO PRODUCE A FIXED 50% DUTY CYCLE FROM THE
300MHz MAX DIRECT DAC” COMPARATOR.
REFERENCE MODE OR 15 TO 75MHz
MAX IN THE 4
–20
+ SWITCH POSTION 2 PROVIDES THE SAME DUTY CYCLE
CLOCK
CLOCK USING QUADRATURE SINUSOIDAL SIGNALS TO THE
MULTIPLIER MODE COMPARATOR OR A DC THRESHOLD VOLTAGE TO
2k ALLOW SETTING OF THE COMPARATOR DUTY CYCLE
RSET (DEPENDS ON THE “Q” DAC’s CONFIGURATION)

CMOS LOGIC “CLOCK” OUT

Figure 8. Frequency Agile Clock Generator Applications for the AD9854

REV. B –13–
AD9854
(continued from page 1) tutorial from Analog Devices called “A Technical Tutorial on
simultaneous quadrature output signals at frequencies up to Digital Signal Synthesis.” This tutorial is available on CD-ROM
150 MHz, which can be digitally tuned at a rate of up to 100 and information on obtaining it can be found at the Analog
million new frequencies per second. The (externally filtered) sine Devices DDS website at www.analog.com/dds. The tutorial
wave output can be converted to a square wave by the internal also provides basic applications information for a variety of
comparator for agile clock generator applications. The device pro- digital synthesis implementations. The DDS background subject
vides two 14-bit phase registers and a single pin for BPSK matter is not covered in this data sheet; the functions and features
operation. For higher order PSK operation, the user may use the of the AD9854 will be individually discussed herein.
I/O Interface for phase changes. The 12-bit I and Q DACs,
coupled with the innovative DDS architecture, provide excel- DESCRIPTION OF AD9854 MODES OF OPERATION
lent wide-band and narrow-band output SFDR. The Q DAC There are five programmable modes of operation of the AD9854.
can also be configured as a user-programmable control DAC Selecting a mode requires that three bits in the Control Register
if the quadrature function is not desired. When configured with (parallel address 1F hex) be programmed as follows in Table I.
the comparator, the 12-bit control DAC facilitates static duty cycle
control in the high-speed clock generator applications. Two Table I. Mode Selection Table
12-bit digital multipliers permit programmable amplitude modula-
tion, shaped on/off keying, and precise amplitude control of the Mode 2 Mode 1 Mode 0 Result
quadrature output. Chirp functionality is also included to 0 0 0 SINGLE-TONE
facilitate wide bandwidth frequency sweeping applications. The 0 0 1 FSK
AD9854’s programmable 4×–20× REFCLK multiplier circuit 0 1 0 RAMPED FSK
generates the 300 MHz system clock internally from a lower fre- 0 1 1 CHIRP
quency external reference clock. This saves the user the expense 1 0 0 BPSK
and difficulty of implementing a 300 MHz system clock source.
Direct 300 MHz clocking is also accommodated with either single- In each mode, engaging certain functions may not be permitted.
ended or differential inputs. Single-pin conventional FSK and Shown in Table II is a listing of some important functions and
the enhanced spectral qualities of “ramped” FSK are supported. their availability for each mode.
The AD9854 uses advanced 0.35 micron CMOS technology Single-Tone (Mode 000)
to provide this high level of functionality on a single 3.3 V supply. This is the default mode when master reset is asserted. It may also
The AD9854 is available in a space-saving 80-lead LQFP be accessed by being user-programmed into the control register.
surface mount package and a thermally enhanced 80-lead LQFP The Phase Accumulator, responsible for generating an output
package. The AD9854 is pin-for-pin compatible with the AD9852 frequency, is presented with a 48-bit value from Frequency Tuning
single-tone synthesizer. It is specified to operate over the extended Word 1 registers whose default values are zero. Default values from
industrial temperature range of –40°C to +85°C. the remaining applicable registers will further define the single-tone
output signal qualities.
OVERVIEW The default values after a master reset configure the device with an
The AD9854 quadrature output digital synthesizer is a highly output signal of 0 Hertz, 0 phase. Upon power-up and reset,
flexible device that will address a wide range of applications. the output from both I and Q DACs will be a dc value equal to the
The device consists of an NCO with 48-bit phase accumulator, midscale output current. This is the default mode amplitude
programmable reference clock multiplier, inverse sinc filters, setting of zero. Refer to the digital multiplier section for further
digital multipliers, two 12-bit/300 MHz DACs, high-speed explanation of the output amplitude control. It will be neces-
analog comparator, and interface logic. This highly integrated sary to program all or some of the 28 program registers to realize a
device can be configured to serve as a synthesized LO, agile clock user-defined output signal.
generator, and FSK/BPSK modulator. The theory of operation of
the functional blocks of the device, and a technical description Figure 9 graphically shows the transition from the default condi-
of the signal flow through a DDS device, can be found in a tion (0 Hz) to a user-defined output frequency (F1).
FREQUENCY

F1

MODE 000 (DEFAULT) 000 (SINGLE TONE)

TW1 0 F1

MASTER RESET

I/O UPDATE
CLOCK

Figure 9. Default State to User-Defined Output Transition

–14– REV. B
 AD9854
Table II. Function Availability vs. Mode of Operation

Single-Pin Single-Pin Phase Amplitude Inverse Frequency Frequency Automatic

Phase Phase FSK/BPSK Shaped- Offset or Control or SINC Tuning Tuning Frequency
Mode Adjust 1 Adjust 2 or HOLD Keying Modulation Modulation Filter Word 1 Word 2 Sweep
Single-Tone ✓ X X ✓ ✓ ✓ ✓ ✓ X X
FSK ✓ X ✓ ✓ ✓ ✓ ✓ ✓ ✓ X
Ramped FSK ✓ X ✓ ✓ ✓ ✓ ✓ ✓ ✓ ✓
CHIRP ✓ X ✓ ✓ ✓ ✓ ✓ ✓ X ✓
BPSK ✓ ✓ ✓ ✓ X ✓ ✓ ✓ X X

As with all Analog Devices DDSs, the value of the frequency Furthermore, all of these qualities can be changed or modulated
tuning word is determined using the following equation: via the 8-bit parallel programming port at a 100 MHz parallel-byte
FTW = (Desired Output Frequency × 2N)/SYSCLK. rate, or at a 10 MHz serial rate. Incorporating this attribute will
permit FM, AM, PM, FSK, PSK, ASK operation in the single-
Where N is the phase accumulator resolution (48 bits in this tone mode.
instance), frequency is expressed in Hertz, and the FTW, Fre-
quency Tuning Word, is a decimal number. Once a decimal Unramped FSK (Mode 001)
number has been calculated, it must be rounded to an integer When selected, the output frequency of the DDS is a function
and then converted to binary format—a series of 48 binary- of the values loaded into Frequency Tuning Word registers 1
weighted 1s or 0s. The fundamental sine wave DAC output and 2 and the logic level of Pin 29 (FSK/BPSK/HOLD). A logic
frequency range is from dc to 1/2 SYSCLK. low on Pin 29 chooses F1 (frequency tuning word 1, parallel
address 4–9 hex) and a logic high chooses F2 (frequency tuning
Changes in frequency are phase-continuous, which means that the word 2, parallel register address A–F hex). Changes in frequency
first sampled phase value of the new frequency will be referenced in are phase-continuous and are internally coincident with the
time from the last sampled phase value of the previous frequency. FSK data pin (29); however, there is deterministic pipeline delay
The I and Q DACs of the AD9854 are always 90 degrees out of between the FSK data signal and the DAC output. (Please refer
phase. The 14-bit phase registers (discussed elsewhere in this to pipeline delays in specification table.)
data sheet) do not independently adjust the phase of each DAC The unramped FSK mode, Figure 10, is representative of
output. Instead, both DACs are affected equally by a change in traditional FSK, RTTY (Radio Teletype) or TTY (Teletype)
phase offset. transmission of digital data. FSK is a very reliable means of
The single-tone mode allows the user to control the following digital communication; however, it makes inefficient use of the
signal qualities: bandwidth in the RF Spectrum. Ramped FSK in Figure 11 is a
method of conserving the bandwidth.
• Output Frequency to 48-Bit Accuracy
• Output Amplitude to 12-Bit Accuracy Ramped FSK (Mode 010)
– Fixed, User-Defined, Amplitude Control This mode is a method of FSK whereby changes from F1
– Variable, Programmable Amplitude Control to F2 are not instantaneous but, instead, are accomplished
– Automatic, Programmable, Single-Pin-Controlled, “Shaped in a frequency sweep or “ramped” fashion, the “ramped” no-
On/Off Keying” tation implies that the sweep is linear. While linear sweeping
• Output Phase to 14-Bit Accuracy or frequency ramping is easily and automatically accomplished,
it is only one of many possibilities. Other frequency transition

F2
FREQUENCY

F1

MODE 000 (DEFAULT) 001 (FSK NO RAMP)

TW1 0 F1

TW2 0 F2

I/O UPDATE CLK

FSK DATA (PIN 29)

Figure 10. Traditional FSK Mode

REV. B –15–
AD9854
F2

FREQUENCY
F1

MODE 000 (DEFAULT) 010 (RAMPED FSK)

TW1 0 F1

TW2 0 F2

DFW REQUIRES A POSITIVE TWO’S COMPLEMENT VALUE

RAMP RATE

I/O UPDATE CLK

FSK DATA (PIN 29)

Figure 11. Ramped FSK Mode

F2
FREQUENCY

F1

MODE 000 (DEFAULT) 010 (RAMPED FSK)

TW1 0 F1

TW2 0 F2

I/O UPDATE
CLOCK

FSK DATA

Figure 12. Ramped FSK Mode

schemes may be implemented by changing the ramp rate and user controls the dwell time at F1 and F2, the number of inter-
ramp step size “on-the-fly,” in piecewise fashion. mediate frequencies and time spent at each frequency. Unlike
Frequency ramping, whether linear or nonlinear, necessitates unramped FSK, ramped FSK requires the lowest frequency to be
that many intermediate frequencies between F1 and F2 will be loaded into F1 registers and the highest frequency into F2 registers.
output in addition to the primary F1 and F2 frequencies. Figures Several registers must be programmed to instruct the DDS
11 and 12 graphically depict the frequency versus time charac- regarding the resolution of intermediate frequency steps (48
teristics of a linear ramped FSK signal. bits) and the time spent at each step (20 bits). Furthermore, the
NOTE: In ramped FSK mode, the Delta Frequency (DFW) is CLR ACC1 bit in the control register should be toggled (low-high-
required to be programmed as a positive two’s complement value. low) prior to operation to assure that the frequency accumulator
Another requirement is that the lowest frequency (F1) be is starting from an “all zeros” output condition. For piecewise,
programmed in the Frequency Tuning Word 1 register. nonlinear frequency transitions, it is necessary to reprogram the
registers while the frequency transition is in progress to affect the
The purpose of ramped FSK is to provide better bandwidth desired response.
containment than traditional FSK by replacing the instantaneous
frequency changes with more gradual, user-defined frequency Parallel register addresses 1A–1C hex comprise the 20-bit “Ramp
changes. The dwell time at F1 and F2 can be equal to or much Rate Clock” registers. This is a countdown counter that outputs
greater than the time spent at each intermediate frequency. The a single pulse whenever the count reaches zero. The counter
is activated any time a logic level change occurs on FSK input

–16– REV. B
 AD9854
Pin 29. This counter is run at the System Clock Rate, 300 MHz frequency is ramped up and down in frequency, according to
maximum. The time period between each output pulse is given as the logic-state of Pin 29. The rate at which this happens is a
(N+1) × (SYSTEM CLOCK PERIOD) function of the 20-bit ramp rate clock. Once the destination
frequency is achieved, the ramp rate clock is stopped, which halts
where N is the 20-bit ramp rate clock value programmed by the the frequency accumulation process.
user. The allowable range of N is from 1 to (220 –1). The output
of this counter clocks the 48-bit Frequency Accumulator Generally speaking, the Delta Frequency Word will be a much
shown below in Figure 13. The Ramp Rate Clock determines the smaller value compared to that of the F1 or F2 tuning word.
amount of time spent at each intermediate frequency between F1 For example, if F1 and F2 are 1 kHz apart at 13 MHz, the
and F2. The counter stops automatically when the destination Delta Frequency Word might be only 25 Hz.
frequency is achieved. The “dwell time” spent at F1 and F2 is
determined by the duration that the FSK input, Pin 29, is held
high or low after the destination frequency has been reached. F2

FREQUENCY
PHASE
F1
ADDER ACCUMULATOR

FREQUENCY 0
ACCUMULATOR
MODE 010 (RAMPED FSK)
48-BIT DELTA- INSTANTANEOUS
FREQUENCY
FSK (PIN 29) PHASE OUT
WORD (TWO’S TW1 F1
COMPLEMENT)

TW2 F2
FREQUENCY FREQUENCY
TUNING TUNING FSK DATA
WORD 1 WORD 2
TRIANGLE
BIT

20-BIT I/O UPDATE

SYSTEM CLOCK
RAMP RATE
CLOCK
CLOCK
Figure 14. Effect of Triangle Bit in Ramped FSK Mode
Figure 13. Block Diagram of Ramped FSK Function Figure 15 shows that premature toggling causes the ramp to
Parallel register addresses 10–15 hex comprise the 48-bit, two’s immediately reverse itself and proceed at the same rate and resolu-
complement, “Delta Frequency Word” registers. This 48-bit tion back to originating frequency.
word is accumulated (added to the accumulator’s output) every The control register contains a triangle bit at parallel register
time it receives a clock pulse from the ramp rate counter. The address 1F hex. Setting this bit high in Mode 010 causes an
output of this accumulator is then added to or subtracted from automatic ramp-up and ramp-down between F1 and F2 to
the F1 or F2 frequency word, which is then fed to the input of the occur without having to toggle Pin 29 as shown in Figure 14. In
48-bit Phase Accumulator that forms the numerical phase steps fact, the logic state of Pin 29 has no effect once the triangle bit
for the sine and cosine wave outputs. In this fashion, the output is set high. This function uses the ramp-rate clock time period

F2
FREQUENCY

F1

MODE 000 (DEFAULT) 010 (RAMPED FSK)

TW1 0 F1

TW2 0 F2

I/O UPDATE
CLOCK

FSK DATA

Figure 15. Effect of Premature Ramped FSK Data

REV. B –17–
AD9854
and the delta-frequency-word step size to form a continuously changes can be precisely timed using the 32-bit Internal Update
sweeping linear ramp from F1 to F2 and back to F1 with equal Clock (see detailed description of Update Clock in this data sheet).
dwell times at every frequency. Using this function, one can Nonlinear ramped FSK will have the appearance of a chirp
automatically sweep between any two frequencies from dc to Nyquist. function that is graphically illustrated in Figure 39. The major
In the ramped FSK mode, with the triangle bit set high, an difference between a ramped FSK function and a chirp function
automatic frequency sweep will begin at either F1 or F2, is that FSK is limited to operation between F1 and F2. Chirp
according to the logic level on Pin 29 (FSK input pin) when the operation has no F2 limit frequency.
triangle bit’s rising edge occurs as shown in Figure 16. If the Two additional control bits are available in the ramped FSK mode
FSK data bit had been high instead of low, F2, rather than F1, that allow even more options. CLR ACC1, register address 1F hex,
would have been chosen as the start frequency. will, if set high, clear the 48-bit frequency accumulator (ACC1)
Additional flexibility in the ramped FSK mode is provided in output with a retriggerable one-shot pulse of one system clock
the ability to respond to changes in the 48-bit delta frequency duration. If the CLR ACC1 bit is left high, a one-shot pulse will
word and/or the 20-bit ramp-rate counter on-the-fly during the be delivered on the rising edge of every Update Clock. The effect
is to interrupt the current ramp, reset the frequency back to the
start point, F1 or F2, and then continue to ramp up (or down)
F2
at the previous rate. This will occur even when a static F1 or F2
FREQUENCY

destination frequency has been achieved.

Next, CLR ACC2 control bit (register address 1F hex) is avail-
able to clear both the frequency accumulator (ACC1) and the phase
F1
accumulator (ACC2). When this bit is set high, the output of the
0 phase accumulator will result in 0 Hz output from the DDS. As
long as this bit is set high, the frequency and phase accumulators
MODE 000 (DEFAULT) 010 (RAMPED FSK)
will be cleared, resulting in 0 Hz output. To return to previous
DDS operation, CLR ACC2 must be set to logic low.
TW1 0 F1
Chirp (Mode 011)
TW2 0 F2 This mode is also known as pulsed FM. Most chirp systems use
a linear FM sweep pattern, but the AD9854 supports nonlinear
FSK DATA
patterns, as well. In radar applications, use of chirp or pulsed
FM allows operators to significantly reduce the output power
TRIANGLE BIT
needed to achieve the same result as a single-frequency radar
Figure 16. Automatic Linear Ramping Using the system would produce. Figure 17 represents a very low-resolution
Triangle Bit nonlinear chirp meant to demonstrate the different “slopes” that
are created by varying the time steps (ramp rate) and frequency
ramping from F1 to F2 or vice versa. To create these nonlinear
steps (delta frequency word).
frequency changes it is necessary to combine several linear ramps,
in a piecewise fashion, with differing slopes. This is done by The AD9854 permits precise, internally generated linear or exter-
programming and executing a linear ramp at some rate or “slope” nally programmed nonlinear pulsed or continuous FM over the
and then altering the slope (by changing the ramp rate clock or complete frequency range, duration, frequency resolution,
delta frequency word or both). Changes in slope are made as often and sweep direction(s). These are all user programmable. A block
as needed to form the desired nonlinear frequency sweep response diagram of the FM chirp components is shown in Figure 18.
before the destination frequency has been reached. These piecewise
FREQUENCY

F1

MODE 000 (DEFAULT) 010 (RAMPED FSK)

TW1 0 F1

DFW

RAMP RATE

I/O UPDATE
CLOCK

Figure 17. Example of a Nonlinear Chirp

–18– REV. B
 AD9854
clock). Instant return to FTW1 is easily achieved, though, and this
option is explained in the next few paragraphs.
OUT
ADDER PHASE Two control bits are available in the FM chirp mode that will
ACCUMULATOR
allow the return to the beginning frequency, FTW1, or to 0 Hz.
FREQUENCY First, when the CLR ACC1 bit (register address 1F hex) is set
ACCUMULATOR CLR ACC2 high, the 48-bit frequency accumulator (ACC1) output is cleared with
48-BIT DELTA- a retriggerable one-shot pulse of one system clock duration.
FREQUENCY
WORD (TWO’S CLR ACC1 The 48-bit delta frequency word input to the accumulator is
COMPLEMENT) unaffected by CLR ACC1 bit. If the CLR ACC1 bit is held high, a
FREQUENCY
TUNING one-shot pulse will be delivered to the Frequency Accumulator
WORD 1
(ACC1) on every rising edge of the I/O Update Clock. The effect
20-BIT
SYSTEM
is to interrupt the current chirp, reset the frequency back to FTW1,
HOLD RAMP RATE
CLOCK
CLOCK and continue the chirp at the previously programmed rate and
direction. Clearing the output of the Frequency Accumulator
Figure 18. FM Chirp Components in the chirp mode is illustrated in Figure 19. Shown in the diagram
is the I/O Update Clock, which is either user-supplied or internally
Basic FM Chirp Programming Steps
generated. A discussion of I/O Update is presented elsewhere
1. Program a start frequency into Frequency Tuning Word 1
in this data sheet.
(parallel register addresses 4–9 hex), hereafter called FTW1.
Next, CLR ACC2 control bit (register address 1F hex) is available
2. Program the frequency step resolution into the 48-bit, two’s
to clear both the frequency accumulator (ACC1) and the phase
complement, Delta Frequency Word (parallel register addresses
accumulator (ACC2). When this bit is set high, the output of the
10–15 hex).
phase accumulator will result in 0 Hz output from the DDS. As
3. Program the rate of change (time at each frequency) into the 20- long as this bit is set high, the frequency and phase accumulators
bit Ramp Rate Clock (parallel register addresses 1A–1C hex). will be cleared, resulting in 0 Hz output. To return to previous
4. When programming is complete, an I/O update pulse at Pin DDS operation, CLR ACC2 must be set to logic low. This bit is
20 will engage the program commands. useful in generating pulsed FM.
The necessity for a two’s complement Delta Frequency Word is Figure 20 graphically illustrates the effect of CLR ACC2 bit upon
to define the direction in which the FM chirp will move. If the the DDS output frequency. Note that reprogramming the registers
48-bit delta frequency word is negative (MSB is high), then the while the CLR ACC2 bit is high allows a new FTW1 frequency
incremental frequency changes will be in a negative direction and slope to be loaded.
from FTW1. If the 48-bit word is positive (MSB is low), then Another function that is available only in the chirp mode is the
the incremental frequency changes will be in a positive direction. HOLD pin, Pin 29. This function will stop the clock signal to the
It is important to note that FTW1 is only a starting point for ramp rate counter, thereby halting any further clocking pulses to
FM chirp. There is no built-in restraint requiring a return to the frequency accumulator, ACC1. The effect is to halt the
FTW1. Once the FM chirp has begun, it is free to move (under chirp at the frequency existing just before HOLD was pulled
program control) within the Nyquist bandwidth (dc to 1/2 system high. When the HOLD pin is returned low, the clocks are resumed
and chirp continues. During a hold condition, the user may
change the programming registers; however, the ramp rate counter
FREQUENCY

F1

MODE 000 (DEFAULT) 011 (CHIRP)

FTW1 0 F1

DFW DELTA FREQUENCY WORD

RAMP RATE RAMP RATE

I/O UPDATE
CLOCK

CLR ACC1

Figure 19. Effect of CLR ACC1 in FM Chirp Mode

REV. B –19–
AD9854

FREQUENCY
F1

MODE 000 (DEFAULT) 011 (CHIRP)

TW1 0

DPW

RAMP RATE

CLR ACC2

I/O UPDATE
CLOCK

Figure 20. Effect of CLR ACC2 in FM Chirp Mode

FREQUENCY

F1

MODE 000 (DEFAULT) 011 (CHIRP)

TW1 0 F1

DFW DELTA FREQUENCY WORD

RAMP RATE RAMP RATE

HOLD

I/O UPDATE
CLOCK

Figure 21. Illustration of HOLD Function

must resume operation at its previous rate until a count of zero is When the chirp destination frequency is reached, there are sev-
obtained before a new ramp rate count can be loaded. Figure 21 eral possible outcomes:
illustrates the effect of the hold function on the DDS output 1. Stop at the destination frequency using the HOLD pin, or by
frequency. loading all zeros into the Delta Frequency Word registers of
The 32-bit automatic I/O Update counter may be used to con- the frequency accumulator (ACC1).
struct complex chirp or ramped FSK sequences. Since this internal 2. Use the HOLD pin function to stop the chirp, then ramp down
counter is synchronized with the AD9854 System Clock, it allows the output amplitude using the digital multiplier stages and
precisely timed program changes to be invoked. In this manner, the Shaped Keying pin, Pin 30, or via program register control
the user is only required to reprogram the desired registers before (addresses 21–24 hex).
the automatic I/O Update Clock is generated.
3. Abruptly terminate the transmission using the CLR ACC2 bit.
In the chirp mode, the destination frequency is not directly
specified. If the user fails to control the chirp, the DDS will natu- 4. Continue chirp by reversing direction and returning to the
rally confine itself to the frequency range between dc and Nyquist. previous, or another, destination frequency in a linear or user-
Unless terminated by the user, the chirp will continue until power directed manner. If this involves going down in frequency, a
is removed. negative 48-bit delta frequency word (the MSB is set to “1”)
must be loaded into registers 10–15 hex. Any decreasing

–20– REV. B
 AD9854
360

PHASE
0

MODE 000 (DEFAULT) 100 (BPSK)

FTW1 0 F1

PHASE ADJUST 1 270 DEGREES

PHASE ADJUST 2 90 DEGREES

BPSK DATA

I/O UPDATE
CLOCK

Figure 22. BPSK Mode

frequency step of the Delta Frequency Word requires the When the user provides an external Update Clock, it is internally
MSB to be set to logic high. synchronized with the system clock to prevent partial transfer
5. Continue chirp by immediately returning to the beginning of program register information due to violation of data setup
frequency (F1) in a sawtooth fashion and repeat the previ- or hold times. This mode gives the user complete control of
ous chirp process. This is where CLR ACC1 control bit is used. when updated program information becomes effective. The
An automatic, repeating chirp can be set up using the 32-bit default mode for Update Clock is internal (Int Update Clk control
Update Clock to issue CLR ACC1 command at precise time register bit is logic high). To switch to External Update Clock
intervals. Adjusting the timing intervals or changing the Delta mode, the Int Update Clk register bit must be set to logic low.
Frequency Word will change the chirp range. It is incumbent The internal update mode generates automatic, periodic update
upon the user to balance the chirp duration and frequency pulses with the time period set by the user.
resolution to achieve the proper frequency range. An internally generated Update Clock can be established by
BPSK (Mode 100) programming the 32-bit Update Clock registers (address 16–19
Binary, biphase, or bipolar phase shift keying is a means to rapidly hex) and setting the Int Update Clk (address 1F hex) control
select between two preprogrammed 14-bit output phase offsets register bit to logic high. The update clock down-counter function
that will identically affect both the I and Q outputs of the AD9854. operates at 1/2 the rate of the system clock (150 MHz maximum)
The logic state of Pin 29, BPSK pin, controls the selection of and counts down from a 32-bit binary value (programmed by
Phase Adjust register number 1 or 2. When low, Pin 29 selects the user). When the count reaches 0, an automatic I/O Update of
Phase Adjust register 1; when high, Phase Adjust register 2 is the DDS output or functions is generated. The Update Clock is
selected. Figure 22 illustrates phase changes made to four cycles internally and externally routed on Pin 20 to allow users to
of an output carrier. synchronize programming of update information with the update
clock rate. The time period between update pulses is given as:
Basic BPSK programming steps:
1. Program a carrier frequency into Frequency Tuning Word 1. (N+1) × (SYSTEM CLOCK PERIOD × 2)

2. Program appropriate 14-bit phase words in Phase Adjust where N is the 32-bit value programmed by the user. The
registers 1 and 2. allowable range of N is from 1 to (232 –1). The internally gen-
erated update pulse output on Pin 20 has a fixed high time of eight
3. Attach BPSK data source to Pin 29. system clock cycles.
4. Activate I/O Update Clock when ready. Programming the Update Clock register for values less than five
NOTE: If higher order PSK modulation is desired, the user should will cause the I/O UD pin to remain high. The Update Clock
select the single-tone mode and program Phase Adjust register 1 functionality still works; however, the user cannot use the signal
using the serial or high-speed parallel programming bus. as an indication that data is transferring. This is an effect of the
minimum high pulse time when I/O UD is an output.
USING THE AD9854 Shaped On/Off Keying
Internal and External Update Clock This feature allows the user to control the amplitude vs. time slope
This function is comprised of a bidirectional I/O pin, Pin 20, and a of the I and Q DAC output signals. This function is used in “burst
programmable 32-bit down-counter. In order for programming transmissions” of digital data to reduce the adverse spectral impact
changes to be transferred from the I/O Buffer registers to the active of short, abrupt bursts of data. Users must first enable the digi-
core of the DDS, a clock signal (low to high edge) must be externally tal multipliers by setting the OSK EN bit (control register address
supplied to Pin 20 or internally generated by the 32-bit Update Clock. 20 hex) to logic high in the control register.

REV. B –21–
AD9854
Otherwise, if the OSK EN bit is set low, the digital multipliers The two fixed elements of the transition time are the period of
responsible for amplitude control are bypassed and the I and Q the system clock (which drives the Ramp Rate Counter) and the
DAC outputs are set to full-scale amplitude. In addition to set- number of amplitude steps (4096). To give an example, assume
ting the OSK EN bit, a second control bit, OSK INT (also at that the system clock of the AD9854 is 100 MHz (10 ns period). If
address 20 hex), must be set to logic high. Logic high selects the the Ramp Rate Counter is programmed for a minimum count of
linear internal control of the output ramp-up or ramp-down three, it will take two system clock periods (one rising edge
function. A logic low in the OSK INT bit switches control of loads the count-down value, the next edge decrements the counter
the digital multipliers to user programmable 12-bit registers from three to two). If the count down value is less than three, the
allowing users to dynamically shape the amplitude transition in Ramp Rate Counter will stall and, therefore, produce a con-
practically any fashion. These 12-bit registers, labeled “Output stant scaling value to the digital multipliers. This stall condition
Shape Key I and Output Shape Key Q,” are located at addresses may have application to the user. The relationship of the 8-bit
21 through 24 hex in Table IV. The maximum output amplitude is count-down value to the time period between output pulses is
a function of the RSET resistor and is not programmable when given as:
OSK INT is enabled. (N+1) × SYSTEM CLOCK PERIOD
ABRUPT ON/OFF KEYING where N is the 8-bit count-down value. It will take 4096 of these
pulses to advance the 12-bit up-counter from zero-scale to full-
ZERO FULL scale. Therefore, the minimum shaped keying ramp time for a
SCALE SCALE
100 MHz system clock is 4096 × 4 × 10 ns = approximately 164 µs.
The maximum ramp time will be 4096 × 256 × 10 ns = approxi-
mately 10.5 ms.
ZERO FULL Finally, changing the logic state of Pin 30, “shaped keying”, will
SCALE SCALE
automatically perform the programmed output envelope functions
when OSK INT is high. A logic high on Pin 30 causes the out-
SHAPED ON/OFF KEYING
puts to linearly ramp up to full-scale amplitude and hold until
Figure 23. Shaped On/Off Keying the logic level is changed to low, causing the outputs to ramp
The transition time from zero-scale to full-scale must also be down to zero-scale.
programmed. The transition time is a function of two fixed I and Q DACs
elements and one variable. The variable element is the program- The sine and cosine outputs of the DDS drive the Q and I DACs,
mable 8-bit Ramp Rate Counter. This is a down-counter that is respectively (300 MSPS maximum). Their maximum output
clocked at the system clock rate (300 MHz max) and generates one amplitudes are set by the DAC RSET resistor at Pin 56. These are
pulse whenever the counter reaches zero. This pulse is routed to current-out DACs with a full-scale maximum output of 20 mA;
a 12-bit counter that increments with each pulse received. The however, a nominal 10 mA output current provides best spurious-
outputs of the 12-bit counter are connected to the 12-bit digital free dynamic range (SFDR) performance. The value of
multiplier. When the digital multiplier has a value of all zeros at RSET = 39.93/IOUT, where IOUT is in amps. DAC output compliance
its inputs, the input signal is multiplied by zero, producing zero- specification limits the maximum voltage developed at the outputs
scale. When the multiplier has a value of all ones, the input signal to –0.5 V to +1 V. Voltages developed beyond this limitation will
is multiplied by a value of 4095/4096, producing nearly full- cause excessive DAC distortion and possibly permanent dam-
scale. There are 4,094 remaining fractional multiplier values age. The user must choose a proper load impedance to limit the
that will produce output amplitudes scaled according to their output voltage swing to the compliance limits. Both DAC out-
binary values. puts should be terminated equally for best SFDR, especially at
higher output frequencies where harmonic distortion errors are
more prominent.

(BYPASS MULTIPLIER)
DIGITAL OSK EN = 0 OSK EN = 0
DDS DIGITAL SIGNAL IN 12 12-BIT DIGITAL 12
OUTPUT MULTIPLIER SINE DAC
OSK EN = 1 OSK EN = 1

12
USER-PROGRAMMABLE
12-BIT Q-CHANNEL OSK INT = 1
MULTIPLIER 12
“OUTPUT SHAPE
KEY Q MULT” OSK INT = 0 12
REGISTER
12-BIT 1 8-BIT RAMP SYSTEM
UP/DOWN RATE CLOCK
COUNTER COUNTER

SHAPED ON/OFF
KEYING PIN

Figure 24. Block Diagram of Q-Pathway of the Digital Multiplier Section Responsible for Shaped-Keying Function

–22– REV. B
 AD9854
Both DACs are preceded by inverse SIN(x)/x filters (a.k.a. inverse over the range of 4× to 20×. Use of this function allows users to
sinc filters) that precompensate for DAC output amplitude varia- input as little as 15 MHz at the REFCLK input to produce a
tions over frequency to achieve flat amplitude response from dc 300 MHz internal system clock. Five bits in control register 1E
to Nyquist. Both DACs can be powered down by setting the hex set the multiplier value as follows in Table III.
DAC PD bit high (address 1D of control register) when not The REFCLK Multiplier function can be bypassed to allow direct
needed. I DAC outputs are designated as IOUT1 and IOUT1B, clocking of the AD9854 from an external clock source. The
Pins 48 and 49 respectively. Q DAC outputs are designated system clock for the AD9854 is either the output of the REFCLK
as IOUT2 and IOUT2B, Pins 52 and 51 respectively. Multiplier (if it is engaged) or the REFCLK inputs. REFCLK
Control DAC may be either a single-ended or differential input by setting Pin
The 12-bit Q DAC can be reconfigured to perform as a “control” 64, DIFF CLK ENABLE, low or high respectively.
or auxiliary DAC. The control DAC output can provide dc PLL Range Bit
control levels to external circuitry, generate ac signals, or enable The PLL Range Bit selects the frequency range of the REFCLK
duty cycle control of the on-board comparator. When the SRC Multiplier PLL. For operation from 200 MHz to 300 MHz
Q DAC bit in the control register (parallel address 1F hex) is (internal system clock rate), the PLL Range Bit should be set to
set high, the Q DAC inputs are switched from internal 12-bit Q Logic 1. For operation below 200 MHz, the PLL Range Bit
data source (default setting) to external 12-bit, two’s complement should be set to Logic 0. The PLL Range Bit adjusts the PLL
data, supplied by the user. Data is channeled through the serial loop parameters for optimized phase noise performance within
or parallel interface to the 12-bit Q DAC register (address 26 and each range.
27 hex) at a maximum 100 MHz data rate. This DAC is clocked
at the system clock, 300 MSPS (maximum), and has the same Pin 61, PLL FILTER
maximum output current capability as that of the I DAC. The This pin provides the connection for the external zero compen-
single RSET resistor on the AD9854 sets the full-scale output sation network of the PLL loop filter. The zero compensation
current for both DACs. The control DAC can be separately network consists of a 1.3 kΩ resistor in series with a 0.01 µF
powered down for power conservation when not needed by set- capacitor. The other side of the network should be connected to
ting the Q DAC POWER-DOWN bit high (address 1D hex). as close as possible to Pin 60, AVDD. For optimum phase noise
Control DAC outputs are designated as IOUT2 and IOUT2B performance, the clock multiplier can be bypassed by setting the
(Pins 52 and 51 respectively). “Bypass PLL” bit in control register address 1E.
4.0 Differential REFCLK Enable
3.5 A high level on this pin enables the differential clock inputs,
3.0 REFCLK and REFCLKB (Pins 69 and 68 respectively). The
2.5
minimum differential signal amplitude required is 400 mV p-p,
2.0 ISF
1.5
at the REFCLK input pins. The centerpoint or common-mode
1.0 range of the differential signal can range from 1.6 V to 1.9 V.
0.5 SYSTEM
When Pin 64 (DIFF CLK ENABLE) is tied low, REFCLK
dB

0
–0.5 (Pin 69) is the only active clock input. This is referred to as
–1.0 the single-ended mode. In this mode, Pin 68 (REFCLKB) should
–1.5 be tied low or high.
–2.0 SINC
–2.5 High-Speed Comparator—Optimized for high speed, >300 MHz
–3.0 toggle rate, low jitter, sensitive input, built-in hysteresis, and an output
–3.5
level of 1 V p-p minimum into 50 Ω or CMOS logic levels into high
–4.0
0 0.1 0.2 0.3 0.4 0.5 impedance loads. The comparator can be separately powered down
FREQUENCY NORMALIZED TO SAMPLE RATE to conserve power. This comparator is used in “clock generator”
Figure 25. Inverse SINC Filter Response applications to square up the filtered sine wave generated by the DDS.
Inverse SINC Function Power-Down—Several individual stages may be powered down
This filter precompensates input data to both DACs for the to reduce power consumption via the programming registers
SIN(x)/x roll-off characteristic inherent in the DAC’s output while still maintaining functionality of desired stages. These
spectrum. This allows wide bandwidth signals (such as QPSK) stages are identified in the Register Layout table, address 1D hex.
to be output from the DACs without appreciable amplitude varia- Power-down is achieved by setting the specified bits to logic high.
tions as a function of frequency. The inverse SINC function may A logic low indicates that the stages are powered up.
be bypassed to significantly reduce power consumption, espe- Furthermore, and perhaps most significantly, the Inverse Sinc
cially at higher clock speeds. When the Q DAC is configured filters and the Digital Multiplier stages, can be bypassed to achieve
as a “control” DAC, the inverse SINC function does not apply significant power reduction through programming of the control
to the Q path. registers in address 20 hex. Again, logic high will cause the stage to
Inverse SINC is engaged by default and is bypassed by bringing be bypassed. Of particular importance is the Inverse Sinc filter;
the “Bypass Inv SINC” bit high in control register 20 (hex) in this stage consumes a significant amount of power.
Table IV. A full power-down occurs when all four PD bits in control register
REFCLK Multiplier 1D hex are set to logic high. This reduces power consumption
This is a programmable PLL-based reference clock multiplier to approximately 10 mW (3 mA).
that allows the user to select an integer clock multiplying value

REV. B –23–
AD9854
Table III. REFCLK Multiplier Control Register Values

Multiplier Value Ref Mult Bit 4 Ref Mult Bit 3 Ref Mult Bit 2 Ref Mult Bit 1 Ref Mult Bit 0
4 0 0 1 0 0
5 0 0 1 0 1
6 0 0 1 1 0
7 0 0 1 1 1
8 0 1 0 0 0
9 0 1 0 0 1
10 0 1 0 1 0
11 0 1 0 1 1
12 0 1 1 0 0
13 0 1 1 0 1
14 0 1 1 1 0
15 0 1 1 1 1
16 1 0 0 0 0
17 1 0 0 0 1
18 1 0 0 1 0
19 1 0 0 1 1
20 1 0 1 0 0

PROGRAMMING THE AD9854 Regardless of mode, the I/O port data is written to a buffer
The AD9854 Register Layout, shown in Table IV, contains memory that does NOT affect operation of the part until the
the information that programs the chip for the desired function- contents of the buffer memory are transferred to the register
ality. While many applications will require very little programming banks. This transfer of information occurs synchronously to the
to configure the AD9854, some will make use of all twelve acces- system clock and occurs in one of two ways:
sible register banks. The AD9854 supports an 8-bit parallel I/O 1. Internally controlled at a rate programmable by the user, or
operation or an SPI-compatible serial I/O operation. All accessible
registers can be written and read back in either I/O operating mode. 2. Externally controlled by the user. I/O operations can occur in
the absence of REFCLK but the data cannot be moved from
S/P SELECT, Pin 70, is used to configure the I/O mode. Systems the buffer memory to the register bank without REFCLK.
that use the parallel I/O mode must connect the S/P SELECT See the Update Clock Operation section of this document
pin to VDD. Systems that operate in the serial I/O mode must tie for details.
the S/P SELECT pin to GND.

A<5:0> A1 A2 A3

D<7:0> D1 D2 D3

RD

TRDHOZ TRDLOV

TAHD TADV

SPECIFICATION VALUE DESCRIPTION

TADV 15ns ADDRESS TO DATA VALID TIME (MAXIMUM)
TAHD 5ns ADDRESS HOLD TIME TO RD SIGNAL INACTIVE (MINIMUM)
TRDLOV 15ns RD LOW TO OUTPUT VALID (MAXIMUM)
TRDHOZ 10ns RD HIGH TO DATA THREE-STATE (MAXIMUM)

Figure 26. Parallel Port Read Timing Diagram

–24– REV. B
 AD9854
Table IV. Register Layout

Parallel Serial
Address Address AD9854 Register Layout

Default
Hex Hex Bit 7 Bit 6 Bit 5 Bit 4 Bit 3 Bit 2 Bit 1 Bit 0 Value

00 0 Phase Adjust Register #1 <13:8> (Bits 15, 14 don’t care) Phase 1 00h
01 Phase Adjust Register #1 <7:0> 00h

02 1 Phase Adjust Register #2 <13:8:> (Bits 15, 14 don’t care) Phase 2 00h
03 Phase Adjust Register #2 <7:0> 00h

04 2 Frequency Tuning Word 1 <47:40> Frequency 1 00h

05 Frequency Tuning Word 1 <39:32> 00h
06 Frequency Tuning Word 1 <31:24> 00h
07 Frequency Tuning Word 1 <23:16> 00h
08 Frequency Tuning Word 1 <15:8> 00h
09 Frequency Tuning Word 1 <7:0> 00h

0A 3 Frequency Tuning Word 2 <47:40> Frequency 2 00h

0B Frequency Tuning Word 2 <39:32> 00h
0C Frequency Tuning Word 2 <31:24> 00h
0D Frequency Tuning Word 2 <23:16> 00h
0E Frequency Tuning Word 2 <15:8> 00h
0F Frequency Tuning Word 2 <7:0> 00h

10 4 Delta Frequency Word <47:40> 00h

11 Delta Frequency Word <39:32> 00h
12 Delta Frequency Word <31:24> 00h
13 Delta Frequency Word <23:16> 00h
14 Delta Frequency Word <15:8> 00h
15 Delta Frequency Word <7:0> 00h

16 5 Update Clock <31:24> 00h

17 Update Clock <23:16> 00h
18 Update Clock <15:8> 00h
19 Update Clock <7:0> 40h

1A 6 Ramp Rate Clock <19:16> (Bits 23, 22, 21, 20 don’t care) 00h
1B Ramp Rate Clock <15:8> 00h
1C Ramp Rate Clock <7:0> 00h

7 Don’t Don’t Don’t Comp PD Reserved, QDAC PD DAC PD DIG PD 10h

1D Care Care Care Always
CR [31] Low
1E Don’t PLL Bypass Ref Mult 4 Ref Mult 3 Ref Mult 2 Ref Mult 1 Ref Mult 0 64h
Care Range PLL
1F CLR CLR Triangle SRC Mode 2 Mode 1 Mode 0 INT/EXT 01h
ACC 1 ACC 2 QDAC Update Clk
Don’t Bypass OSK EN OSK INT Don’t Don’t LSB First SDO 20h
Care Inv Care Care Active
20 Sinc CR [0]
21 8 Output Shape Key I Mult <11:8> (Bits 15, 14, 13, 12 don’t care) 00h
22 Output Shape Key I Mult <7:0> 00h

23 9 Output Shape Key Q Mult <11:8> (Bits 15, 14, 13, 12 don’t care) 00h
24 Output Shape Key Q Mult <7:0> 00h

25 A Output Shape Key Ramp Rate <7:0> 80h

26 B QDAC <11:8> (Bits 15, 14, 13, 12 don’t care) 00h

27 QDAC <7:0> (Data is required to be in two’s complement format) 0

NOTE
Shaded sections comprise the control register.

REV. B –25–
AD9854
TWR

A<5:0> A1 A2 A3

D<7:0> D1 D2 D3

WR
TASU TAHD
TDSU

TWRHIGH TWRLOW TDHD

SPECIFICATION VALUE DESCRIPTION

TASU 8.0ns ADDRESS SETUP TIME TO WR SIGNAL ACTIVE
TDSU 3.0ns DATA SETUP TIME TO WR SIGNAL ACTIVE
TADH 0ns ADDRESS HOLD TIME TO WR SIGNAL INACTIVE
TDHD 0ns DATA HOLD TIME TO WR SIGNAL INACTIVE
TWRLOW 2.5ns WR SIGNAL MINIMUM LOW TIME
TWRHIGH 7ns WR SIGNAL MINIMUM HIGH TIME
TWR 10.5ns MINIMUM WRITE TIME

Figure 27. Parallel Port Write Timing Diagram

Master RESET—Logic high active, must be held high for a mini- this information to transfer to the register bank, putting the de-
mum of 10 system clock cycles. This causes the communications vice in external update mode.
bus to be initialized and loads default values listed in Table IV.
Parallel I/O Operation Table V. Serial I/O Pin Requirements
With the S/P SELECT pin tied high, the parallel I/O mode is Pin Pin
active. The I/O port is compatible with industry standard DSPs Number Name Serial I/O Description
and microcontrollers. Six address bits, eight bidirectional data
bits and separate write/read control inputs make up the I/O 1, 2, 3, 4, D[7:0] The parallel data pins are not active, tie
port pins. 5, 6, 7, 8 to VDD or GND.
Parallel I/O operation allows write access to each byte of any regis- 14, 15, 16 A[5:3] The parallel address Pins A5, A4, A3
ter in a single I/O operation up to 1/10.5 ns. Read back capability are not active, tie to VDD or GND.
for each register is included to ease designing with the AD9854. 17 A2 IO RESET
Reads are not guaranteed at 100 MHz as they are intended for 18 A1 SDO
software debug only. 19 A0 SDIO
Parallel I/O operation timing diagrams are shown in Figures 20 I/O UD Update Clock. Same functionality for
26 and 27. CLOCK Serial Mode as Parallel Mode.
21 WRB SCLK
Serial Port I/O Operation
With the S/P SELECT pin tied low, the serial I/O mode is active. 22 RDB CSB—Chip Select
The AD9854 serial port is a flexible, synchronous, serial com-
munications port allowing easy interface to many industry-standard GENERAL OPERATION OF THE SERIAL INTERFACE
microcontrollers and microprocessors. The serial I/O is compat- There are two phases to a serial communication cycle with the
ible with most synchronous transfer formats, including both the AD9854. Phase 1 is the instruction cycle, which is the writing
Motorola 6905/11 SPI and Intel 8051 SSR protocols. The inter- of an instruction byte into the AD9854, coincident with the first
face allows read/write access to all twelve registers that configure eight SCLK rising edges. The instruction byte provides the
the AD9854 and can be configured as a single pin I/O (SDIO) AD9854 serial port controller with information regarding the data
or two unidirectional pins for in/out (SDIO/SDO). Data transfers transfer cycle, which is Phase 2 of the communication cycle. The
are supported in most significant bit (MSB) first format or least Phase 1 instruction byte defines whether the upcoming data trans-
significant bit (LSB) first format at up to 10 MHz. fer is read or write, and the register address to be acted upon.
When configured for serial I/O operation, most pins from the The first eight SCLK rising edges of each communication cycle
AD9854 parallel port are inactive; some are used for the serial are used to write the instruction byte into the AD9854. The
I/O. Table V describes pin requirements for serial I/O. remaining SCLK edges are for Phase 2 of the communication
Note: When operating in the serial I/O mode, it is best to use cycle. Phase 2 is the actual data transfer between the AD9854
the external I/O update clock mode to avoid an I/O update CLK and the system controller. The number of data bytes transferred
during a serial communication cycle. Such an occurrence could in Phase 2 of the communication cycle is a function of the regis-
cause incorrect programming due to partial data transfer. To exit ter address. The AD9854 internal serial I/O controller expects
the default internal update mode, at power-up, before starting every byte of the register being accessed to be transferred. Table
the REFCLK signal, but after a Master Reset program the device VI describes how many bytes must be transferred. Hence the user
for external update operation. Starting the REFCLK will cause would want to write between I/O update clocks.

–26– REV. B
 AD9854
Table VI. Register Address vs. Data Bytes Transferred R/W—Bit 7 of the instruction byte determines whether a read or
write data transfer will occur following the instruction byte.
Serial Number Logic high indicates read operation. Logic zero indicates a write
Register of Bytes operation.
Address Register Name Transferred
Bits 6, 5, and 4 of the instruction byte are dummy bits (don’t care).
0 Phase Offset Tuning Word Register #1 2 Bytes
1 Phase Offset Tuning Word Register #2 2 Bytes A3, A2, A1, A0—Bits 3, 2, 1, 0 of the instruction byte determine
2 Frequency Tuning Word #1 6 Bytes which register is accessed during the data transfer portion of the
3 Frequency Tuning Word #2 6 Bytes communications cycle. See Table VI for register address details.
4 Delta Frequency Register 6 Bytes Serial Interface Port Pin Description
5 Update Clock Rate Register 4 Bytes
6 Ramp Rate Clock Register 3 Bytes
SCLK
7 Control Register 4 Bytes Serial Clock (Pin 21). The serial clock pin is used to synchronize
8 I Path Digital Multiplier Register 2 Bytes data to and from the AD9854 and to run the internal state
9 Q Path Digital Multiplier Register 2 Bytes machines. SCLK maximum frequency is 10 MHz.
A Shaped On/Off Keying Ramp Rate Register 1 Byte CS
B Q DAC Register 2 Bytes Chip Select (Pin 22). Active low input that allows more than
one device on the same serial communications lines. The SDO
At the completion of any communication cycle, the AD9854 and SDIO pins will go to a high impedance state when this
serial port controller expects the next eight rising SCLK edges input is high. If driven high during any communications cycle,
to be the instruction byte of the next communication cycle. In that cycle is suspended until CS is reactivated low. Chip Select
addition, an active high input on the IO RESET pin immediately can be tied low in systems that maintain control of SCLK.
terminates the current communication cycle. After IO RESET
returns low, the AD9854 serial port controller requires the next SDIO
eight rising SCLK edges to be the instruction byte of the next Serial Data I/O (Pin 19). Data is always written into the AD9854
communication cycle. on this pin. However, this pin can be used as a bidirectional
data line. The configuration of this pin is controlled by Bit 0 of
All data input to the AD9854 is registered on the rising edge of register address 20h. The default is logic zero, which configures
SCLK. All data is driven out of the AD9854 on the falling edge the SDIO pin as bidirectional.
of SCLK.
SDO
Figures 28 and 29 are useful in understanding the general opera- Serial Data Out (Pin 18). Data is read from this pin for proto-
tion of the AD9854 Serial Port. cols that use separate lines for transmitting and receiving data.
In the case where the AD9854 operates in a single bidirectional
CS
I/O mode, this pin does not output data and is set to a high
impedance state.
INSTRUCTION
BYTE DATA BYTE 1 DATA BYTE 2 DATA BYTE 3 IO RESET
SDIO Synchronize I/O Port (Pin 17). Synchronizes the I/O port state
INSTRUCTION DATA TRANSFER machines without affecting the contents of the addressable regis-
CYCLE
ters. An active high input on IO RESET pin causes the current
communication cycle to terminate. After IO RESET returns low
Figure 28. Using SDIO as a Read/ Write Transfer
(Logic 0) another communication cycle may begin, starting with
the instruction byte.
CS Notes on Serial Port Operation
INSTRUCTION The AD9854 serial port configuration bits reside in Bits 1 and 0
BYTE of register address 20h. It is important to note that the configura-
SDIO
tion changes immediately upon a valid I/O update. For multibyte
INSTRUCTION DATA TRANSFER transfers, writing this register may occur during the middle of a
CYCLE
DATA BYTE 1 DATA BYTE 2 DATA BYTE 3 communication cycle. Care must be taken to compensate for
SDO this new configuration for the remainder of the current commu-
DATA TRANSFER nication cycle.
Figure 29. Using SDIO as an Input, SDO as an Output The system must maintain synchronization with the AD9854 or
the internal control logic will not be able to recognize further
Instruction Byte instructions. For example, if the system sends the instruction to
The instruction byte contains the following information. write a 2-byte register, then pulses the SCLK pin for a 3-byte
register (24 additional SCLK rising edges), communication
Table VII. Instruction Byte Information synchronization is lost. In this case, the first 16 SCLK rising edges
MSB D6 D5 D4 D3 D2 D1 LSB after the instruction cycle will properly write the first two data
bytes into the AD9854, but the next eight rising SCLK edges
R/W X X X A3 A2 A1 A0 are interpreted as the next instruction byte, NOT the final byte
of the previous communication cycle.

REV. B –27–
AD9854
In the case where synchronization is lost between the system and CR[31:29] are open.
the AD9854, the IO RESET pin provides a means to reestablish CR[28] is the comparator power-down bit. When this bit is set
synchronization without reinitializing the entire chip. Asserting (Logic 1), this signal indicates to the comparator that a power-
the IO RESET pin (active high) resets the AD9854 serial port state down mode is active. This bit is an output of the digital section
machine, terminating the current IO operation and putting the and is an input to the analog section.
device into a state in which the next eight SCLK rising edges
are understood to be an instruction byte. The IO RESET pin CR[27] must always be written to logic zero. Writing this bit to
must be deasserted (low) before the next instruction byte write can Logic 1 causes the AD9854 to stop working until a master
begin. Any information that had been written to the AD9854 reset is applied.
registers during a valid communication cycle prior to loss of CR[26] is the Q DAC power-down bit. When this bit is set
synchronization will remain intact. (Logic 1), this signal indicates to the Q DAC that a power-down
mode is active.
TPRE
TSCLK CR[25] is the full DAC power-down bit. When this bit is set
CS
(Logic 1), this signal indicates to both the I and Q DACs as well
TDSU TSCLKPWH TSCLKPWL
as the reference that a power-down mode is active.
SCLK CR[24] is the digital power-down bit. When this bit is set
(Logic 1), this signal indicates to the digital section that a
TDHLD
power-down mode is active. Within the digital section, the
SDIO 1ST BIT 2ND BIT clocks will be forced to dc, effectively powering down the digital
section. The PLL will still accept the REFCLK signal and con-
SYMBOL MIN DEFINITION
tinue to output the higher frequency.
TPRE 30ns CS SETUP TIME
TSCLK 100ns PERIOD OF SERIAL DATA CLOCK CR[23] is reserved. Write to zero.
TDSU 30ns SERIAL DATA SETUP TIME
TSCLKPWH 40ns SERIAL DATA CLOCK PULSEWIDTH HIGH CR[22] is the PLL range bit. The PLL range bit controls the
TSCLKPWL 40ns SERIAL DATA CLOCK PULSEWIDTH LOW
TDHLD 0ns SERIAL DATA HOLD TIME
VCO gain. The power-up state of the PLL range bit is Logic 1,
higher gain for frequencies above 200 MHz.
Figure 30. Timing Diagram for Data Write to AD9854
CR[21] is the bypass PLL bit, active high. When this bit is active,
the PLL is powered down and the REFCLK input is used to
CS drive the system clock signal. The power-up state of the bypass
PLL bit is Logic 1, PLL bypassed.
SCLK CR[20:16] bits are the PLL multiplier factor. These bits are the
REFCLK multiplication factor unless the bypass PLL bit is set.
SDIO
1ST BIT 2ND BIT
The PLL multiplier valid range is from 4 to 20, inclusive.
SDO
TDV
CR[15] is the clear accumulator 1 bit. This bit has a one-shot
type function. When this bit is written active, Logic 1, a clear
SYMBOL MAX DEFINITION
accumulator 1 signal is sent to the DDS logic, resetting the accu-
TDV 30ns DATA VALID TIME
mulator value to zero. The bit is then automatically reset, but the
Figure 31. Timing Diagram for Read from AD9854 buffer memory is not reset. This bit allows the user to easily cre-
ate a sawtooth frequency sweep pattern with minimal user
MSB/LSB TRANSFERS intervention. This bit is intended for chirp mode only, but its
The AD9854 serial port can support both most significant bit function is still retained in other modes.
(MSB) first or least significant bit (LSB) first data formats. This CR[14] is the clear accumulator bit. This bit, active high, holds
functionality is controlled by Bit 1 of serial register bank 20h. both the accumulator 1 and accumulator 2 values at zero for as
When this bit is set active high, the AD9854 serial port is in LSB long as the bit is active. This allows the DDS phase to be initial-
first format. This bit defaults low, to the MSB first format. The ized via the I/O port.
instruction byte must be written in the format indicated by Bit 1
of serial register bank 20h. That is, if the AD9854 is in LSB first CR[13] is the triangle bit. When this bit is set, the AD9854 will
mode, the instruction byte must be written from least significant automatically perform a continuous frequency sweep from F1 to
bit to most significant bit. F2 frequencies and back. The effect is a triangular frequency
sweep. When this bit is set, the operating mode must be set to
Control Register Description ramped FSK.
The Control Register is located in the shaded portion of Table IV
at address 1D through 20 hex. It is composed of 32 bits. CR[12] is the source Q DAC bit. When this bit is set high, the
Bit 31 is located at the top left position and Bit 0 is located in the Q path DAC accepts data from the Q DAC Register.
lower right position of the shaded table portion. The register has CR[11:9] are the three bits that describe the five operating modes
been subdivided below to make it easier to locate the text asso- of the AD9854:
ciated with specific control categories. 0h = Single-Tone Mode
1h = FSK Mode
2h = Ramped FSK mode
3h = Chirp Mode
4h = BPSK Mode
–28– REV. B
 AD9854
INSTRUCTION CYCLE DATA TRANSFER CYCLE
CS

SCLK

SDIO I7 I6 I5 I4 I3 I2 I1 I0 D7 D6 D5 D4 D3 D2 D1 D0

Figure 32. Serial Port Write Timing–Clock Stall Low

INSTRUCTION CYCLE DATA TRANSFER CYCLE

CS

SCLK

SDIO I7 I6 I5 I4 I3 I2 I1 I0 DON'T CARE

SDO DO 7 DO 6 DO 5 DO 4 DO 3 DO 2 DO 1 DO 0

Figure 33. 3-Wire Serial Port Read Timing–Clock Stall Low

INSTRUCTION CYCLE DATA TRANSFER CYCLE

CS

SCLK

SDIO I7 I6 I5 I4 I3 I2 I1 I0 D7 D6 D5 D4 D3 D2 D1 D0

Figure 34. Serial Port Write Timing–Clock Stall High

INSTRUCTION CYCLE DATA TRANSFER CYCLE

CS

SCLK

SDIO I7 I6 I5 I4 I3 I2 I1 I0 DO 7 DO 6 DO 5 DO 4 DO 3 DO 2 DO 1 DO 0

Figure 35. 2-Wire Serial Port Read Timing–Clock Stall High

CR[8] is the internal update active bit. When this bit is set to CR[3:2] are reserved. Write to zero.
Logic 1, the I/O UD pin is an output and the AD9854 generates CR[1] is the serial port MSB/LSB first bit. Defaults low,
the I/O UD signal. When set to Logic 0, external I/O UD function- MSB first.
ality is performed and the I/O UD pin is configured as an input.
CR[0] is the serial port SDO active bit. Defaults low, inactive.
CR[7] is reserved. Write to zero.
CR[6] is the inverse sinc filter BYPASS bit. When this bit is POWER DISSIPATION AND THERMAL
set, the data from the DDS block goes directly to the output CONSIDERATIONS
shaped-keying logic and the clock to the inverse sinc filter is The AD9854 is a multifunctional, very high-speed device that
stopped. Default is clear, filter enabled. targets a wide variety of synthesizer and agile clock applications.
CR[5] is the shaped-keying enable bit. When this bit is set, the The set of numerous innovative features contained in the device
output ramping function is enabled and is performed in accor- each consume incremental power. If enabled in combination,
dance with the CR[4] bit requirements. the safe thermal operating conditions of the device may be
exceeded. Careful analysis and consideration of power dissipa-
CR[4] is the internal/external output shaped-keying control tion and thermal management is a critical element in the successful
bit. When this bit is set to Logic 1, the shaped-keying factor will application of the AD9854 device.
be internally generated and applied to both the I and Q paths.
When cleared (default), the output shaped-keying function is The AD9854 device is specified to operate within the industrial
externally controlled by the user and the shaped-keying factor is temperature range of –40°C to +85°C. This specification is
the I and Q output shaped-keying factor register value. The two conditional, however, such that the absolute maximum junction
registers that are the shaped-keying factors also default low such temperature of 150°C is not exceeded. At high operating tempera-
that the output is off at power-up and until the device is pro- tures, extreme care must be taken in the operation of the device
grammed by the user.

REV. B –29–
AD9854
to avoid exceeding the junction temperature, which results in a Clock Speed—This directly and linearly influences the total
potentially damaging thermal condition. power dissipation of the device, and, therefore, junction tem-
Many variables contribute to the operating junction tempera- perature. As a rule, the user should always select the lowest
ture within the device, including: internal clock speed possible to support a given application, to
minimize power dissipation. Normally the usable frequency out-
1. Package Style put bandwidth from a DDS is limited to 40% of the clock rate
2. Selected Mode of Operation to keep reasonable requirements on the output low-pass filter.
3. Internal System Clock Speed For the typical DDS application, the system clock frequency
4. Supply Voltage should be 2.5 times the highest desired output frequency.
5. Ambient Temperature.
Mode of Operation—The selected mode of operation for the
The combination of these variables determines the junction AD9854 has a great influence on total power consumption. The
temperature within the AD9854 device for a given set of operating AD9854 offers many features and modes, each of which imposes
conditions. an additional power requirement. The collection of features
The AD9854 device is available in two package styles: a thermally contained in the AD9854 target a wide variety of applications
enhanced, surface-mount package with an exposed heat sink, and the device was designed under the assumption that only a
and a nonthermally enhanced, surface-mount package. The few features would be enabled for any given application. In fact,
thermal impedance of these packages is 16°C/W and 38°C/W the user must understand that enabling multiple features at higher
respectively, measured under still-air conditions. clock speeds may cause the maximum junction temperature of
the die to be exceeded. This can severely limit the long-term
THERMAL IMPEDANCE reliability of the device. Figures 36a and 36b provide a summary
The thermal impedance of a package can be thought of as a of the power requirements associated with the individual fea-
thermal resistor that exists between the semiconductor surface tures of the AD9854. These charts should be used as a guide
and the ambient air. The thermal impedance of a package is in determining the optimum application of the AD9854 for
determined by package material and its physical dimensions. The reliable operation.
dissipation of the heat from the package is directly dependent upon As can be seen in Figure 36b, the Inverse Sinc filter function
the ambient air conditions and the physical connection made requires a significant amount of power. As an alternate approach
between the IC package and the PCB. Adequate dissipation of to maintaining flatness across the output bandwidth, the digital
power from the AD9854 relies upon all power and ground pins multiplier function may be used to adjust the output signal level,
of the device being soldered directly to a copper plane on a PCB. at a dramatic savings in power consumption. Careful planning and
In addition, the thermally enhanced package of the AD9854ASQ management in the use of the feature set will minimize power dis-
contains a heat sink on the bottom of the package that must be sipation and avoid exceeding junction temperature requirements
soldered to a ground pad on the PCB surface. This pad must be within the IC.
connected to a large copper plane which, for convenience, may be
ground plane. Sockets for either package style of the AD9854 Figure 36a shows the supply current consumed by the AD9854
device are not recommended. over a range of frequencies for two possible configurations: All
circuits enabled means the output scaling multipliers, the inverse
JUNCTION TEMPERATURE CONSIDERATIONS sinc filter, the Q DAC, and the on-board comparator are all
The power dissipation (PDISS) of the AD9854 device in a given enabled, while basic configuration means the output scaling
application is determined by many operating conditions. Some multipliers, the inverse sinc filter, the Q DAC, and the on-board
of the conditions have a direct relationship with PDISS, such as comparator are all disabled.
supply voltage and clock speed, but others are less deterministic.
1400
The total power dissipation within the device, and its effect
on the junction temperature, must be considered when using the
1200
device. The junction temperature of the device is given by: ALL CIRCUITS ENABLED
Junction Temperature = (Thermal Impedance ×
SUPPLY CURRENT – mA

1000
Power Consumption) + Ambient Temperature
800
Given that the junction temperature should never exceed 150°C
for the AD9854, and that the ambient temperature can be 85°C, 600
the maximum power consumption for the AD9854AST is 1.7 W
and the AD9854ASQ (thermally-enhanced package) is 4.1 W. 400
Factors affecting the power dissipation are:
Supply Voltage—This obviously affects power dissipation and 200
BASIC CONFIGURATION
junction temperature since PDISS equals V × I. Users should design
0
for 3.3 V nominal; however, the device is guaranteed to meet 20 60 100 140 180 220 260 300
specifications, over the full temperature range and over the sup- FREQUENCY – MHz

ply voltage range of 3.135 V to 3.465 V. Figure 36a. Current Consumption vs. Clock Frequency

–30– REV. B
 AD9854
Figure 36b shows the approximate current consumed by each of THERMALLY ENHANCED PACKAGE MOUNTING
four functions. GUIDELINES
The following are general recommendations for mounting the
500 thermally enhanced exposed heat sink package (AD9854ASQ)
450 INVERSE SINC FILTER to printed circuit boards. The exceptional thermal characteristics of
this package depend entirely upon proper mechanical attachment.
400
Figure 37 depicts the package from the bottom and shows the
SUPPLY CURRENT – mA

350
dimensions of the exposed heat sink. A solid conduit of solder
300 needs to be established between this pad and the surface of
250 the PCB.
200 OUTPUT SCALING
MULTIPLIERS
150

100 Q DAC
COMPARATOR
50

0
20 60 100 140 180 220 260 300
FREQUENCY – MHz

Figure 36b. Current Consumption by Function vs. Clock

Frequency 10mm 14mm

EVALUATION OF OPERATING CONDITIONS

The first step in applying the AD9854 is to select the internal
clock frequency. Clock frequency selections above 200 MHz U NT
will require the thermally enhanced package (AD9854ASQ);
CO

RY
clock frequency selections of 200 MHz and below may allow
the use of the standard plastic surface-mount package, but more
information will be needed to make that determination.
The second step is to determine the maximum required operating
temperature for the AD9854 in the given application. Subtract Figure 37.
this value from 150°C, which is the maximum junction tem- Figure 38 depicts a general PCB land pattern for such an exposed
perature allowed for the AD9854. For the extended industrial heat sink device. Note that this pattern is for a 64-lead device, not
temperature range, the maximum operating temperature is 85°C, an 80-lead, but the relative shapes and dimensions still apply.
which results in a difference of 65°C. This is the maximum In this land pattern, a solid copper plane exists inside the indi-
temperature gradient that the device may experience due to vidual lands for device leads. Note also that the solder mask
power dissipation. opening is conservatively dimensioned to avoid any assembly
The third step is to divide this maximum temperature gradient problems.
by the thermal impedance, to arrive at the maximum power dis-
SOLDER MASK
sipation allowed for the application. For the example so far, 65°C OPENING
divided by both versions of the AD9854 package’s thermal imped- THERMAL LAND
ances of 38°C/W and 16°C/W, yields a total power dissipation
limit of 1.7 W and 4.1 W (respectively). This means that for a
3.3 V nominal power supply voltage, the current consumed by the
device under full operating conditions must not exceed 515 mA
in the standard plastic package and 1242 mA in the thermally
enhanced package. The total set of enabled functions and
operating conditions of the AD9854 application must support
these current consumption limits.
Figures 36a and Figure 36b may be used to determine the
suitability of a given AD9854 application vs. power dissipation
requirements. These graphs assume that the AD9854 device will
be soldered to a multilayer PCB per the recommended best
manufacturing practices and procedures for the given package
type. This ensures that the specified thermal impedance spec-
ifications will be achieved.

Figure 38.

REV. B –31–
AD9854
The thermal land itself must be able to distribute heat to an even EVALUATION BOARD INSTRUCTIONS
larger copper plane such as an internal ground plane. Vias must be Introduction
uniformly provided over the entire thermal pad to connect to this The AD9852/AD9854 Rev E evaluation board includes either
internal plane. A proposed via pattern is shown in Figure 39. Via an AD9852ASQ or AD9854ASQ IC.
holes should be small (12 mils, 0.3 mm) such that they can be The ASQ package permits 300 MHz operation by virtue of its
plated and plugged. These will provide the mechanical conduit thermally enhanced design. This package has a bottom-side heat
for heat transfer. “slug” that must be soldered to the ground plane of the PCB
directly beneath the IC. In this manner, the evaluation board
PCB ground plane layer extracts heat from the AD9852/AD9854
IC package. If device operation is limited to 200 MHz and below,
the AST package without a heat slug may be used in customer
installations over the full temperature range. The AST package
is less expensive than the ASQ package and those costs are
reflected in the price of the IC.
Evaluation boards for both the AD9852 and AD9854 are identi-
cal except for the installed IC.
The AD9852 or AD9854 data sheet is essential to understand-
ing all the modes of operation. While various Preliminary data
sheets have been prepared and disseminated, only the released
data sheet should be used since errors and omissions in the pre-
liminary data sheets are inevitable. A released data sheet will
Figure 39. have no Preliminary markings and will display a revision status
Finally, a proposed stencil design is shown in Figure 40 for screen such as “REV 0” or “REV A” at the lower left corner of each page.
solder placement. Note that if vias are not plugged, wicking will To assist in proper placement of the pin-header shorting-jumpers,
occur, which will displace solder away from the exposed heat sink, the instructions will refer to direction (left, right, top, bottom)
and the necessary mechanical bond will not be established. as well as header pins to be shorted. Pin #1 for each three pin-
header has been marked on the PCB corresponding with the
schematic diagram. When following these instructions, position the
PCB so that the PCB text can be read from left to right. The board
is shipped with the pin-headers configuring the board as follows:
1. REFCLK for the AD9852/AD9854 is configured as differ-
ential. The differential clock signals are provided by the
MC100LVEL16D differential receiver.
2. Input clock for the MC100LVEL16D is single-ended via
J25. This signal may be 3.3 V CMOS or a 2 V p-p sine
wave capable of driving 50 Ω (R13).
3. Both DAC outputs from the AD9852/AD9854 are routed
through the two 120 MHz elliptical LP filters and their out-
puts connected to J7 (Q or Control DAC) and J6 (I or
Cosine DAC).
4. The board is set up for software control via the printer port
connector.
5. The DAC’s output currents are configured for 10 mA.
Figure 40.
GENERAL OPERATING INSTRUCTIONS
EVALUATION BOARD Load the Version 1.71 software from the provided CD onto your
An evaluation board is available that supports the AD9854 DDS PC’s hard disk. Connect a printer cable from the PC to the
devices. This evaluation board consists of a PCB, software, and AD9854 Evaluation Board printer port connector labeled “J11.”
documentation to facilitate bench analysis of the performance of Version 1.71 software will support Windows 95, Windows 98,
the AD9854 device. It is recommended that users of the AD9854 and Windows NT.
familiarize themselves with the operation and performance Hardware Preparation: Using the schematic in conjunction with
capabilities of the device with the evaluation board. The evaluation these instructions will be helpful in acquainting the user with
board should also be used as a PCB reference design to ensure the electrical functioning of the evaluation board.
optimum dynamic performance from the device.
Attach power wires to connector labeled “TB1” using the screw-
down terminals. This is a plastic connector that press-fits over a
4-pin header soldered to the board. Table VIII shows connec-
tions to each pin. DUT = “device under test.”

–32– REV. B
 AD9854
Table VIII. Power Requirements for DUT Pins equipment offers 50 Ω inputs, the DAC will develop only
0.25 V p-p due to the double termination.
AVDD 3.3 V DVDD 3.3 V VCC 3.3 V Ground—
for All DUT for All DUT for All Other for All 1. Install shorting jumpers at W7 and W10.
Analog Pins Digital Pins Devices Devices 2. Remove shorting jumper at W16.
3. Remove shorting jumper from 3-pin header W1.
Attach REFCLK to clock input, J25.
4. Install shorting jumper on Pins 1 and 2 (bottom two pins) of
Clock Input, J25
3-pin header W4.
This is actually a single-ended input that will be routed to the
MC100LVEL16D for conversion to differential PECL output. If using the AD9852 evaluation board, IOUT2, the Control
This is accomplished by attaching a 2 V p-p clock or sine wave DAC output is under user control through the serial or parallel
source to J25. Note that this is a 50 Ω impedance point set by R13. ports. The 12-bit, two’s complement value(s) is/are written to
The input signal will be ac-coupled and then biased to the cen- the Control DAC register that will set the IOUT2 output to a
ter-switching threshold of the MC100LVEL16D. To engage the static dc level. Allowable hexadecimal values are 7FF (maximum)
differential-clocking mode of the AD9854, W3 Pins 2 and 3 (the to 800 (minimum) with all zeros being midscale. Rapidly changing
bottom two pins) must be connected with a shorting jumper. the contents of the Control DAC register (up to 100 MSPS)
allows IOUT2 to assume any waveform that can be programmed.
The signal arriving at the AD9854 is called the Reference Clock.
If you choose to engage the on-chip PLL clock multiplier, this Observing the Filtered IOUT1 and the Filtered IOUT2
signal is the reference clock for the PLL and the multiplied PLL This allows the viewer to observe the filtered “I” and “Q” (or
output becomes the System Clock. If the user chooses to bypass Control) DAC outputs at J6 (the “I” signal) and J7 (the “Q” or
the PLL clock multiplier, the reference clock supplied by the Control signal). This places the 50 Ω (input and output Z) low-
user is directly operating the AD9854 and is, therefore, the pass filters in the “I” and “Q” (or Control) DAC pathways to
system clock. remove images and aliased harmonics and other spurious signals
above approximately 120 MHz. For “I” and “Q” signals, these
Three-State Control
signals will appear as nearly pure sine waves and 90 degrees
Three control or switch headers W9, W11, W12, W13, W14,
out-of-phase with each other. These filters are designed with the
and W15 must be shorted to allow the provided software to
assumption that the system clock speed is at or near maximum
control the AD9854 evaluation board via the printer port
(300 MHz). If your system clock speed is much less than
connector J11.
300 MHz, for example 200 MHz, it is possible or inevitable
Programming that unwanted DAC products other than the fundamental sig-
If programming of the AD9854 is not to be provided by the nal will be passed by the low-pass filters.
user’s PC and ADI software, Headers W9, W11, W12, W13,
If you are using the AD9852 evaluation board, any reference to
W14, and W15 should be opened (shorting jumpers removed).
the “Q” signal should be interpreted to mean “Control DAC.”
This effectively detaches the PC interface and allows the 40-pin
header, J10, and J1, to assume control without bus contention. 1. Install shorting jumpers at W7 and W10.
Input signals on J10 and J1 going to the AD9854 should be 3.3 V 2. Install shorting jumper at W16.
CMOS logic levels.
3. Install shorting jumper on Pins 1 and 2 (bottom two pins) of
Low-Pass Filter Testing 3-pin header W1.
The purpose of 2-pin headers W7 and W10 (associated with J4
4. Install shorting jumper on Pins 1 and 2 (bottom two pins) of
and J5) is to allow the two 50 Ω, 120 MHz filters to be tested
3-pin header W4.
during PCB assembly without interference from other circuitry
attached to the filter inputs. Normally, a shorting jumper will be 5. Install shorting jumper on Pins 2 and 3 (bottom two pins) of
attached to each header to allow the DAC signals to be routed to the 3-pin header W2 and W8.
filters. If the user wishes to test the filters, the shorting jumpers Observing the Filtered IOUT1 and the Filtered IOUT1B
at W7 and W10 should be removed and 50 Ω test signals applied This allows the viewer to observe only the filtered “I” DAC
at J4 and J5 inputs to the 50 Ω elliptic filters. Users should refer outputs at J6 (the “true” signal) and J7 (the “complementary”
to the provided schematic and the following sections to properly signal). This places the 120 MHz low-pass filters in the true and
position the remaining shorting jumpers. complementary outputs paths of the “I” DAC to remove images
Observing the Unfiltered IOUT1 and the Unfiltered IOUT2 and aliased harmonics and other spurious signals above approxi-
DAC Signals mately 120 MHz. These signals will appear as nearly pure sine
This allows the viewer to observe the unfiltered DAC outputs waves and 180 degrees out-of-phase with each other. If your
at J5 (the “I” or Cosine signal) and J4 (the “Q” or Control system clock speed is much less than 300 MHz, for example
DAC signal). The procedure below simply routes the two 200 MHz, it is possible or inevitable that unwanted DAC prod-
50 Ω terminated analog DAC outputs to the SMB connectors ucts other than the fundamental signal will be passed by the
and disconnects any other circuitry. The “raw” DAC outputs low-pass filters.
may appear as a series of quantized (stepped) output levels that 1. Install shorting jumpers at W7 and W10.
may not resemble a sine wave until they have been filtered. The
default 10 mA output current will develop a 0.5 V p-p signal 2. Install shorting jumper at W16.
across the on-board 50 Ω termination. If your “observation” 3. Install shorting jumper on Pins 2 and 3 (top two pins) of
equipment offers 50 Ω inputs, the DAC will develop only 3-pin header W1.
0.25 V p-p due to the double termination. If your “observation”

REV. B –33–
AD9854
4. Install shorting jumper on Pins 2 and 3 (top two pins) of • Detects the Windows Platform (Windows 95, Windows 98,
3-pin header W4. Windows NT).
5. Install shorting jumpers on Pins 2 and 3 (bottom two pins) • Installs the correct version of the software (Windows 95/98
of 3-pin header W2 and W8. or Windows NT).
To Connect the High-Speed Comparator • Detects if Windows NT has Service Pack 3 installed, and if it
To connect the high-speed comparator to the DAC output sig- does not, gives the option to install it.
nals, either the quadrature filtered output configuration (AD9854 • Allows access to the data sheets for both products through
only) or the complementary filtered output configuration out- hyperlinks. (The hyperlinks bring up the executable that is
lined above (both AD9854 and AD9852) can be chosen. Follow currently associated with Acrobat files.)
Steps 1 through 4 for either filtered configuration as above. Step
5 below will reroute the filtered signals away from their output The CD-ROM contains the following:
connectors (J6 and J7) and to the 100 Ω configured comparator • The AD9852/AD9854 Evaluation Software.
inputs. This sets up the comparator for differential input without
• Service Pack 3 for Windows NT. This is required for Visual
control of the comparator output duty cycle. The comparator out-
Basic 6 applications to run on Windows NT 4.0.
put duty cycle should be close to 50% in this configuration.
• Acrobat Reader 4.0 for Windows 95/98 and Windows NT.
5. Install shorting jumper on Pins 1 and 2 (top two pins) of
3-pin header W2 and W8. Several numerical entries, such as frequency and phase informa-
tion, require that the ENTER key be pressed to register that
Users may elect to change the RSET resistor, R2 from 3.9 kΩ to
information. So, for example, if a new frequency is input, the
1.95 kΩ to receive a more robust signal at the comparator inputs.
load button is hit, and nothing new happens, it is probably because
This will decrease jitter and extend comparator-operating range.
the user neglected to press the enter key after typing the new
Users can accomplish this by installing a shorting jumper at W6,
frequency information.
which provides a second 3.9 kΩ chip resistor (R20) in parallel
with the provided R2. This boosts the DAC output current 1. Normal operation of the AD9852/AD9854 evaluation board
from 10 mA to 20 mA and doubles the p-p output voltage de- begins with a master reset. Many of the default register values
veloped across the loads. after reset are depicted in the software “control panel.” The
reset command sets the DDS output amplitude to minimum
Single-Ended Configuration
and 0 Hz, 0 phase-offset as well as other states that are listed
To connect the high-speed comparator in a single-ended configu-
in the AD9852/AD9854 Register Layout table in the data sheet.
ration that will allow duty cycle or pulsewidth control requires that
a dc threshold voltage be present at one of the comparator inputs. 2. The next programming block should be the “Reference Clock
You may supply this voltage using the control DAC. A 12-bit, and Multiplier” since this information is used to determine
two’s complement value is written to the Control DAC register the proper 48-bit frequency tuning words that will be entered
that will set the IOUT2 output to a static dc level. Allowable and calculated later.
hexadecimal values are 7FF (maximum) to 800 (minimum) 3. The output amplitude defaults to the 12-bit straight binary
with all zeros being midscale. The IOUT1 channel will continue multiplier values of the “I or Cosine” multiplier register of
to output a filtered sine wave programmed by the user. These 000hex and no output (dc) should be seen from the DAC.
two signals are routed to the comparator using W2 and W8 Set the multiplier amplitude in the Output Amplitude window
3-pin header switches. The user must be in the configura- to a substantial value, such as FFF hex. The digital multiplier
tion described in the section “Observing the Filtered IOUT1 may be bypassed by clicking the box “Output Amplitude is
and the Filtered IOUT2.” Follow Steps 1 through 4 in that sec- always Full-Scale,” but experience has shown that doing so
tion and then the following: does not result in best SFDR. Best SFDR, as much as 11 dB
5. Install shorting jumper on Pins 1 and 2 (top two pins) of better, is obtained by routing the signal through the digital
3-pin header W2 and W8. multiplier and “backing off” on the multiplier amplitude. For
instance, FC0 hex produces less spurious signal amplitude
The user may elect to change the RSET resistor, R2 from 3.9 kΩ
than FFF hex. It is an exploitable and repeatable phenomenon
to 1.95 kΩ to receive a more robust signal at the comparator
that should be investigated in your application if SFDR (spuri-
inputs. This will decrease jitter and extend comparator-operating
ous-free dynamic range) must be maximized. This phenomenon
range. The user can accomplish this by installing a shorting
is more readily observed at higher output frequencies where
jumper at W6, which provides a second 3.9 kΩ chip resistor
good SFDR becomes more difficult to achieve.
(R20) in parallel with the provided R2.
4. Refer to this data sheet and evaluation board schematic to
USING THE PROVIDED SOFTWARE understand all the functions of the AD9854 available to the
The software is provided on a CD. This brief set of instructions user and to gain an understanding of what the software is
should be used in conjunction with the AD9852 or AD9854 data doing in response to programming commands.
sheet and the AD9852/AD9854 Evaluation Board schematic. Applications assistance is available for the AD9854, the
Version 1.71 Software has been improved from previous versions AD9854/PCB evaluation board, and all other Analog Devices
in the following ways: products. Please call 1/800-ANALOGD.
• Detects old versions of the software installed and gives
option to uninstall them.

–34– REV. B
REV. B
GND GND DVDD
J15
W3

AVDD

DVDD
DVDD
CLK

GND
RESET
PMODE
CLK8
GND J16 J8
1 GND
J17 J6 GND
GND AVDD J18 J11
R4 C1 J19 J12
80 79 78 77 76 75 74 73 72 71 70 69 68 67 66 65 64 63 62 61 1.3k 0.01F J20 J13
J22 J14
J24 J21 J6

NC5

GND4
GND3
J23

DVDD7
DVDD6

DVDD8
DVDD9
DGND8
DGND6
DGND7

DGND9
120 MHz LOW-PASS FILTER

PLLFLT

REFCLK
CLKVDD

MRESET
OPTGND
60

CLKGND
1 D7

REFCLKB
D7 PLLVDD AVDD C32 C33 C34
GND

SPSELECT
DIFFCLKEN
D6 2 D6 PLLGND 59 2.2pF 12pF 8.2pF
D5 3 D5 NC4 58 W6 R20 3.9k R1 W2
50 GND J4
D4 4 D4 NC3 57 GND 1
R2 3.9k L4 L5 L2
D3 5 D3 RSET 56 82nH 68nH 68nH
C45 0.1F
D2 6 D2 DACBYPASS 55 AVDD W7 GND
7 D1 54 GND C4 C5 C30 C31
D1 AVDD2 AVDD 1 27pF 47pF 39pF 22pF
D0 8 D0 U1 AGND2 53 W1
GND
DVDD 9 DVDD1 AD9854 IOUT2 52 GND GND GND GND
R3 25
DVDD 10 DVDD2 IOUT2B 51 J7
TOP VIEW
11 DGND1 (Not to Scale) AVDD 50 AVDD
GND 120 MHz LOW-PASS FILTER
GND 12 DGND2 IOUT1B 49 W4
C41 C42 C43 GND
GND 13 NC IOUT1 48 J5 1 R6 2.2pF 12pF 8.2pF
50
ADDR5 14 ADDR5 AGND 47 R7 W8
46 GND 25
ADDR4 15 ADDR4 GND2 L6 1
GND L3 L1
ADDR3 16 ADDR3 COMPGND 45 GND GND 82nH 68nH 68nH
W10 W16 GND
ADDR2 17 ADDR2 COMPVDD 44 GND
AVDD C37 C38 C39 C40
ADDR1 18 ADDR1 VINB 43
27pF 47pF 39pF 22pF
ADDR0 19 ADDR0 VIN 42 R5
50

–35–
UDCLK 20 UPDCLK GND 41
GND GND GND GND GND
GND
R8
100

RD
WR
FSK/BPSK/HOLD
DACDVDD2

OUTRAMP
COUTGND

DVDD3
DVDD4
DVDD5
DACDGND
COUTVDD
COUTGND2

VOUT

DGND3
DGND5
NC2

DGND4
DACDVDD
COUTVDD2

DACDGND2
GND Y2
21 22 23 24 25 26 27 28 29 30 31 32 33 34 35 36 37 38 39 40 3 2
OUT GND
NC = NO CONNECT
R10 J25 1 4
100 1 DVDD 3.3V NC GND

RD
GND W17

WR
GND U3 CLKB J3

GND
GND
GND

GND
GND
GND
GND
AVDD
AVDD

DVDD
AVDD
AVDD

DVDD
DVDD

Figure 41a. Evaluation Board Schematic

2 7

DRAMP
D Q
3 6 R19
J1 J26 R13 C2 D Q
GND R8 0 GND
50 0.01F 2k MC100LVEL16
FDATA GND GND R14 J2
0
VBB

VEE
VCC

DVDD
+
C25 C24 C23 C22 C27 C8 C44 5 4 8 CLK
W5 R11
10F 0.1F 0.1F 0.1F 0.1F 0.1F 0.1F
W18 50

RD
WR

ADR5
ADR4
ADR3
ADR2
ADR1
ADR0
ORAMP

D7
UDCLK
PMODE
RESET

D6
D5
D4
D3
D2
D1
D0
GND DVDD R12 GND
W19 50
W20 GND
VCC

40
39
38
37
36
35
34
33
32
31
30
29
28
27
26
25
24
23
22
21
+ C18 C17 C16 C26
J10 TB1 C21 C20 C19 C14 C28 GND
1 10F 0.1F 0.1F 0.1F 0.1F 0.1F 0.1F 0.1F 0.1F

9
8
7
1

6
5
4
3
2

20
19
18
17
16
15
14
13
12
11
10
AVDD
2
DVDD AVDD
3 GND
VCC C6 + C7 C29 C9 C10 C11 C12 C13
4 10F 0.1F 0.1F 0.1F 0.1F 0.1F 0.1F 0.1F
GND

GND
AD9854

GND
 VCC
AD9854
R18
10k

W15
VCC RP1
10k
1 2 3 4 5 6 7 8 9 10 U8 U9
1 GND
EN VCC: 20
1 EN VCC: 20
11 11
1 C1 GND: 10 C1 GND: 10
C0 U5 74HC574 74HC574
2 2 9 12 9 12
A0
1 1A 1Y 8D D0 8D ADDR5
3 8 13 8 13
A1
3 2A
2Y 4 D1 ADDR4
4 6 7 14 7 14
A2
5 3A
3Y D2 ADDR3
5 9 8 6 15 6 15 ADDR2
A3 4A 4Y D3
6 10 5 16 5 16 U10
A4 11 5A
5Y D4 1
7 4 17 EN VCC: 20
13 4 17
A5 6A 6Y 12 D5 11 C1 GND: 10
3 18 3 18
8 74HC14 D6 74HC574
A6 2 19 2 19 9 12
VCC GND 1D D7 1D 8D WR
A7 9 8 13
14 7 RD
7 14 RESET
VCC
J11 6 15
VCC GND UDCLK
36PINCONN 16 W12
5
GND:[19:30] PMODE
4 17
U6 ORAMP
W13
1 1A 3 18
10 1Y 2 FDATA
B6 2 1D 19 W9
3 2A 2Y 4
11
B7 VCC 5
3A 3Y 6

–36–
12
B5 9
13 4A 4Y 8
B4 11
5A 5Y 10
13 6A 6Y 12
U2
74HC14
14 1 1G
VCC GND VCC 14 VCC
C1 U4 2 1A
14 7 4G 13
1 1A 1Y 2 3 1Y
R15 VCC 4A 12
GND 4 ADDR1
10k 3 2A
2Y W11 4 2G 11
4Y

Figure 41b. Evaluation Board Schematic

5 3A 6 5 2A 10
VCC 3Y 3G
8 6 2Y 9
9 4A 4Y 3A
10 ADDR0 W14 7 GND 8
11 5A 5Y 3Y
VCC
U7 13 6A 6Y 12 74HC14
GND
R16 1 1A 1Y 2 74HC14
10k
3 2A 2Y 4 VCC GND
31 VCC
5 3A 6 14 7
C2 3Y
9 4A 4Y 8 VCC GND
VCC
32 10
B3 VCC
11 5A 5Y
13 6A 6Y 12
74HC14
VCC
VCC GND
14 7
R17 VCC GND
10k
36
C3

REV. B
 AD9854
AD9852/54 Customer Evaluation Board
(AD9852 PCB > U1 = AD9852ASQ, AD9854 PCB > U1 = AD9854ASQ)
# Quantity REFDES Device Package Value Mfg. Part No.
1 3 C1, C2, C45 CAP 0805 0.01 µF
2 21 C7, C8, C9, C10, C11, CAP 0603 0.1 µF
C12, C13, C14, C16, C17,
C18, C19, C20, C22, C23,
C24, C26, C27, C28, C29,
C44
3 2 C4, C37 CAP 1206 27 pF
4 2 C5, C38 CAP 1206 47 pF
5 3 C6, C21, C25 BCAPT TAJD 10 µF
6 2 C30, C39 CAP 1206 39 pF
7 2 C31, C40 CAP 1206 22 pF
8 2 C32, C41 CAP 1206 2.2 pF
9 2 C33, C42 CAP 1206 12 pF
10 2 C34, C43 CAP 1206 8.2 pF
11 9 J1, J2, J3, J4, J5, J6, J7 SMB STR-PC MNT ITT INDUSTRIES
J25, J26 B51–351–0000220
12 16 J8, J9, J11, J12, J13, J14, W-HOLE
J15, J16, J17, J18, J19, J20,
J21, J22, J23, J24
13 1 J10 DUAL ROW 40 PINS SAMTEC
HEADER TSW-120-23-L-D
14 4 L1, L2, L3, L5 IND-COIL 1008CS 68 nH COILCRAFT
1008CS-680XGBB
15 2 L4, L6 IND-COIL 1008CS 82 nH COILCRAFT
1008CS-820XGBB
16 2 R1, R5, R6, R11, R12, R13 RES 1206 50 Ω (49.9 Ω, 1%)
17 2 R2, R20 RES 1206 3900 Ω
18 2 R3, R7 RES 1206 25 Ω (24.9 Ω, 1%)
19 1 R4 RES 1206 1300 Ω
20 1 R8 RES 1206 2000 Ω
21 2 R9, R10 RES 1206 100 Ω
22 4 R15, R16, R17, R18 RES 1206 10 kΩ
23 1 RP1 RES NETWORK SIP-10P 10 kΩ Bourns
4610X-101-103
24 1 TB1 TERMINAL 4-POSITION WIELAND
BLOCK & PINS 25.602.2453.0 Block
Z5.530.3425.0 Pins

25 1 U1 AD9852 or 80 LQFP AD9852ASQ or

AD9854 AD9854ASQ
26 1 U2 74HC125 14 SO1C SN74HC125D
27 1 U3 MC100LVEL16D 8 SO1C MC100LVEL16D
28 4 U4, U5, U6, U7 74HC14 14 SO1C SN74HC14D
29 3 U8, U9, U10 74HC574 20 SO1C SN74HC574DW
30 1 J11 36-PIN AMP 552742-1
CONNECTOR
31 6 W1, W2, W3, W4, W8, W17 3-PIN JUMPER SAMTEC
32 10 W6, W7, W9, W10, W11, 2-PIN JUMPER SAMTEC
W12, W13, W14, W15, W16
33 2 SELF-TAPPING 4–40, PHILIPS,
SCREW ROUND HEAD
34 4 RUBBER SQUARE 3M
BUMPER BLACK SJ-5018SPBL
35 1 AD9852/54 PCB GSO2669 REV. E
36 2 R14, R19 Zero Ω JUMPER 1206 Zero Ω
37 4 Pin Socket AMP 5-330808-6
38 1 Y1 (Not Supplied) XTAL COSC (Not Supplied)

REV. B –37–
AD9854

Figure 42. Assembly Drawing

Figure 43. Top Routing Layer, Layer 1

–38– REV. B
 AD9854

Figure 45. Ground Plane Layer, Layer 2

Figure 44. Power Plane Layer, Layer 3

REV. B –39–
AD9854

Figure 46. Bottom Routing Layer, Layer 4

–40– REV. B
 AD9854
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).

80-Lead LQFP_ED
(SQ-80)

0.063 (1.60) 0.630 (16.00) BSC SQ

MAX 0.394 (10.00)
0.551 (14.00) BSC SQ
0.030 (0.75) REF SQ
0.024 (0.60) 80 61 61 80
0.018 (0.45) 1 60 60 1

SEATING PIN 1
PLANE

THERMAL
TOP VIEW SLUG
(PINS DOWN)
BOTTOM VIEW

COPLANARITY
0.004 (0.10) 20 41 41 20
MAX 21 40 40 21

0.006 (0.15)
0.002 (0.05) 0.057 (1.45)
0.055 (1.40)
0.053 (1.35)
0.008 (0.20) 0.0256 (0.65) 0.015 (0.38) 7
0.004 (0.09) BSC 0.013 (0.32) 3.5
0.009 (0.22) 0

CONTROLLING DIMENSIONS IN MILLIMETERS.

CENTER FIGURES ARE NOMINAL UNLESS OTHERWISE NOTED.

80-Lead LQFP
(ST-80)

0.063 (1.60) 0.630 (16.00) BSC SQ

MAX
0.551 (14.00) BSC SQ
0.030 (0.75)
0.024 (0.60) 80 61
0.018 (0.45) 1 60

SEATING PIN 1
PLANE

TOP VIEW
(PINS DOWN)

COPLANARITY
0.004 (0.10) 20 41
MAX 21 40

0.006 (0.15)
0.002 (0.05) 0.057 (1.45)
0.055 (1.40)
0.053 (1.35)
0.008 (0.20) 0.0256 (0.65) 0.015 (0.38) 7
0.004 (0.09) BSC 0.013 (0.32) 3.5
0.009 (0.22) 0

CONTROLLING DIMENSIONS IN MILLIMETERS.

CENTER FIGURES ARE NOMINAL UNLESS OTHERWISE NOTED.

REV. B –41–
AD9854
Revision History
Location Page

Data Sheet changed from Rev. A to Rev. B

Edits to GENERAL DESCRIPTION . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1
Edits to FUNCTIONAL BLOCK DIAGRAM . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1
Edits to SPECIFICATIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 - 5
Edit to Absolute Maximum Ratings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
Edits to PIN FUNCTION DESCRIPTIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
Edits to Figure 1 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
Deletion of two Typical Performance Characteristics graphs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
Edits to Inverse SINC Function section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
Edits to Differential REFCLK Enable section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
Edits to Figure 27 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
Edits to Parallel I/O Operation section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
Edits to GENERAL OPERATION OF THE SERIAL INTERFACE section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
Edit to Figure 31 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28
Replacement of OPERATING INSTRUCTIONS section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32
Edits to Figure 41a . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
Edits to Figure 41b . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36
Edits to AD9852/AD9854 Customer Evaluation Board chart . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37

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PRINTED IN U.S.A. C00636–0–3/02(B)