a 16-Bit 100 kSPS

Sampling ADC
AD676
FEATURES FUNCTIONAL BLOCK DIAGRAM
Autocalibrating
On-Chip Sample-Hold Function
Parallel Output Format VIN 15 ANALOG
CHIP
16 Bits No Missing Codes AGND SENSE 14 INPUT 16-BIT
COMP
61 LSB INL VREF 16 BUFFERS DAC

–97 dB THD AGND 13

90 dB S/(N+D) CAL
DAC
1 MHz Full Power Bandwidth
LOGIC & TIMING

LEVEL TRANSLATORS

DIGITAL
CHIP 7 BUSY
SAR
CAL 8 1
PAT L
MICRO-CODED GEN A 6
SAMPLE 9 T BIT 1 – BIT 16
CONTROLLER 19
ALU C
PRODUCT DESCRIPTION CLK 10 H
28
RAM
The AD676 is a multipurpose 16-bit parallel output analog-to-
digital converter which utilizes a switched-capacitor/charge AD676
redistribution architecture to achieve a 100 kSPS conversion
rate (10 µs total conversion time). Overall performance is opti-
mized by digitally correcting internal nonlinearities through
on-chip autocalibration.
The AD676 operates from +5 V and ± 12 V supplies and typi-
The AD676 circuitry is segmented onto two monolithic chips— cally consumes 360 mW during conversion. The digital supply
a digital control chip fabricated on Analog Devices DSP CMOS (VDD) is separated from the analog supplies (VCC, VEE) for re-
process and an analog ADC chip fabricated on our BiMOS II duced digital crosstalk. An analog ground sense is provided for
process. Both chips are contained in a single package. the analog input. Separate analog and digital grounds are also
The AD676 is specified for ac (or “dynamic”) parameters such provided.
as S/(N+D) Ratio, THD and IMD which are important in sig- The AD676 is available in a 28-pin plastic DIP or 28-pin side-
nal processing applications. In addition, dc parameters are brazed ceramic package. A serial-output version, the AD677, is
specified which are important in measurement applications. available in a 16-pin 300 mil wide ceramic or plastic package.

REV. A
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which may result from its use. No license is granted by implication or One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
otherwise under any patent or patent rights of Analog Devices. Tel: 617/329-4700 Fax: 617/326-8703
AD676* PRODUCT PAGE QUICK LINKS
Last Content Update: 02/23/2017

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AD676–SPECIFICATIONS
AC SPECIFICATIONS (T MIN to TMAX, VCC = +12 V 6 5%, VEE = –12 V 6 5%, VDD = +5 V 6 10%)1
AD676J/A AD676K/B
Parameter Min Typ Max Min Typ Max Units
2
Total Harmonic Distortion (THD)
@ 83 kSPS, TMIN to TMAX –96 –88 –97 –90 dB
0.0016 0.004 0.0014 0.003 %
@ 100 kSPS, +25°C –96 –97 dB
0.0016 0.0014 %
@ 100 kSPS, TMIN to TMAX –92 –92 dB
0.0025 0.0025 %
Signal-to-Noise and Distortion Ratio (S/(N+D))2, 3
@ 83 kSPS, TMIN to TMAX 85 89 87 90 dB
@ 100 kSPS, +25°C 89 90 dB
@ 100 kSPS, TMIN to TMAX 86 86 dB
Peak Spurious or Peak Harmonic Component –98 –98 dB
Intermodulation Distortion (IMD)4
2nd Order Products –102 –102 dB
3rd Order Products –98 –98 dB
Full Power Bandwidth 1 1 MHz
Noise 160 160 µV rms

DIGITAL SPECIFICATIONS (for all grades T MIN to TMAX, VCC = +12 V 6 5%, VEE = –12 V 6 5%, VDD = +5 V 6 10%)
Parameter Test Conditions Min Typ Max Units
LOGIC INPUTS
VIH High Level Input Voltage 2.4 VDD + 0.3 V
VIL Low Level Input Voltage –0.3 0.8 V
IIH High Level Input Current VIH = VDD –10 +10 µA
IIL Low Level Input Current VIL = 0 V –10 +10 µA
CIN Input Capacitance 10 pF
LOGIC OUTPUTS
VOH High Level Output Voltage IOH = 0.1 mA VDD –1 V V
IOH = 0.5 mA 2.4 V
VOL Low Level Output Voltage IOL = 1.6 mA 0.4 V
NOTES
1
VREF = 10.0 V, (Conversion Rate (fs) = 83 kSPS, f IN = 1.0 kHz, VIN = –0.05 dB, Bandwidth = fs/2 unless otherwise indicated. All measurements referred to a 0 dB
(20 V p-p) input signal. Values are post-calibration.
2
For other input amplitudes, refer to Figure 13.
3
For other input ranges/voltages reference values see Figure 12.
4
fa = 1008 Hz. fb = 1055 Hz. See Definition of Specifications section and Figure 15.
Specifications subject to change without notice.

–2– REV. A
 AD676
DC SPECIFICATIONS (T MIN to TMAX, VCC = +12 V 6 5%, VEE = –12 V 6 5%, VDD = +5 V 6 1O%)1
AD676J/A AD676K/B
Parameter Min Typ Max Min Typ Max Units
TEMPERATURE RANGE
J, K Grades 0 +70 0 +70 °C
A, B Grades –40 +85 –40 +85 °C
ACCURACY
Resolution 16 16 Bits
Integral Nonlinearity (INL)
@ 83 kSPS, TMIN to TMAX ±1 ±1 ± 1.5 LSB
@ 100 kSPS, +25°C ±1 ±1 LSB
@ 100 kSPS, TMIN to TMAX ±2 ±2 LSB
Differential Nonlinearity (DNL)–No Missing Codes 16 16 Bits
Bipolar Zero Error2 (at Nominal Supplies) 0.005 0.005 % FSR
Gain Error (at Nominal Supplies)
@ 83 kSPS2 0.005 0.005 % FSR
@ 100 kSPS, +25°C 0.005 0.005 % FSR
@ 100 kSPS2 0.01 0.01 % FSR
Temperature Drift, Bipolar Zero3 % FSR
J, K Grades 0.0015 0.0015 % FSR
A, B Grades 0.003 0.003 % FSR
Temperature Drift, Gain3
J, K Grades 0.0015 0.0015 % FSR
A, B Grades 0.003 0.003 % FSR
VOLTAGE REFERENCE INPUT RANGE4 (VREF) 5 10 5 10 V
ANALOG INPUT5
Input Range (VIN) ± VREF ± VREF V
Input Impedance * *
Input Settling Time 2 2 µs
Input Capacitance During Sample 50* 50* pF
Aperture Delay 6 6 ns
Aperture Jitter 100 100 ps
POWER SUPPLIES
Power Supply Rejection
VCC = +12 V ± 5% ±1 ±1 LSB
VEE = –12 V ± 5% ±1 ±1 LSB
VDD = +5 V ± 10% ±1 ±1 LSB
Operating Current
ICC 14.5 18 14.5 18 mA
IEE 14.5 18 14.5 18 mA
IDD 2 5 2 5 mA
Power Consumption 360 480 360 480 mW
NOTES
1
VREF = 5.0 V, Conversion Rate = 83 kSPS unless otherwise noted. Values are post-calibration.
2
Values shown apply to any temperature from TMIN to TMAX after calibration at that temperature.
3
Values shown are based upon calibration at +25°C with no additional calibration at temperature. Values shown are the worst case variation from the value at +25 °C.
4
See “APPLICATIONS” section for recommended voltage reference circuit, and Figure 12 for dynamic performance with other reference voltage values.
5
See “APPLICATIONS” section for recommended input buffer circuit.
*For explanation of input characteristics, see “ANALOG INPUT” section.
Specifications subject to change without notice.

REV. A –3–
AD676
TIMING SPECIFICATIONS(T MIN to TMAX VCC = +12 V 6 5%, VEE = –12 V 6 5%, VDD = +5 V 6 10%, VREF = 10.0 V)1
Parameter Symbol Min Typ Max Units
Conversion Time 2
tC 10 1000 µs
CLK Period3 tCLK 480 ns
Calibration Time tCT 85,530 tCLK
Sampling Time (Included in tC) tS 2 µs
CAL to BUSY Delay tCALB 75 150 ns
BUSY to SAMPLE Delay tBS 2 µs
SAMPLE to BUSY Delay tSB 15 100 ns
CLK HIGH4 tCH 50 ns
CLK LOW4 tCL 50 ns
SAMPLE LOW to 1st CLK Delay tSC 50 ns
SAMPLE LOW tSL 100 ns
Output Delay tOD 125 200 ns
Status Delay tSD 50 ns
CAL HIGH Time tCALH 50 ns
NOTES
1
See the “CONVERSION CONTROL” and “AUTOCALIBRATION” sections for detailed explanations of the above timing.
2
Depends upon external clock frequency; includes acquisition time and conversion time. The maximum conversion time is specified to account for the droop of the
internal sample/hold function. Longer conversion times may degrade performance. See “General Conversion Guidelines” for additional explanation of maximum con-
version time.
3
580 ns is recommended for optimal accuracy over temperature.
4
tCH + t CL = tCLK and must be greater than 480 ns.

t CALH

CAL
t CT
t CALB

t CLK
BUSY
t CH t OD

CLK

t CL

Figure 1. Calibration Timing

tC tC tS
tS tSL tS tSL

SAMPLE SAMPLE
(INPUT) tSC t CL (INPUT) tSC t CL
1 2 3 4 5 13 14 15 16 17 1 2 3 4 5 13 14 15 16 17
CLK CLK
(INPUT) t CLK t CH (INPUT) t CLK t CH
BIT 1 – BIT 16 BIT 1 – BIT 16
(PREVIOUS CONVERSION) (NEW DATA) (PREVIOUS CONVERSION) (NEW DATA)
(OUTPUTS) (OUTPUTS)
t OD t OD
tSD tSD
BUSY tBS BUSY t BS
(OUTPUT) (OUTPUT)

tSB tSB

Figure 2a. General Conversion Timing Figure 2b. Continuous Conversion Timing

–4– REV. A
 AD676
ORDERING GUIDE

Package
Model Temperature Range1 S/(N+D) Max INL Package Description Option2
AD676JD 0°C to +70°C 85 dB Ceramic 28-Pin DIP D-28
AD676KD 0°C to +70°C 87 dB ± 1.5 LSB Ceramic 28-Pin DIP D-28
AD676AD –40°C to +85°C 85 dB Ceramic 28-Pin DIP D-28
AD676BD –40°C to +85°C 87 dB ± 1.5 LSB Ceramic 28-Pin DIP D-28
NOTES
1
For details on grade and package offerings screened in accordance with MIL-STD-883, refer to the AD676/883 data sheet.
2
D = Ceramic DIP.

ABSOLUTE MAXIMUM RATINGS*

VCC to VEE . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +26.4 V
VDD to DGND . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +7 V
VCC to AGND . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +18 V
VEE to AGND . . . . . . . . . . . . . . . . . . . . . . . . –18 V to +0.3 V
AGND to DGND . . . . . . . . . . . . . . . . . . . . . . . . . . . . ± 0.3 V
Digital Inputs to DGND . . . . . . . . . . . . . . . . . . 0 V to +5.5 V
Analog Inputs, VREF to AGND
. . . . . . . . . . . . . . . . . . . . . . . (VCC + 0.3 V) to (VEE – 0.3 V)
Soldering . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +300°C, 10 sec
Storage Temperature . . . . . . . . . . . . . . . . . . –65°C to +150°C
*Stresses greater than those listed under “Absolute Maximum Ratings” may cause
permanent damage to the device. This is a stress rating only and functional
operation of the device at these or any other conditions above those indicated in
the operational section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect device reliability.

CAUTION
The AD676 features input protection circuitry consisting of large “distributed” diodes and
polysilicon series resistors to dissipate both high energy discharges (Human Body Model) and fast,
low energy pulses (Charged Device Model). Per Method 3015.2 of MIL-STD-883C, the AD676
has been classified as a Category 1 Device. WARNING!
Proper ESD precautions are strongly recommended to avoid functional damage or performance
degradation. Charges as high as 4000 volts readily accumulate on the human body and test
ESD SENSITIVE DEVICE
equipment, and discharge without detection. Unused devices must be stored in conductive foam
or shunts, and the foam discharged to the destination socket before devices are removed. For further
information on ESD Precaution. Refer to Analog Devices’ ESD Prevention Manual.

REV. A –5–
AD676
PIN DESCRIPTION

Pin Name Type Description

1–6 BIT 11-BIT 16 DO BIT 11–BIT 16 represent the six LSBs of data.
7 BUSY DO Status Line for Converter. Active HIGH, indicating a conversion or calibration in progress.
BUSY should be buffered when capacitively loaded.
8 CAL DI Calibration Control Pin (Asynchronous).
9 SAMPLE DI VIN Acquisition Control Pin. Active HIGH. During conversion, SAMPLE controls the state
of the internal sample-hold amplifier and the falling edge initiates conversion (see “Conver-
sion Control” paragraph). During calibration, SAMPLE should be held LOW. If HIGH dur-
ing calibration, diagnostic information will appear on the two LSBs (Pins 5 and 6).
10 CLK DI Master Clock Input. The AD676 requires 17 clock cycles to execute a conversion.
11 DGND P Digital Ground.
12 VCC P +12 V Analog Supply Voltage.
13 AGND P/AI Analog Ground.
14 AGND SENSE AI Analog Ground Sense.
15 VIN AI Analog Input Voltage.
16 VREF AI External Voltage Reference Input.
17 VEE P –12 V Analog Supply Voltage. Note: the lid of the ceramic package is internally connected to
VEE.
18 VDD P +5 V Logic Supply Voltage.
19–28 BIT 1–BIT 10 DO BIT 1–BIT 10 represent the ten MSB of data.
Type: AI = Analog Input
DI = Digital Input
DO = Digital Output
P = Power

BIT 11 1 28 BIT 10 VIN 15 ANALOG

CHIP
BIT 12 2 27 BIT 9 AGND SENSE 14 INPUT 16-BIT
COMP
VREF 16 BUFFERS DAC
BIT 13 3 26 BIT 8
AGND 13
BIT 14 4 25 BIT 7 CAL
DAC
BIT 15 5 24 BIT 6
LOGIC & TIMING
BIT 16 (LSB) 6 23 BIT 5
AD676 LEVEL TRANSLATORS
BUSY 7 22 BIT 4
TOP VIEW
CAL 8 (Not to Scale) 21 BIT 3
DIGITAL
SAMPLE CHIP 7 BUSY
9 20 BIT 2
SAR
CLK 10 19 BIT 1 (MSB) CAL 8 1
PAT L
DGND 11 18 VDD MICRO-CODED GEN A 6
SAMPLE 9 T BIT 1 – BIT 16
CONTROLLER 19
VCC 12 17 VEE ALU C
CLK 10 H
28
AGND 13 16 VREF RAM

AGND SENSE 14 15 VIN

AD676

Package Pinout Functional Block Diagram

–6– REV. A
 Definition of Specifications–AD676
NYQUIST FREQUENCY BANDWIDTH
An implication of the Nyquist sampling theorem, the “Nyquist The full-power bandwidth is that input frequency at which the
frequency” of a converter is that input frequency which is one amplitude of the reconstructed fundamental is reduced by 3 dB
half the sampling frequency of the converter. for a full-scale input.

TOTAL HARMONIC DISTORTION INTERMODULATION DISTORTION (IMD)

Total harmonic distortion (THD) is the ratio of the rms sum of With inputs consisting of sine waves at two frequencies, fa and
the harmonic components to the rms value of a full-scale input fb, any device with nonlinearities will create distortion products,
signal and is expressed in percent (%) or decibels (dB). For in- of order (m+n), at sum and difference frequencies of mfa ± nfb,
put signals or harmonics that are above the Nyquist frequency, where m, n = 0, 1, 2, 3. . . . Intermodulation terms are those for
the aliased components are used. which m or n is not equal to zero. For example, the second or-
der terms are (fa + fb) and (fa – fb), and the third order terms
SIGNAL-TO-NOISE PLUS DISTORTION RATIO are (2 fa + fb), (2 fa – fb), (fa + 2 fb) and (fa – 2 fb). The IMD
Signal-to-noise plus distortion is defined to be the ratio of the products are expressed as the decibel ratio of the rms sum of the
rms value of the measured input signal to the rms sum of all measured input signals to the rms sum of the distortion terms.
other spectral components below the Nyquist frequency, includ- The two signals applied to the converter are of equal amplitude,
ing harmonics but excluding dc. and the peak value of their sum is –0.5 dB from full scale. The
IMD products are normalized to a 0 dB input signal.
GAIN ERROR
The last transition should occur at an analog value 1.5 LSB be- APERTURE DELAY
low the nominal full scale (4.99977 volts for a ± 5 V range). The Aperture delay is the time required after SAMPLE pin is taken
gain error is the deviation of the actual difference between the LOW for the internal sample-hold of the AD676 to open, thus
first and last code transition from the ideal difference between holding the value of VlN.
the first and last code transition.
APERTURE JITTER
BIPOLAR ZERO ERROR Aperture jitter is the variation in the aperture delay from sample
Bipolar zero error is the difference between the ideal midscale to sample.
input voltage (0 V) and the actual voltage producing the
midscale output code. POWER SUPPLY REJECTION
DC variations in the power supply voltage will affect the overall
DIFFERENTIAL NONLINEARITY (DNL) transfer function of the ADC, resulting in zero error and gain er-
In an ideal ADC, code transitions are one LSB apart. Differen- ror changes. Power supply rejection is the maximum change in
tial nonlinearity is the maximum deviation from this ideal value. either the bipolar zero error or gain error value. Additionally,
It is often specified in terms of resolution for which no missing there is another power supply variation to consider. AC ripple
codes are guaranteed. on the power supplies can couple noise into the ADC, resulting
in degradation of dynamic performance. This is displayed in
INTEGRAL NONLINEARITY (INL) Figure 16.
The ideal transfer function for an ADC is a straight line bisect-
ing the center of each code drawn between “zero” and “full INPUT SETTLING TIME
scale.” The point used as “zero” occurs 1/2 LSB before the Settling time is a function of the SHA’s ability to track fast
most negative code transition. “Full scale” is defined as a level slewing signals. This is specified as the maximum time required
1.5 LSB beyond the most positive code transition. Integral in track mode after a full-scale step input to guarantee rated
nonlinearity is the worst-case deviation of a code center average conversion accuracy.
from the straight line.

REV. A –7–
AD676
FUNCTIONAL DESCRIPTION LOW and completes in 85,530 clock cycles, indicated by BUSY
The AD676 is a multipurpose 16-bit analog-to-digital converter going LOW. During calibration, it is preferable for SAMPLE to
and includes circuitry which performs an input sample/hold be held LOW. If SAMPLE is HIGH, diagnostic data will appear
function, ground sense, and autocalibration. These functions on Pins 5 and 6. This data is of no value to the user.
are segmented onto two monolithic chips—an analog signal pro- The AD676 requires one clock cycle after BUSY goes LOW to
cessor and a digital controller. Both chips are contained within complete the calibration cycle. If this clock cycle is not pro-
the AD676 package. vided, it will be taken from the first conversion, likely resulting
The AD676 employs a successive-approximation technique to in first conversion error.
determine the value of the analog input voltage. However, in- In most applications, it is sufficient to calibrate the AD676 only
stead of the traditional laser-trimmed resistor-ladder approach, upon power-up, in which case care should be taken that the
this device uses a capacitor-array, charge redistribution tech- power supplies and voltage reference have stabilized first. If not
nique. Binary-weighted capacitors subdivide the input sample to calibrated, the AD676 accuracy may be as low as 10 bits.
perform the actual analog-to-digital conversion. The capacitor
array eliminates variation in the linearity of the device due to CONVERSION CONTROL
temperature-induced mismatches of resistor values. Since a ca- The AD676 is controlled by two signals: SAMPLE and CLK, as
pacitor array is used to perform the data conversions, the shown in Figures 2a and 2b. It is assumed that the part has been
sample/hold function is included without the need for additional calibrated and the digital I/O pins have the levels shown at the
external circuitry. start of the timing diagram.
Initial errors in capacitor matching are eliminated by an auto- A conversion consists of an input acquisition followed by 17
calibration circuit within the AD676. This circuit employs an clock pulses which execute the 16-bit internal successive ap-
on-chip microcontroller and a calibration DAC to measure and proximation routine. The analog input is acquired by taking the
compensate capacitor mismatch errors. As each error is deter- SAMPLE line HIGH for a minimum sampling time of tS. The
mined, its value is stored in on-chip memory (RAM). Subse- actual sample taken is the voltage present on VIN one aperture
quent conversions use these RAM values to improve conversion delay after the SAMPLE line is brought LOW, assuming the
accuracy. The autocalibration routine may be invoked at any previous conversion has completed (signified by BUSY going
time. Autocalibration insures high performance while eliminat- LOW). Care should he taken to ensure that this negative edge is
ing the need for any user adjustments and is described in detail well defined and jitter free in ac applications to reduce the un-
below. certainty (noise) in signal acquisition. With SAMPLE going
The microcontroller controls all of the various functions within LOW, the AD676 commits itself to the conversion—the input at
the AD676. These include the actual successive approximation VIN is disconnected from the internal capacitor array, BUSY
algorithm, the autocalibration routine, the sample/hold opera- goes HIGH, and the SAMPLE input will be ignored until the
tion, and the internal output data latch. conversion is completed (when BUSY goes LOW). SAMPLE
must be held LOW for a minimum period of time tSL. A period
AUTOCALIBRATION of time tSC after bringing SAMPLE LOW, the 17 CLK cycles
The AD676 achieves rated performance without the need for are applied; CLK pulses that start before this period of time are
user trims or adjustments. This is accomplished through the use ignored. BUSY goes HIGH tSB after SAMPLE goes LOW, sig-
of on-chip autocalibration. nifying that a conversion is in process, and remains HIGH until
In the autocalibration sequence, sample/hold offset is nulled by the conversion is completed. BUSY goes LOW during the 17th
internally connecting the input circuit to the ground sense cir- CLK cycle at the point where the data outputs have changed
cuit. The resulting offset voltage is measured and stored in and are valid. The AD676 will ignore CLK after BUSY has
RAM for later use. Next, the capacitor representing the most gone LOW and the output data will remain constant until a new
significant bit (MSB) is charged to the reference voltage. This conversion is completed. The data can, therefore, be read any
charge is then transferred to a capacitor of equal size (composed time after BUSY goes LOW and before the 17th CLK of the
of the sum of the remaining lower weight bits). The difference next conversion (see Figures 2a and 2b). The section on Micro-
in the voltage that results and the reference voltage represents processor Interfacing discusses how the AD676 can be inter-
the amount of capacitor mismatch. A calibration digital-to-ana- faced to a 16-bit databus.
log converter (DAC) adds an appropriate value of error correc- Typically BUSY would be used to latch the AD676 output data
tion voltage to cancel this mismatch. This correction factor is into buffers or to interrupt microprocessors or DSPs. It is rec-
also stored in RAM. This process is repeated for each of the ommended that the capacitive load on BUSY be minimized by
capacitors representing the remaining top eight bits. The accu- driving no more than a single logic input. Higher capacitive
mulated values in RAM are then used during subsequent con- loads such as cables or multiple gates may degrade conversion
versions to adjust conversion results accordingly. quality unless BUSY is buffered.
As shown in Figure 1, when CAL is taken HIGH the AD676 in-
ternal circuitry is reset, the BUSY pin is driven HIGH, and the
ADC prepares for calibration. This is an asynchronous hard-
ware reset and will interrupt any conversion or calibration cur-
rently in progress. Actual calibration begins when CAL is taken

–8– REV. A
 AD676
CONTINUOUS CONVERSION Figure 3 also illustrates the use of a counter (74HC393) to de-
For maximum throughput rate, the AD676 can be operated in a rive the AD676 SAMPLE command from the system clock
continuous convert mode (see Figure 2b). This is accomplished when a continuous convert mode is desirable. Pin 9 (2QC) pro-
by utilizing the fact that SAMPLE will no longer be ignored af- vides a 96 kHz sample rate for the AD676 when used with a
ter BUSY goes LOW, so an acquisition may be initiated even 12.288 MHz system clock. Alternately, Pin 8 (2QD) could be
during the HIGH time of the 17th CLK pulse for maximum used for a 48 kHz rate.
throughput rate while enabling full settling of the sample/hold If a continuous clock is used, then the user must avoid CLK
circuitry. If SAMPLE is already HIGH when BUSY goes LOW edges at the instant of disconnecting VIN which occurs at the
at the end of a conversion, then an acquisition is immediately falling edge of SAMPLE (see tSC specification). The duty cycle
initiated and tS and tC start from that time. Data from the previ- of CLK may vary, but both the HIGH (tCH) and LOW (tCL )
ous conversion may be latched up to tSD before BUSY goes phases must conform to those shown in the timing specifica-
LOW or tOD after the rising edge of the 17th clock pulse. How- tions. The internal comparator makes its decisions on the rising
ever, it is preferred that latching occur on or after the falling edge of CLK. To avoid a negative edge transition disturbing the
edge of BUSY. comparator’s settling, tCL should be at least half the value of tCLK.
Care must he taken to adhere to the minimum/maximum timing To also avoid transitions disturbing the internal comparator’s
requirements in order to preserve conversion accuracy. settling, it is not recommended that the SAMPLE pin change
state toward the end of a CLK cycle.
GENERAL CONVERSION GUIDELINES During a conversion, internal dc error terms such as comparator
During signal acquisition and conversion, care should be taken voltage offset are sampled, stored on internal capacitors and
with the logic inputs to avoid digital feedthrough noise. It is pos- used to correct for their corresponding errors when needed. Be-
sible to run CLK continuously, even during the sample period. cause these voltages are stored on capacitors, they are subject to
However, CLK edges during the sampling period, and especially leakage decay and so require refreshing. For this reason there is
when SAMPLE goes LOW, may inject noise into the sampling a maximum conversion time tC (1000 µs). From the time
process. The AD676 is tested with no CLK cycles during the SAMPLE goes HIGH to the completion of the 17th CLK pulse,
sampling period. The BUSY signal can be used to prevent the no more than 1000 µs should elapse for specified performance.
clock from running during acquisition, as illustrated in Figure 3. However, there is no restriction to the maximum time between
In this circuit BUSY is used to reset the circuitry which divides conversions.
the system clock down to provide the AD676 CLK. This serves
to interrupt the clock until after the input signal has been ac- Output coding for the AD676 is twos complement, as shown in
quired, which has occurred when BUSY goes HIGH. When the Table I. By inverting the MSB, the coding can be converted to
conversion is completed and BUSY goes LOW, the circuit in offset binary. The AD676 is designed to limit output coding in
Figure 3 truncates the 17th CLK pulse width which is tolerable the event of out-of-range inputs.
because only its rising edge is critical.
Table I. Output Coding
11 3Q 2Q 7
VIN Output Code
4 1D 3D 12
12.288MHz >Full Scale 011 . . . 11
9 CLK CLR 1 7 BUSY
SYSTEM Full Scale 011 . . . 11
CLOCK SAMPLE 9 Full Scale – 1 LSB 011 . . . 10
1Q 2 10 CLK
Midscale + 1 LSB 000 . . . 01
Midscale 000 . . . 00
74HC175
2D 5 Midscale – 1 LSB 111 . . . 11
AD676
–Full Scale + 1 LSB 100 . . . 01
–Full Scale 100 . . . 00
1 1CLK
<–Full Scale 100 . . . 00
2QC 9
13 2CLK
2QD 8
6 1QD

12 2CLR

2 1CLR

74HC393

Figure 3.

REV. A –9–
AD676
POWER SUPPLIES AND DECOUPLING Additionally, it is beneficial to have large capacitors (>47 µF)
The AD676 has three power supply input pins. VCC and VEE located at the point where the power connects to the PCB with
provide the supply voltages to operate the analog portions of the 10 µF capacitors located in the vicinity of the ADC to further
AD676 including the ADC and sample-hold amplifier (SHA). reduce low frequency ripple. In systems that will be subjected to
VDD provides the supply voltage which operates the digital por- particularly harsh environmental noise, additional decoupling
tions of the AD676 including the data output buffers and the may be necessary. RC-filtering on each power supply combined
autocalibration controller. with dedicated voltage regulation can substantially decrease
As with most high performance linear circuits, changes in the power supply ripple effects (this is further detailed in Figure 7).
power supplies can produce undesired changes in the perfor-
mance of the circuit. Optimally, well regulated power supplies BOARD LAYOUT
with less than 1% ripple should be selected. The ac output im- Designing with high resolution data converters requires careful
pedance of a power supply is a complex function of frequency, attention to board layout. Trace impedance is a significant issue.
and in general will increase with frequency. In other words, high A 1.22 mA current through a 0.5 Ω trace will develop a voltage
frequency switching such as that encountered with digital cir- drop of 0.6 mV, which is 4 LSBs at the 16-bit level for a 10 V
cuitry requires fast transient currents which most power supplies full-scale span. In addition to ground drops, inductive and ca-
cannot adequately provide. This results in voltage spikes on the pacitive coupling need to be considered, especially when high
supplies. If these spikes exceed the ± 5% tolerance of the ± 12 V accuracy analog signals share the same board with digital
supplies or the ± 10% limits of the +5 V supply, ADC perfor- signals.
mance will degrade. Additionally, spikes at frequencies higher Analog and digital signals should not share a common return
than 100 kHz will also degrade performance. To compensate for path. Each signal should have an appropriate analog or digital
the finite ac output impedance of the supplies, it is necessary to return routed close to it. Using this approach, signal loops en-
store “reserves” of charge in bypass capacitors. These capacitors close a small area, minimizing the inductive coupling of noise.
can effectively lower the ac impedance presented to the AD676 Wide PC tracks, large gauge wire, and ground planes are highly
power inputs which in turn will significantly reduce the magni- recommended to provide low impedance signal paths. Separate
tude of the voltage spikes. For bypassing to be effective, certain analog and digital ground planes are also desirable, with a single
guidelines should be followed. Decoupling capacitors, typically interconnection point at the AD676 to minimize interference
0.1 µF, should be placed as closely as possible to each power between analog and digital circuitry. Analog signals should be
supply pin of the AD676. It is essential that these capacitors be routed as far as possible from digital signals and should cross
placed physically close to the IC to minimize the inductance of them, if at all, only at right angles. A solid analog ground plane
the PCB trace between the capacitor and the supply pin. The around the AD676 will isolate it from large switching ground
logic supply (VDD) should be decoupled to digital common and currents. For these reasons, the use of wire wrap circuit con-
the analog supplies (Vcc and VEE) to analog common. The ref- struction will not provide adequate performance; careful printed
erence input is also considered as a power supply pin in this re- circuit board construction is preferred.
gard and the same decoupling procedures apply. These points
are displayed in Figure 4. GROUNDING
The AD676 has three grounding pins, designated ANALOG
GROUND (AGND), DIGITAL GROUND (DGND) and
+5V 18 VDD AD676 ANALOG GROUND SENSE (AGND SENSE). The analog
ground pin is the “high quality” ground reference point for the
DGND AGND VCC VEE VREF
0.1µF
device, and should be connected to the analog common point in
11 13 12 17 11
the system.
0.1µF
AGND SENSE is intended to be connected to the input signal
0.1µF ground reference point. This allows for slight differences in level
between the analog ground point in the system and the input
0.1µF signal ground point. However no more than 100 mV is recom-
SYSTEM SYSTEM 12V –12V mended between the AGND and the AGND SENSE pins for
DIGITAL ANALOG
COMMON COMMON
specified performance.

Figure 4. Grounding and Decoupling the AD676

–10– REV. A
 AD676
Using AGND SENSE to remotely sense the ground potential of VOLTAGE REFERENCE
the signal source can be useful if the signal has to be carried The AD676 requires the use of an external voltage reference.
some distance to the A/D converter. Since all IC ground cur- The input voltage range is determined by the value of the refer-
rents have to return to the power supply and no ground leads ence voltage; in general, a reference voltage of n volts allows an
are free from resistance and inductance, there are always some input range of ± n volts. The AD676 is specified for both 10 V
voltage differences from one ground point in a system to and 5.0 V references. A 10 V reference will typically require
another. support circuitry operated from ± 15 V supplies; a 5.0 V refer-
Over distance this voltage difference can easily amount to sev- ence may be used with ± 12 V supplies. Signal-to-noise perfor-
eral LSBs (in a 10 V input span, 16-bit system each LSB is mance is increased proportionately with input signal range. In
about 0.15 mV). This would directly corrupt the A/D input sig- the presence of a fixed amount of system noise, increasing the
nal if the A/D measures its input with respect to power ground LSB size (which results from increasing the reference voltage)
(AGND) as shown in Figure 5a. To solve this problem the will increase the effective S/(N+D) performance. Figure 12
AD676 offers an AGND SENSE pin. Figure 5b shows how the illustrates S/(N+D) as a function of reference voltage. In
AGND SENSE can be used to eliminate the problem in Figure contrast, INL will be optimal at lower reference voltage values
5a. Figure 5b also shows how the signal wires should be (such as 5 V) due to capacitor nonlinearity at higher voltage
shielded in a noisy environment to avoid capacitive coupling. If values.
inductive (magnetic) coupling is expected to be dominant such During a conversion, the switched capacitor array of the AD676
as where motors are present, twisted-pair wires should be used presents a dynamically changing current load at the voltage ref-
instead. erence as the successive-approximation algorithm cycles through
The digital ground pin is the reference point for all of the digital various choices of capacitor weighting. (See the following sec-
signals that operate the AD676. This pin should be connected tion “Analog Input” for a detailed discussion of the VREF input
to the digital common point in the system. As Figure 4 illus- characteristics.) The output impedance of the reference circuitry
trated, the analog and digital grounds should be connected to- must be low so that the output voltage will remain sufficiently
gether at one point in the system, preferably at the AD676. constant as the current drive changes. In some applications, this
may require that the output of the voltage reference be buffered
AD676
by an amplifier with low impedance at relatively high frequen-
cies. In choosing a voltage reference, consideration should be
VIN
made for selecting one with low noise. A capacitor connected
SOURCE between REF IN and AGND will reduce the demands on the
VS
reference by decreasing the magnitude of high frequency com-
∆V AGND ponents required to be sourced by the reference.
TO POWER Figures 6 and 7 represent typical design approaches.
SUPPLY GND
GROUND LEAD I GROUND > 0 +12V

Figure 5a. Input to the A/D Is Corrupted by IR Drop in 2

Ground Leads: VIN = VS + ∆V VIN

8 AD586 6 16 VREF
CN
SHIELDED CABLE AD676 +
1.0µF 10µF
VIN
AGND 4
SOURCE
SENSE 13 AGND
VS

AD676
AGND

TO POWER
SUPPLY GND Figure 6.
GROUND LEAD I GROUND > 0
Figure 6 shows a voltage reference circuit featuring the 5 V out-
Figure 5b. AGND SENSE Eliminates the Problem in put AD586. The AD586 is a low cost reference which utilizes a
Figure 5a. buried Zener architecture to provide low noise and drift. Over
the 0°C to +70°C range, the AD586L grade exhibits less than
2.25 mV output change from its initial value at +25°C. A noise-
reduction capacitor, CN, reduces the broadband noise of the

REV. A –11–
AD676
AD586 output, thereby optimizing the overall performance of The AD676 analog inputs (VIN, VREF and AGND SENSE) ex-
the AD676. It is recommended that a 10 µF to 47 µF high qual- hibit dynamic characteristics. When a conversion cycle begins,
ity tantalum capacitor be tied between the VREF input of the each analog input is connected to an internal, discharged 50 pF
AD676 and ground to minimize the impedance on the capacitor which then charges to the voltage present at the corre-
reference. sponding pin. The capacitor is disconnected when SAMPLE is
taken LOW, and the stored charge is used in the subsequent
AD587 conversion. In order to limit the demands placed on the external
10Ω VO 6 source by this high initial charging current, an internal buffer
2 VIN
10µF
amplifier is employed between the input and this capacitance for
NR 8
0.1µF GND a few hundred nanoseconds. During this time the input pin ex-
4 1µF
hibits typically 20 kΩ input resistance, 10 pF input capacitance
and ± 40 µA bias current. Next, the input is switched directly to
10Ω 0.1µF
the now precharged capacitor and allowed to fully settle. During
+15V 78L12 this time the input sees only a 50 pF capacitor. Once the sample
100µF 0.01µF 10µF 12 is taken, the input is internally floated so that the external input
VCC VREF 16
10Ω source sees a very high input resistance and a parasitic input ca-
+5V 18 VDD pacitance of typically only 2 pF. As a result, the only dominant
AD676 10µF
100µF 0.1µF
VEE VIN
input characteristic which must be considered is the high cur-
rent steps which occur when the internal buffers are switched in
10 Ω 17 15
–15V 79L12
and out.
10µF 0.1µF
100µF 0.01µF In most cases, these characteristics require the use of an external
op amp to drive the input of the AD676. Care should he taken
VIN
with op amp selection; even with modest loading conditions,
most available op amps do not meet the low distortion require-
Figure 7. ments necessary to match the performance capabilities of the
Using the AD676 with ± 10 V input range (VREF = 10 V) typi- AD676. Figure 8 represents a circuit, based upon the AD845,
cally requires ± 15 V supplies to drive op amps and the voltage recommended for low noise, low distortion ac applications.
reference. If ± 12 V is not available in the system, regulators For applications optimized more for low bias and low offset than
such as 78L12 and 79L12 can be used to provide power for the speed or bandwidth, the AD845 of Figure 8 may be replaced by
AD676. This is also the recommended approach (for any input the OP27.
range) when the ADC system is subjected to harsh environ-
ments such as where the power supplies are noisy and where 1kΩ

voltage spikes are present. Figure 7 shows an example of such a +12V

system based upon the 10 V AD587 reference, which provides a ±5V 0.1µF
300 µV LSB. Circuitry for additional protection against power INPUT 1kΩ AD676
2
supply disturbances has been shown. A 100 µF capacitor at each 7

regulator prevents very large voltage spikes from entering the 499Ω AD845 6 15 VIN

regulators. Any power line noise which the regulators cannot 3 4 0.1µF

eliminate will be further filtered by an RC filter (10 Ω/10 µF)

having a –3 dB point at 1.6 kHz. For best results the regulators 13 AGND
–12V
should be within a few centimeters of the AD676.
AGND
14
SENSE
ANALOG INPUT
As previously discussed, the analog input voltage range for the
AD676 is ± VREF. For purposes of ground drop and common
mode rejection, the VIN and VREF inputs each have their own Figure 8.
ground. VREF is referred to the local analog system ground
(AGND), and VIN is referred to the analog ground sense pin
(AGND SENSE) which allows a remote ground sense for the
input signal.

–12– REV. A
 AD676
AC PERFORMANCE This limit is described by S/(N+D) = (6.02n + 1.76 + 10 log
AC parameters, which include S/(N+D), THD, etc., reflect the FS/2FA) dB, where n is the resolution of the converter in bits, FS
AD676’s effect on the spectral content of the analog input sig- is the sampling frequency, and Fa is the signal bandwidth of in-
nal. Figures 12 through 16 provide information on the AD676’s terest. For audio bandwidth applications, the AD676 is capable
ac performance under a variety of conditions. of operating at a 2 3 oversample rate (96 kSPS), which typically
As a general rule, averaging the results from several conversions produces an improvement in S/(N+D) of 3 dB compared with
reduces the effects of noise, and therefore improves such param- operating at the Nyquist conversion rate of 48 kSPS. Over-
eters as S/(N+D). AD676 performance may be optimized by sampling has another advantage as well; the demands on the
operating the device at its maximum sample rate of 100 kSPS antialias filter are lessened. In summary, system performance is
and digitally filtering the resulting bit stream to the desired signal optimized by running the AD676 at or near its maximum sam-
bandwidth. This succeeds in distributing noise over a wider pling rate of 100 kHz and digitally filtering the resulting spec-
frequency range, thus reducing the noise density in the fre- trum to eliminate undesired frequencies.
quency band of interest. This subject is discussed in the follow-
ing section. DC CODE UNCERTAINTY
Ideally, a fixed dc input should result in the same output code
OVERSAMPLING AND NOISE FILTERING for repetitive conversions. However, as a consequence of system
The Nyquist rate for a converter is defined as one-half its sam- noise and circuit noise, for a given input voltage there is a range
pling rate. This is established by the Nyquist theorem, which re- of output codes which may occur. Figure 9 is a histogram of the
quires that a signal he sampled at a rate corresponding to at codes resulting from 1000 conversions of a typical input voltage
least twice its highest frequency component of interest in order by the AD676 used with a 10 V reference.
to preserve the informational content. Oversampling is a conver- 800
sion technique in which the sampling frequency is more than
twice the frequency bandwidth of interest. In audio applications,
the AD676 can operate at a 2 3 FS oversampling rate, where 600

NUMBER OF CODE HITS

FS = 48 kHz.
In quantized systems, the informational content of the analog
input is represented in the frequency spectrum from dc to the 400
Nyquist rate of the converter. Within this same spectrum are
higher frequency noise and signal components. Antialias, or low
pass, filters are used at the input to the ADC to reduce these 200
noise and signal components so that their aliased components
do not corrupt the baseband spectrum. However, wideband
noise contributed by the AD676 will not be reduced by the
0
antialias filter. The AD676 quantization noise is evenly distrib- –1 0 1 2
uted from dc to the Nyquist rate, and this fact can be used to DEVIATION FROM CORRECT CODE – LSBs
minimize its overall affect.
The AD676 quantization noise effects can be reduced by Figure 9. Distribution of Codes from 1000 Conversions,
oversampling–sampling at a rate higher than that defined by the Relative to the Correct Code
Nyquist theorem. This spreads the noise energy over a band- The standard deviation of this distribution is approximately 0.5
width wider than the frequency band of interest. By judicious LSBs. If less uncertainty is desired, averaging multiple conver-
selection of a digital decimation filter, noise frequencies outside sions will narrow this distribution by the inverse of the square
the bandwidth of interest may be eliminated. root of the number of samples; i.e., the average of 4 conversions
The process of analog to digital conversion inherently produces would have a standard deviation of 0.25 LSBs.
noise, known as quantization noise. The magnitude of this noise
is a function of the resolution of the converter, and manifests it-
self as a limit to the theoretical signal-to-noise ratio achievable.

REV. A –13–
AD676
MICROPROCESSOR INTERFACE The AD676 CLK and SAMPLE can be generated by dividing
The AD676 is ideally suited for use in both traditional dc mea- down the system clock as described earlier (Figure 3), or if the
surement applications supporting a microprocessor, and in ac ADSP-2101 serial port clocks are not being used, they can be
signal processing applications interfacing to a digital signal pro- programmed to generate CLK and SAMPLE.
cessor. The AD676 is designed to interface with a 16-bit data
bus, providing all output data bits in a single read cycle. A vari-
A13
ety of external buffers, such as 74HC541, can be used with the
A12
AD676 to provide 3-state outputs, high driving capability, and
to prevent bus noise from coupling into the ADC. The following CS
sections illustrate the use of the AD676 with a representative A11
digital signal processor and microprocessor. These circuits pro-
DMS
vide general interface practices which are applicable to other
processor choices. Figure 10b.

ADSP-2101 80286
Figure 10a shows the AD676 interfaced to the ADSP-2101 DSP The 80286 16-bit microprocessor can be interfaced to a buff-
processor. The AD676 buffers are mapped in the ADSP-2101’s ered AD676 without any generation of wait states. As seen in
memory space, requiring one wait state when using a 12.5 MHz Figure 11, BUSY can be used both to control the AD676 clock
processor clock. and to alert the processor when new data is ready. In the system
The falling edge of BUSY interrupts the processor, indicating shown, the 80286 should be configured in an edge triggered, di-
that new data is ready. The ADSP-2101 automatically jumps to rect interrupt mode (integrated controller provides the interrupt
the appropriate service routine with minimal overhead. The in- vector). Since the 80286 does not latch interrupt signals, the in-
terrupt routine then instructs the processor to read the new data terrupt needs to be internally acknowledged before BUSY goes
using a memory read instruction. HIGH again during the next AD676 conversion (BUSY = 0).
Depending on whether the AD676 buffers are mapped into
IRQ2 memory or 1/0 space, the interrupt service routine will read the
A0
data by using either the MOV or the IN instruction. To be able
ADDRESS BUS to read all the 16 bits at once, and thereby increase the 80286’s
A13 efficiency, the buffers should be located at an even address.
RD CS G1
DECODER
DMS AD0 – AD15 Y1 – Y8
G1 8
16 A1 – A8
D8 – D23 Y1 – Y8 8
8 RD CS 74HC541
16 G2
A1 – A3 PCSO – 6
ADSP-2101 DECODER
BUSY ALE BIT1 – BIT16
8
74HC541 S2 G1 16
G2 Y1 – Y8
8
BIT 1 – BIT 16 A1 – A8
80286 8 AD676
G1
16
74HC541
Y1 – Y8 G2
8
AD676 CLKOUT DIVIDER SAMPLE
A1 – A3 2MHz
8 D Q D Q CLK
74HC541
Q Q BUSY
G2 CLR CLR
INT 0
74HC04 74HC74
Figure 10a.
Figure 10b shows circuitry which would be included by a typical
address decoder for the output buffers. In this case, a data Figure 11.
memory access to any address in the range 3000H to 37FFH
will result in the output buffers being enabled.

–14– REV. A
 Typical Dynamic Performance– AD676
102 100

100
THD 90
98 THD
96 80

94
70
92

dB
dB

S/(N+D)
90 S/(N+D) 60
88
50
86

84
40
82

80 30
2.5 3.5 4.5 5.5 6.5 7.5 8.5 9.5 10.5 –60 –50 –40 –30 –20 –10 0

VREF – Volts INPUT AMPLITUDE – dB

Figure 12. S/(N+D) and THD vs. VREF Figure 13. S/(N+D) and THD vs. Input Amplitude

Figure 14. 4096 Point FFT at 96 kSPS, fIN = 1.06 kHz Figure 15. IMD Plot for fIN = 1008 Hz (fa),
1055 Hz (fb) at 96 kSPS

+5V
90

80
+12V
70
–12V
S/(N+D) –dB

60

50

40

30

20
0 100 1k 10k 100k 1M
RIPPLE FREQUENCY – Hz

Figure 16. AC Power Supply Rejection (fIN = 1.06 kHz)

fSAMPLE = 96 kSPS, VRIPPLE = 0.13 V p-p

REV. A –15–
AD676
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).

28-Pin Ceramic DIP Package (D-28)

0.005 (0.13) 0.100 (2.54)

C1679–24–7/92
MIN MAX

28 15

1 14

0.610 (15.49)
1.490 (37.85) MAX 0.060 (1.52)
0.500 (12.70)
0.015 (0.38)
0.225 (5.72)
MAX
0.150 (3.81) 0.018 (0.46)
MIN 0.008 (0.20)
0.620 (15.75)
0.070 (1.78)
0.590 (14.99)
0.200 (5.08) 0.026 (0.66) 0.100 (2.54) 0.030 (0.76)
0.125 (3.18) 0.014 (0.36) BSC

PRINTED IN U.S.A.

–16– REV. A